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Search Results (469)

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25 pages, 954 KiB  
Article
SAC-Based Intelligent Load Relief Attitude Control Method for Launch Vehicles
by Shou Zhou, Hao Yang, Shifeng Zhang, Xibin Bai and Feng Wang
Aerospace 2025, 12(3), 203; https://doi.org/10.3390/aerospace12030203 - 28 Feb 2025
Viewed by 165
Abstract
This paper proposes an intelligent control method based on Soft Actor-Critic (SAC) to address uncertainties faced by flight vehicles during flight. The method effectively reduces aerodynamic loads and enhances the reliability of structural strength under significant wind disturbances. A specific launch vehicle is [...] Read more.
This paper proposes an intelligent control method based on Soft Actor-Critic (SAC) to address uncertainties faced by flight vehicles during flight. The method effectively reduces aerodynamic loads and enhances the reliability of structural strength under significant wind disturbances. A specific launch vehicle is taken as the research subject, and its dynamic model is established. A deep reinforcement learning (DRL) framework suitable for the attitude control problem is constructed, along with a corresponding training environment. A segmented reward function is designed: the initial stage emphasizes tracking accuracy, the middle stage, with a detrimental effect due to the high-altitude wind region, focuses on load relief, and the final stage gradually resumes following tracking accuracy on the basis of maintaining the effect of load relief. The reward function dynamically switches between stages using a time factor. The improved SAC algorithm is employed to train the agent over multiple epochs, ultimately resulting in an intelligent load relief attitude controller applicable to the launch vehicle. Simulation experiments demonstrate that this method effectively solves the attitude control problem under random wind disturbances, particularly reducing the aerodynamic loads of launch vehicles in the high-altitude wind region. Full article
(This article belongs to the Section Aeronautics)
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Figure 1

Figure 1
<p>Force analysis diagram of the launch vehicle.</p>
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<p>Illustration of RL principles.</p>
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<p>Flowchart of the SAC Algorithm.</p>
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<p>The variation curves of the reward weights over time.</p>
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<p>Diagram of the training environment for the simulation in Pytorch.</p>
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<p>Mean reward curve during agent training without wind disturbances.</p>
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<p>Vertical wind shear curve.</p>
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<p>Mean reward curve during agent training with wind disturbances.</p>
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<p>Comparison of SAC and PID controllers in the no-wind scenario.</p>
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<p>Comparison of SAC and PID controllers under the maximum wind disturbance.</p>
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21 pages, 5943 KiB  
Article
Application of a Soft-Switching Adaptive Kalman Filter for Over-Range Measurements in a Low-Frequency Extension of MHD Sensors
by Junze Tong, Shaocen Shi, Fuchao Wang and Dapeng Tian
Aerospace 2025, 12(3), 192; https://doi.org/10.3390/aerospace12030192 - 27 Feb 2025
Viewed by 192
Abstract
The increasing demand for image quality in aerospace remote sensing has led to higher performance requirements for inertial stabilization platforms equipped with image sensors, particularly in terms of bandwidth. To achieve wide-bandwidth control in optical stabilization platforms, engineers employ magneto-hydrodynamic (MHD) sensors as [...] Read more.
The increasing demand for image quality in aerospace remote sensing has led to higher performance requirements for inertial stabilization platforms equipped with image sensors, particularly in terms of bandwidth. To achieve wide-bandwidth control in optical stabilization platforms, engineers employ magneto-hydrodynamic (MHD) sensors as key components to enhance system performance because of their wide measurement bandwidth (5–1000 Hz). While MHD sensors offer a wide-frequency response, they are limited by a narrow measuring range and low sensitivity at low frequencies, making them unsuitable as standalone sensors. To address the challenges of over-range measurement and the loss of low-frequency signals, in this study, we developed a soft-switching adaptive Kalman filter method, which enables us to dynamically adjust the fusion weights in the Kalman filter so we can obtain wide-band measurement signals even when the MHD sensor experiences over-range conditions. The proposed method was validated with fusion experiments involving a fiber-optic gyroscope and an MHD sensor; the results demonstrate its ability to expand the sensing bandwidth, regardless of the operating conditions of the MHD sensor. Full article
(This article belongs to the Topic Multi-Sensor Integrated Navigation Systems)
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Figure 1
<p>The mechanism of an MHD sensor.</p>
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<p>Adaptive function <math display="inline"><semantics> <mrow> <mi>f</mi> <mo>(</mo> <mi>z</mi> <mo>)</mo> </mrow> </semantics></math>.</p>
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<p>Simulation results for fusion of sensor measurements without over-range. (<b>a</b>) Original velocity and measurement results, (<b>b</b>) fusion results, (<b>c</b>) comparison of fusion error and measurement error, and (<b>d</b>) statistics of fusion error.</p>
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<p>Simulation results for fusion of sensor measurements with over-range. (<b>a</b>) Original velocity and measurement results, (<b>b</b>) fusion results, (<b>c</b>) comparison of fusion error and measurement error, and (<b>d</b>) power spectrum of fusion errors.</p>
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<p>Simulation results for SSAKF of sensor measurements with over-range. (<b>a</b>) Fusion results and (<b>b</b>) comparison of fusion error and measurement error.</p>
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<p>Frequency identification result. (<b>a</b>) FOG frequency response and (<b>b</b>) MHD frequency response.</p>
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<p>Complementary filtering method.</p>
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<p>Closed-loop control filtering method.</p>
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<p>Experimental setup diagram.</p>
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<p>Static noise PSD analysis.</p>
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<p>Composite–frequency signal fusion. (<b>a</b>) General view. (<b>b</b>) Details of each signal.</p>
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<p>Integration of composite–frequency signal fusion.</p>
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<p>Random signal fusion. (<b>a</b>) Normal condition. (<b>b</b>) Over-range condition.</p>
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<p>Time–domain response signal of noise sweep.</p>
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<p>Frequency–domain Bode plots. (<b>a</b>) Magnitude response. (<b>b</b>) Phase response.</p>
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21 pages, 2535 KiB  
Article
A Bidirectional Resonant Converter Based on Partial Power Processing
by Junfeng Liu, Zhouzhou Wu and Qinglin Zhao
Electronics 2025, 14(5), 910; https://doi.org/10.3390/electronics14050910 - 25 Feb 2025
Viewed by 158
Abstract
This article proposes a bidirectional half-bridge resonant converter based on partial power regulation. The converter adopts an LLC converter as a DC-DC transformer (LLC-DCX) in the main power circuit and works in the open loop at the resonant frequency to give full play [...] Read more.
This article proposes a bidirectional half-bridge resonant converter based on partial power regulation. The converter adopts an LLC converter as a DC-DC transformer (LLC-DCX) in the main power circuit and works in the open loop at the resonant frequency to give full play to the performance advantages of the LLC resonant converter. The partial power regulation circuit incorporates a synchronous Buck converter, enabling forward and backward power transmission by controlling the power flow direction. The converter achieves soft switching in both forward and backward directions, thereby reducing switching losses and enhancing conversion efficiency. Compared with the LLC-DCX converter, this converter can achieve wide voltage gain regulation while having high efficiency, which makes it suitable for charge–discharge applications between energy storage systems and DC Buses. In order to verify the performance of the proposed converter, a 1 kW prototype was constructed, maintaining a constant primary voltage of 400 V and a secondary voltage range of 350 V to 450 V. Experimental results indicate that the prototype achieves peak efficiencies of 97.74% in forward operation and 96.92% in backward operation, thoroughly demonstrating the feasibility and effectiveness of the proposed converter. Full article
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Figure 1

Figure 1
<p>Partial power regulation architecture.</p>
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<p>Voltage and power distribution of the converter.</p>
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<p>Bidirectional half-bridge LLC resonant circuit (<b>a</b>) Main circuit topology. (<b>b</b>) LLC fundamental equivalent circuit.</p>
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<p>The main waveform of the split-capacitor LLC volt-doubling resonant converter. (<b>a</b>) Key forward waveform. (<b>b</b>) Key backward waveform.</p>
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<p>Partial power-processing bidirectional resonant converter architecture.</p>
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<p>Key waveform of Buck converter in FCCM mode.</p>
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<p>Mode of synchronous Buck converter in FCCM mode. (<b>a</b>) Mode 1 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>0</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>1</mn> </msub> </semantics></math>]. (<b>b</b>) Mode 2 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>1</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>2</mn> </msub> </semantics></math>]. (<b>c</b>) Mode 3 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>2</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>3</mn> </msub> </semantics></math>]. (<b>d</b>) Mode 4 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>3</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>4</mn> </msub> </semantics></math>]. (<b>e</b>) Mode 5 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>4</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>5</mn> </msub> </semantics></math>]. (<b>f</b>) Mode 6 [<math display="inline"><semantics> <msub> <mi>t</mi> <mn>5</mn> </msub> </semantics></math>, <math display="inline"><semantics> <msub> <mi>t</mi> <mn>6</mn> </msub> </semantics></math>].</p>
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<p>Partial power regulation circuit gain characteristic diagram.</p>
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<p>Photograph of the designed converter prototype.</p>
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<p>The key waveforms of the LLC-DCX circuit under the output condition of 400 V/2.22 A. (<b>a</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>2</mn> </msub> </semantics></math> and resonant current. (<b>b</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>4</mn> </msub> </semantics></math> and resonant current. (<b>c</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>6</mn> </msub> </semantics></math> and resonant current.</p>
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<p>The key waveforms of synchronous Buck circuit under different output voltages (<math display="inline"><semantics> <msub> <mi>I</mi> <mrow> <mi>o</mi> <mn>2</mn> </mrow> </msub> </semantics></math> = 2.22 A). (<b>a</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>7</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 350 V). (<b>b</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>7</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 400 V). (<b>c</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>7</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 450 V).</p>
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<p>The key waveforms of the LLC-DCX circuit under the output condition of 400V/2.22A. (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 400 V). (<b>a</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>2</mn> </msub> </semantics></math> and resonant current. (<b>b</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>4</mn> </msub> </semantics></math> and resonant current. (<b>c</b>) Key waveforms of <math display="inline"><semantics> <msub> <mi>S</mi> <mn>6</mn> </msub> </semantics></math> and resonant current.</p>
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<p>The key waveforms of the synchronous Boost circuit with different input parameters on the secondary side. (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>1</mn> </msub> </semantics></math> = 400 V). (<b>a</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>8</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 350 V). (<b>b</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>8</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 400 V). (<b>c</b>) <math display="inline"><semantics> <msub> <mi>S</mi> <mn>8</mn> </msub> </semantics></math> and inductor current waveforms (<math display="inline"><semantics> <msub> <mi>U</mi> <mn>2</mn> </msub> </semantics></math> = 450 V).</p>
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<p>The distribution of power in the forward mode.</p>
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<p>Constant-current–constant-voltage charging efficiency curve in forward mode.</p>
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<p>Power distribution in the backward mode.</p>
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<p>Efficiency graph of the converter during backward operation.</p>
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26 pages, 14205 KiB  
Article
Design and Implementation of a DC–DC Resonant LLC Converter for Electric Vehicle Fast Chargers
by Joao Rocha, Saghir Amin, Sergio Coelho, Gonçalo Rego, Joao L. Afonso and Vitor Monteiro
Energies 2025, 18(5), 1099; https://doi.org/10.3390/en18051099 - 24 Feb 2025
Viewed by 239
Abstract
This article presents the design and implementation of a DC–DC power converter for application in electric vehicle (EV) fast-charging systems. The prototype is of the resonant LLC type and consists of a high-power transformer operating at high frequency, which is an essential feature [...] Read more.
This article presents the design and implementation of a DC–DC power converter for application in electric vehicle (EV) fast-charging systems. The prototype is of the resonant LLC type and consists of a high-power transformer operating at high frequency, which is an essential feature for the adequate behavior of the EV fast-charging system as a whole. As demonstrated throughout the article, by using this converter topology as well as its specific operating modes, such as for achieving zero-voltage switching (ZVS) and zero-current switching (ZCS), it is possible to enhance efficiency by reducing conduction and switching losses as well as to increase power density. The details of the high-power high-frequency transformer (HFT), considering different designs, are presented and discussed. With the implemented laboratorial prototype fully developed with silicon carbide (SiC) power semiconductor devices, it was possible to demonstrate and validate the main features of the resonant LLC converter, including high efficiency, under distinct conditions of operation. Full article
Show Figures

Figure 1

Figure 1
<p>Global EV stock over the last 10 years (2013–2023) [<a href="#B3-energies-18-01099" class="html-bibr">3</a>].</p>
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<p>Composition of an EV fast charger, based on an AC–DC three-phase and an isolated DC–DC converter.</p>
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<p>Block diagram of an EV fast-charging system highlighting the internal constitution of the resonant LLC converter.</p>
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<p>Topology of the DC–DC resonant LLC converter.</p>
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<p>Plots of different effects in the curve gain: (<b>a</b>) <span class="html-italic">m</span> effect; (<b>b</b>) <span class="html-italic">Q</span> effect.</p>
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<p>Representation of the principle of operation with hard switching.</p>
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<p>Representation of the principle of operation with soft switching.</p>
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<p>Examples of switching techniques: (<b>a</b>) ZVS switching; (<b>b</b>) ZCS switching.</p>
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<p>Equivalent electric circuit of the HFT.</p>
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<p>Possibilities for the arrangement of the windings in the HFT: (<b>a</b>) Proposed design for HFT1, with one primary and one secondary winding together in one part of the transformer and one primary and one secondary winding separated. (<b>b</b>) Proposed design for HFT2, with the secondary winding wound on top of the primary winding on both sides.</p>
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<p>Simulation results showing the voltage variation in <span class="html-italic">C<sub>r</sub></span> (<span class="html-italic">V<sub>cr</sub></span>) for different case studies: (<b>a</b>) Case 1, with peak voltage value equal to 87 V; (<b>b</b>) Case 2, with peak voltage value equal to 205 V; (<b>c</b>) Case 3, with peak voltage value equal to 318 V.</p>
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<p>Simulation results showing the current variation (<span class="html-italic">I<sub>lr</sub></span>) for different case studies: (<b>a</b>) Case 1, with peak current value equal to 44 A; (<b>b</b>) Case 2, with peak voltage value equal to 44.2 A; (<b>c</b>) Case 3, with peak voltage value equal to 44.3 A.</p>
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<p>Simulation results showing the voltage waveforms for the DC–DC resonant LLC converter at (<b>a</b>) the input side (<span class="html-italic">V<sub>in</sub></span>) and (<b>b</b>) the output side (<span class="html-italic">V<sub>out</sub></span>).</p>
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<p>Simulation results showing the voltage waveforms on the HFT: (<b>a</b>) Primary side (<span class="html-italic">V<sub>prim</sub></span>); (<b>b</b>) Secondary side (<span class="html-italic">V<sub>sec</sub></span>).</p>
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<p>Simulation results showing the voltage (<span class="html-italic">V<sub>DSq3</sub></span>) and current (<span class="html-italic">I<sub>DSq3</sub></span>) waveforms on the SiC power device <span class="html-italic">q3</span>, which validates the ZVS and ZCS operation.</p>
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<p>Simulation results showing the influence of decreasing the operating power.</p>
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<p>Simulation results showing the influence of increasing the operating power.</p>
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<p>Simulation results show the influence of decreasing the <span class="html-italic">f<sub>sw</sub></span> on the system (50 kHz).</p>
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<p>Simulation results showing the influence of increasing the <span class="html-italic">f<sub>sw</sub></span> on the system (150 kHz).</p>
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<p>A 3D mold was implemented to create the ‘O’-shaped core with the dimensions 17 cm long, 12 cm wide, 6 cm deep, and 3 mm material thickness.</p>
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<p>Three different designs according to the shape of the body: (<b>a</b>) Design 5, (<b>b</b>) Design 8, (<b>c</b>) Design 3.</p>
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<p>Values obtained through Design 1 tests: (<b>a</b>) <span class="html-italic">L<sub>o</sub></span><sub>1</sub> equal to 144 µH; (<b>b</b>) <span class="html-italic">L<sub>o</sub></span><sub>2</sub> equal to 154 µH; (<b>c</b>) <span class="html-italic">L<sub>s</sub></span><sub>1</sub> equal to 1.47 µH.</p>
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<p>Values obtained through Design 4 tests: (<b>a</b>) <span class="html-italic">L<sub>o</sub></span><sub>1</sub> equal to 196 µH; (<b>b</b>) <span class="html-italic">L<sub>o</sub></span><sub>2</sub> equal to 199 µH; (<b>c</b>) <span class="html-italic">L<sub>s</sub></span><sub>1</sub> equal to 70 µH.</p>
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<p>Values obtained through Design 8 tests: (<b>a</b>) <span class="html-italic">L<sub>o</sub></span><sub>1</sub> equal to 157 µH; (<b>b</b>) <span class="html-italic">L<sub>o</sub></span><sub>2</sub> equal to 158 µH; (<b>c</b>) <span class="html-italic">L<sub>s</sub></span><sub>1</sub> equal to 2.5 µH.</p>
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<p>HFTs development: (<b>a</b>) The first HFT had <span class="html-italic">L<sub>o</sub></span><sub>1</sub> equal to 173 µH, <span class="html-italic">L<sub>o</sub></span><sub>2</sub> equal to 172 µH, and <span class="html-italic">L<sub>s</sub></span><sub>1</sub> equal to 3 µH. (<b>b</b>) The second HFT had <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>L</mi> </mrow> <mrow> <mi>o</mi> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> equal to 147 µH, <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>L</mi> </mrow> <mrow> <mi>o</mi> <mn>2</mn> </mrow> </msub> </mrow> </semantics></math> equal to 148 µH, and <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>L</mi> </mrow> <mrow> <mi>s</mi> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> equal to 12.5 µH.</p>
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<p>Implemented prototype with the different PCBs and all the connections with the user.</p>
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<p>Experimental test using the first transformer with 1 kW of power and 100 kHz of <span class="html-italic">f<sub>sw</sub></span>. Yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal at the HFT input side (<span class="html-italic">V<sub>prim</sub></span>); red line: voltage signal at the HFT output side (<span class="html-italic">V<sub>sec</sub></span>); green line: current signal at the HFT input side (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output side (<span class="html-italic">I<sub>sec</sub></span>).</p>
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<p>Experimental test using the first transformer with 1 kW of power and 100 kHz of <span class="html-italic">f<sub>sw</sub></span>. Channel 1 yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal at the HFT input side (<span class="html-italic">V<sub>prim</sub></span>); red line: voltage signal at the HFT output side (<span class="html-italic">V<sub>sec</sub></span>); green line: current signal at the HFT input side (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output side (<span class="html-italic">I<sub>sec</sub></span>).</p>
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<p>Experimental test using the first transformer with 1 kW of power and 127 kHz of <span class="html-italic">f<sub>sw</sub></span>. Yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal at the HFT input side (<span class="html-italic">V<sub>prim</sub></span>); red line: voltage signal at the HFT output side (<span class="html-italic">V<sub>sec</sub></span>); green line: current signal at the HFT input side (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output side (<span class="html-italic">I<sub>sec</sub></span>).</p>
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<p>Experimental test using the first transformer with 1 kW of power and 127 kHz of <span class="html-italic">F<sub>sw</sub></span>. Yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal (<span class="html-italic">V<sub>ds</sub></span>); red line: anode-to-cathode <span class="html-italic">(V<sub>ak</sub></span>) voltage signal; green line: current signal at the HFT input (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output (<span class="html-italic">I<sub>sec</sub></span>).</p>
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<p>Experimental test using the second transformer with 1 kW of power and 50 kHz of <span class="html-italic">f<sub>sw</sub></span>. Yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal (<span class="html-italic">V<sub>ds</sub></span>); red line: anode-to-cathode <span class="html-italic">(V<sub>ak</sub></span>) voltage signal; green line: current signal at the HFT input (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output (<span class="html-italic">I<sub>sec</sub></span>).</p>
Full article ">Figure 32
<p>Experimental test using the second transformer with 1 kW of power and 50 kHz of <span class="html-italic">f<sub>sw</sub></span>. Yellow line: PWM signal at MOSFET 1 (<span class="html-italic">V<sub>_S</sub></span><sub>1</sub>); blue line: voltage signal (<span class="html-italic">V<sub>ds</sub></span>); red line: anode-to-cathode <span class="html-italic">(V<sub>ak</sub></span>) voltage signal; green line: current signal at the HFT input (<span class="html-italic">I<sub>prim</sub></span>); orange line: current signal at the HFT output (<span class="html-italic">I<sub>sec</sub></span>).</p>
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<p>Efficiency for an operating power range from 5 kW to 25 kW in the simulation environment.</p>
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19 pages, 1513 KiB  
Article
Factors Influencing Energy Drink Usage Amongst Pupils in the Mahikeng Sub-District, Northwest
by Karabo Dina Thini, Kebogile Elizabeth Mokwena and Mohora Feida Malebatja
Nutrients 2025, 17(5), 770; https://doi.org/10.3390/nu17050770 - 21 Feb 2025
Viewed by 473
Abstract
Background/Objectives: The high consumption rate of energy drinks among pupils is a serious public health concern in various countries, including South Africa. Excessive consumption of energy drinks that contain elevated caffeine and sugar levels has the potential to lead to the development of [...] Read more.
Background/Objectives: The high consumption rate of energy drinks among pupils is a serious public health concern in various countries, including South Africa. Excessive consumption of energy drinks that contain elevated caffeine and sugar levels has the potential to lead to the development of addictions, strokes, dehydration, sleeping disorders, mental health and central nervous disorders, hypertension, digestive problems, and anxiety. Most pupils regard energy drinks as regular soft drinks and lack knowledge of the active ingredients contained in energy drinks and their side effects. The objective of this study was to investigate factors influencing energy drink usage amongst pupils in the Mahikeng sub-district, Northwest Province. Methods: A quantitative cross-sectional survey was conducted amongst 505 pupils in the Mahikeng sub-district, Northwest, using self-administered questionnaires. Data were analysed using STATA software version 18 to examine associations between variables. Results: The energy drinks consumed most by pupils were Dragon (38.21%), Switch (28.97%), and Red Bull (14.62%). Factors and reasons influencing energy drink usage among pupils include all-night parties (3.1%), concentration (20.3%), being awake (43.1%), curiosity (2.2%), energy levels (23.1%), exams (13.8%), sports (8.7%), fatigue (6.9%), and health (2.3%). There was a strong association (p ≤ 0.05) between energy drink usage and sports activities amongst pupils. Conclusions: It is concluded that health education and promotion intervention programmes are required to educate pupils about the dangers of energy drink usage to prevent public health risks. Further studies, including research on primary school pupils, are necessary, considering that a substantial number of pupils were exposed to energy drinks at an early age. Full article
(This article belongs to the Section Nutrition and Public Health)
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<p>Map of the Mahikeng Sub-District.</p>
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<p>Multi-stage sampling technique applied in the study.</p>
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<p>Energy drinks consumed by learners.</p>
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<p>Reasons for choosing an energy drink brand.</p>
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17 pages, 6751 KiB  
Article
Study of Improved Active Clamp Phase-Shifted Full-Bridge Converter
by Xinyao Guo, Runquan Meng, Xiang Bai, Huajian Li, Jiahui Zhang and Xin He
Electronics 2025, 14(5), 834; https://doi.org/10.3390/electronics14050834 - 20 Feb 2025
Viewed by 120
Abstract
The polar energy router is a key device in the polar clean energy system which converges the output of wind power, photovoltaic units, energy storage units and hydrogen fuel cells through the power electronic power converter to the DC bus, which requires the [...] Read more.
The polar energy router is a key device in the polar clean energy system which converges the output of wind power, photovoltaic units, energy storage units and hydrogen fuel cells through the power electronic power converter to the DC bus, which requires the use of a variety of specifications of DC/DC converters; as a result, the efficiency of the DC/DC converter is directly connected to the efficiency of the polar energy router. This paper presents an enhanced isolated DC/DC converter with a phase-shifted full-bridge topology designed to meet the high-efficiency conversion requirements of polar energy routers. Although soft switching can be realized naturally in phase-shifted full-bridge topology, it also faces challenges, such as the difficulty of realizing soft switching under light load conditions, large circulation losses, a loss of duty cycle and oscillation in the secondary-side voltage. To solve these problems, an improved scheme of the phase-shifted full-bridge converter with an active clamp circuit is proposed in this paper. The scheme realized zero-voltage switch (ZVS) under light load by utilizing clamp capacitor energy. The on-state loss was reduced by zeroing the primary-side current during the circulating phase. This paper provides a detailed description of the topology, working principle and performance characteristics of the improved scheme, and its feasibility has been verified through experiments. Full article
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<p>Schematic diagram of active clamp circuit as current enhancement module.</p>
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<p>Schematic diagram of current enhancement principle.</p>
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<p>Schematic diagram of active clamp circuit as current enhancement module.</p>
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<p>Active clamp phase-shifted full-bridge converter with improved control.</p>
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<p>Timing diagram of primary waveforms of active clamp phase-shifted full-bridge converter.</p>
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<p>Mode 1 equivalent circuit.</p>
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<p>Mode 2 equivalent circuit.</p>
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<p>Mode 3 equivalent circuit.</p>
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<p>Mode 4 equivalent circuit.</p>
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<p>Mode 5 equivalent circuit.</p>
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<p>Mode 6 equivalent circuit.</p>
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<p>A schematic diagram of the free flow of the rectifier devices: (<b>a</b>) the output current flows through one rectifier and (<b>b</b>) the output current flows through two rectifiers.</p>
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<p>Experimental test platform.</p>
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<p>Physical diagram of converter.</p>
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<p>Primary switch gate PWM signal waveform.</p>
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<p>ZVS waveform of Q<sub>4</sub>.</p>
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<p>ZVS waveform of Q<sub>2</sub>.</p>
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<p>ZVS waveform of Q<sub>4</sub>.</p>
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<p>ZVS waveform of Q<sub>2</sub>.</p>
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<p>Synchronous rectifier drive waveform: (<b>a</b>) 100 A load current; (<b>b</b>) 170 A load current.</p>
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<p>Synchronous rectifier drive waveform: (<b>a</b>) 100 A load current; (<b>b</b>) 170 A load current.</p>
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<p>Efficiency curve.</p>
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15 pages, 5253 KiB  
Article
Fully Soft Switched Coupled Inductor-Based Semi Dual Active Half Bridge Converter with Voltage Match Control
by Liting Li, Mei Su, Wenjing Xiong, Yu Han and Guo Xu
Energies 2025, 18(4), 886; https://doi.org/10.3390/en18040886 - 13 Feb 2025
Viewed by 378
Abstract
This article proposes a fully soft-switched coupled-inductor-based semi dual-active half-bridge (SDAHB) converter designed for wide voltage range and low power applications. The converter employs a negatively coupled inductor to link a boost-type half-bridge with a semi half-bridge, utilizing a single magnetic component. This [...] Read more.
This article proposes a fully soft-switched coupled-inductor-based semi dual-active half-bridge (SDAHB) converter designed for wide voltage range and low power applications. The converter employs a negatively coupled inductor to link a boost-type half-bridge with a semi half-bridge, utilizing a single magnetic component. This approach addresses the issue of low power density often caused by multiple magnetic components, such as leakage inductor and transformer. Additionally, the proposed SDAHB converter includes only three active switches, making it cost-effective. To simplify control implementation, the converter operates under a voltage-matching condition using pulse width modulation plus phase shift (PPS) modulation. This also ensures full-range zero-voltage switching (ZVS) for all switches and zero-current switching (ZCS) for the diode. Finally, an experimental prototype is constructed, and the results confirm the validity of the theoretical analysis and design approach. Full article
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<p>The topology of the proposed SDAHB converter. (<b>a</b>) The topology with a coupled inductor. (<b>b</b>) The equivalent topology with Γ model.</p>
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<p>Theoretical operating waveforms of the proposed SDAHB converter.</p>
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<p>Equivalent circuits of the main operating intervals. (<b>a</b>) Stage 1 (<span class="html-italic">t</span><sub>0</sub>–<span class="html-italic">t</span><sub>1</sub>). (<b>b</b>) Stage 2 (<span class="html-italic">t</span><sub>1</sub>–<span class="html-italic">t</span><sub>2</sub>). (<b>c</b>) Stage 3 (<span class="html-italic">t</span><sub>2</sub>–<span class="html-italic">t</span><sub>3</sub>). (<b>d</b>) Stage 4 (<span class="html-italic">t</span><sub>3</sub>–<span class="html-italic">t</span><sub>4</sub>). (<b>e</b>) Stage 5 (<span class="html-italic">t</span><sub>4</sub>–<span class="html-italic">t</span><sub>5</sub>). (<b>f</b>) Stage 6 (<span class="html-italic">t</span><sub>5</sub>–<span class="html-italic">t</span><sub>6</sub>). (<b>g</b>) Stage 7 (<span class="html-italic">t</span><sub>6</sub>–<span class="html-italic">t</span><sub>7</sub>). (<b>h</b>) Stage 8 (<span class="html-italic">t</span><sub>7</sub>–<span class="html-italic">t</span><sub>8</sub>).</p>
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<p>Power curves under different <span class="html-italic">k</span>.</p>
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<p>RMS current curves under different <span class="html-italic">k</span>.</p>
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<p>The relationship between voltage conversion ratio and duty cycle and the comparison between the proposed method and the method in [<a href="#B21-energies-18-00886" class="html-bibr">21</a>].</p>
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<p>The control diagram of the proposed voltage match control method.</p>
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<p>Parameter design flowchart.</p>
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<p>Experimental prototype of the proposed SDAHB converter.</p>
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<p>Steady state waveforms of the semi DAHB converter. (<b>a</b>) <span class="html-italic">V<sub>in</sub></span> = 36 V, <span class="html-italic">P</span> = 100 W. (<b>b</b>) <span class="html-italic">V<sub>in</sub></span> = 72 V, <span class="html-italic">P</span> = 100 W. (<b>c</b>) <span class="html-italic">V<sub>in</sub></span> = 36 V, <span class="html-italic">P</span> = 20 W. (<b>d</b>) <span class="html-italic">V<sub>in</sub></span> = 72 V, <span class="html-italic">P</span> = 20 W.</p>
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<p>Soft switching waveforms when <span class="html-italic">V<sub>in</sub></span> = 36 V. (<b>a</b>) ZVS of <span class="html-italic">S</span><sub>1</sub> and ZCS of <span class="html-italic">D</span> at <span class="html-italic">P</span> = 100 W. (<b>b</b>) ZVS of <span class="html-italic">S</span><sub>2</sub> and <span class="html-italic">S</span><sub>3</sub> at <span class="html-italic">P</span> = 100 W. (<b>c</b>) ZVS of <span class="html-italic">S</span><sub>1</sub> and ZCS of <span class="html-italic">D</span> at <span class="html-italic">P</span> = 20 W. (<b>d</b>) ZVS of <span class="html-italic">S</span><sub>2</sub> and <span class="html-italic">S</span><sub>3</sub> at <span class="html-italic">P</span> = 20 W.</p>
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<p>Soft switching waveforms when <span class="html-italic">V<sub>in</sub></span> = 72 V. (<b>a</b>) ZVS of <span class="html-italic">S</span><sub>1</sub> and ZCS of <span class="html-italic">D</span> at <span class="html-italic">P</span> = 100 W. (<b>b</b>) ZVS of <span class="html-italic">S</span><sub>2</sub> and <span class="html-italic">S</span><sub>3</sub> at <span class="html-italic">P</span> = 100 W. (<b>c</b>) ZVS of <span class="html-italic">S</span><sub>1</sub> and ZCS of <span class="html-italic">D</span> at <span class="html-italic">P</span> = 20 W. (<b>d</b>) ZVS of <span class="html-italic">S</span><sub>2</sub> and <span class="html-italic">S</span><sub>3</sub> at <span class="html-italic">P</span> = 20 W.</p>
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<p>Load switching waveforms.</p>
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<p>Power loss breakdown under different input voltages and loads.</p>
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<p>Experimental efficiency curves.</p>
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17 pages, 3226 KiB  
Article
Single-Level and Two-Level Circuit Solutions for Buck-Boost AC Voltage Regulators with Phase-by-Phase Switches
by Aleksey Udovichenko, Evgeniy Grishanov, Evgeniy Kosykh, Maksim Filippov and Maksim Dybko
Electricity 2025, 6(1), 6; https://doi.org/10.3390/electricity6010006 - 12 Feb 2025
Viewed by 328
Abstract
Forming required AC voltage levels is currently one of the most pressing problems. Unstable voltage levels can lead to the failure of household and industrial equipment. This can lead to a pure effect on the production cycle. In this regard, the development of [...] Read more.
Forming required AC voltage levels is currently one of the most pressing problems. Unstable voltage levels can lead to the failure of household and industrial equipment. This can lead to a pure effect on the production cycle. In this regard, the development of AC voltage regulators has become relevant. Such regulators can perform the function of voltage level asymmetry compensators in a three-phase power supply network. In turn, new topologies should be energy-efficient and reliable. This can be achieved by reducing the number of semiconductor elements, thus reducing losses and increasing efficiency. Also, AC voltage regulators have found applications as soft-start devices for motors and have become relevant to frequency converters. The power level of such devices can vary from units to tens of kilowatts. This paper presents several circuit design solutions for AC voltage regulators with fewer switches. These solutions are made according to both a single-level and two-level system, where the level refers to the number of links that increase the transmission coefficient. The schemes were analyzed, and efficiency was evaluated through their harmonic coefficients, power factor, and efficiency coefficient. For the basic scheme, a photo of the experimental layout and its results are provided. Full article
(This article belongs to the Special Issue Recent Advances in Power and Smart Grids)
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<p>A family of new energy-efficient buck-boost RAV with phase-by-phase switches (<b>a</b>)—basic LCC circuit, (<b>b</b>)—CC circuit, (<b>c</b>)—C circuit, (<b>d</b>)—two-level LCC variant, (<b>e</b>)—AC switch.</p>
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<p>A family of new energy-efficient buck-boost RAV with phase-by-phase switches (<b>a</b>)—basic LCC circuit, (<b>b</b>)—CC circuit, (<b>c</b>)—C circuit, (<b>d</b>)—two-level LCC variant, (<b>e</b>)—AC switch.</p>
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<p>An example of thermal models setting up the GD200CEY120C2S transistor.</p>
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<p>Output voltage diagrams (green) for LC RAV combined with input voltage (blue) and load current (red).</p>
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<p>A family of control characteristics of buck-boost RAV with phase-by-phase switches.</p>
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<p>A family of load characteristics of buck-boost RAV with phase-by-phase switches.</p>
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<p>The dependence of the input shift factor on the modulation depth.</p>
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<p>Dependence of the input current THD of buck-boost RAV with phase-by-phase switches.</p>
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<p>Dependence of the output voltage THD of buck-boost RAV with phase-by-phase switches.</p>
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<p>Experimental layout of the RAV (<b>a</b>), (<b>b</b>) (red—choke block, blue—capacitor block, white—ADUM4135BRWZ-based drivers, yellow—GD200CEY120C2S IGBT power modules, brown—LA55-P current sensors and LV25-P voltage sensors), (<b>c</b>) control system based on STM32F407ZGT6 and (<b>d</b>) photo of the experimental layout.</p>
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<p>Experimental layout of the RAV (<b>a</b>), (<b>b</b>) (red—choke block, blue—capacitor block, white—ADUM4135BRWZ-based drivers, yellow—GD200CEY120C2S IGBT power modules, brown—LA55-P current sensors and LV25-P voltage sensors), (<b>c</b>) control system based on STM32F407ZGT6 and (<b>d</b>) photo of the experimental layout.</p>
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<p>The current (red) and voltage (blue) of the old experimental model of RAV (<b>a</b>,<b>b</b>) with different LC proportions).</p>
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14 pages, 3350 KiB  
Article
Optimization Study of Rare Earth-Free Metal Amorphous Nanocomposite Axial Flux-Switching Permanent Magnet Motor
by Kyle P. Schneider, Satoru Simizu, Michael E. McHenry and Maarten P. de Boer
Energies 2025, 18(3), 640; https://doi.org/10.3390/en18030640 - 30 Jan 2025
Viewed by 543
Abstract
Metal amorphous nanocomposite (MANC) soft magnetic materials exhibit remarkably low iron loss and high saturation magnetization. However, they have not been widely used in electric motors largely due to a lack of demonstrated manufacturing processing methods and an absence of proven motor designs [...] Read more.
Metal amorphous nanocomposite (MANC) soft magnetic materials exhibit remarkably low iron loss and high saturation magnetization. However, they have not been widely used in electric motors largely due to a lack of demonstrated manufacturing processing methods and an absence of proven motor designs well suited for their use. Recent developments in these two areas have prompted the optimization study of flux-switching with permanent magnet motor topology using MANCs presented here. This study uses population-based optimization in conjunction with a simplified electromagnetics model to seek rare earth-free designs that attain or exceed the state of the art in power density and efficiency. To predict the maximum mechanically safe rotational speed for each design with minimal computational effort, a new method of quantifying the rotor assembly mechanical limit is presented. The resulting population of designs includes motor designs with a specific power of up to 6.1 kW/kg and efficiency of up to 99% without the use of rare earth permanent magnets. These designs, while exhibiting drawbacks of high electrical frequency and significant manufacturing complexity, exceed the typical power density of representative state-of-the-art EV motors while increasing efficiency and eliminating rare earth elements. Full article
(This article belongs to the Special Issue Advances in Permanent Magnet Motor and Motor Control)
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<p>(<b>a</b>) Anhysteretic magnetization curves and (<b>b</b>) core loss vs. frequency at 1.0 T of materials relevant to axial flux motors, Somaloy 700HR 5P [<a href="#B10-energies-18-00640" class="html-bibr">10</a>], M235-35A [<a href="#B11-energies-18-00640" class="html-bibr">11</a>] (losses extrapolated), and Fe-Ni80-based MANC of interest here (process-dependent, BH curve approximated from ref. [<a href="#B4-energies-18-00640" class="html-bibr">4</a>], loss plot generated from parameters in ref. [<a href="#B9-energies-18-00640" class="html-bibr">9</a>]).</p>
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<p>(<b>a</b>) Three-dimensional geometry of axial FSWPM motor with coils removed for clarity and PMs shown in blue. (<b>b</b>) Two-dimensional plot of flux density x component for initial FSWPM motor design and (<b>c</b>) y component of flux density for revised “direct flux” design that eliminates the need for rotor back iron. See <a href="#sec2dot1-energies-18-00640" class="html-sec">Section 2.1</a> for more information.</p>
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<p>(<b>a</b>) Rotor wedge, hub, and carbon fiber sleeve geometry, shown with sector symmetry and transparent hub parts. (<b>b</b>) Stress heat map of one hub part when viewed from inside the hub, where blue is the lowest and red the highest stress.</p>
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<p>(<b>a</b>) Pareto-optimal front of direct flux motor optimization after 200 individuals across 100 generations. (<b>b</b>) Dependence of motor electromagnetic efficiency (top) and specific power (bottom) on each free parameter in <a href="#energies-18-00640-t003" class="html-table">Table 3</a>. Here, the vertical axis gives the normalized variable value, and left to right (blue to green to red) in each parameter demonstrates increasing efficiency and specific power.</p>
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17 pages, 23729 KiB  
Article
Topology and Control Strategy of Multi-Port DC Power Electronic Transformer Based on Soft Switching
by Jialin Zhang, Kunpeng Zha, Xiaojun Tang, Yuefeng Yang, Lanfang Li and Jiafei Li
Energies 2025, 18(2), 400; https://doi.org/10.3390/en18020400 - 17 Jan 2025
Viewed by 352
Abstract
Multi-port DC power electronic transformer (PET) is a core equipment for achieving transformation of different voltage levels and flexible interconnection of different DC buses in a DC distribution system. It is capable of bidirectional energy flow, flexible regulation of power flow, port fault [...] Read more.
Multi-port DC power electronic transformer (PET) is a core equipment for achieving transformation of different voltage levels and flexible interconnection of different DC buses in a DC distribution system. It is capable of bidirectional energy flow, flexible regulation of power flow, port fault isolation, and other functions. A new five-port DC transformer topology based on soft switching technology is proposed in this paper. In this topology, different DC voltage levels can be interconnected efficiently, such as 20 kV, 750 V, ±375 V, and 300 to 500 V adjustable. The control of each port is simple and flexible. The output voltage is stable, and they are independent of each other, which can improve the system reliability. The topology of the proposed multi-port DC transformer is introduced in detail. The working principle, control strategy, and parameter design method of the transformer are analyzed. Simulations and experimental results are provided to validate the theoretical analysis. Full article
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<p>Topology of the proposed five-port DC transformer.</p>
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<p>Topology of the proposed IBC.</p>
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<p>The working waveform of the IBC.</p>
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<p>Working modes of IBC: (<b>a</b>) Mode 1; (<b>b</b>) Mode 2; (<b>c</b>) Mode 3; (<b>d</b>) Mode 4; (<b>e</b>) Mode 5; (<b>f</b>) Mode 6. (The direction of the arrow indicates the direction of voltage and current).</p>
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<p>Topology of the TC-VBU.</p>
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<p>Working modes of TC-VBU: (<b>a</b>) Mode 1; (<b>b</b>) Mode 2. (The direction of the arrow indicates the direction of current).</p>
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<p>The working waveform of TC-VBU under power-unbalanced condition.</p>
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<p>Experimental circuit based on power circulation. (The dashed arrows represent voltage and current sampling).</p>
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<p>The picture of the experimental platform. (<b>a</b>) Main circuit; (<b>b</b>) Control circuit.</p>
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<p>The control diagram: (<b>a</b>) closed-loop control of IBC; (<b>b</b>) open-loop control of TC-VBU.</p>
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<p>The experimental waveform of IBC output voltage, output current, and inductor (<span class="html-italic">L</span>) current: (<b>a</b>) forward mode; (<b>b</b>) backward mode. (The triangles represent the coordinate origin).</p>
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<p>The experimental waveform of switch voltage and auxiliary inductor current of IBC: (<b>a</b>) forward mode; (<b>b</b>) backward mode. (The triangles represent the coordinate origin).</p>
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<p>The experimental waveform of switch voltage and resonant current of TC-VBU: (<b>a</b>) forward mode; (<b>b</b>) backward mode. (The triangles represent the coordinate origin).</p>
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<p>The transient performance of experimental waveform (forward mode): (<b>a</b>) IBC output voltage, output current, and inductor (<span class="html-italic">L</span>) current; (<b>b</b>) switch voltage and auxiliary inductor current of IBC; (<b>c</b>) switch voltage and resonant current of TC-VBU. (The triangles represent the coordinate origin).</p>
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<p>The transient performance of experimental waveform (backward mode): (<b>a</b>) IBC output voltage, output current, and inductor (<span class="html-italic">L</span>) current; (<b>b</b>) switch voltage and auxiliary inductor current of IBC; (<b>c</b>) switch voltage and resonant current of TC-VBU. (The triangles represent the coordinate origin).</p>
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<p>The efficiency curves of IBC and TC-VBU under different power points.</p>
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16 pages, 1856 KiB  
Article
Design of Resonant Inverters with Energy Dosing, Based on Optimization with Reference Curve
by Nikolay Madzharov, Bogdan Gilev and Nikolay Hinov
Electronics 2025, 14(2), 327; https://doi.org/10.3390/electronics14020327 - 15 Jan 2025
Viewed by 387
Abstract
This paper presents an optimization-based approach for the design of energy-dosing resonant inverters (RI) using a reference curve. RI are widely used in areas such as wireless charging, induction heating, and high-frequency power supplies due to their high efficiency and soft-switching capability. The [...] Read more.
This paper presents an optimization-based approach for the design of energy-dosing resonant inverters (RI) using a reference curve. RI are widely used in areas such as wireless charging, induction heating, and high-frequency power supplies due to their high efficiency and soft-switching capability. The proposed methodology uses a reference current curve in the alternating current (AC) circuit that describes the ideal behavior of the inverter during the start-up transient in order to operate in energy metering mode and minimize switching losses. This approach analyzes and optimizes the values of the circuit elements whose initial values are determined by applying a rational design methodology. The results of the study demonstrate increased dynamics of the power circuit and better stability compared to applying traditional design methods. The presented simulation results confirm the effectiveness of the proposed optimization and the improvement of the characteristics of the studied device. Full article
(This article belongs to the Special Issue Power Electronics and Its Applications in Power System)
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<p>Some types of resonant inverters with energy dosing (<b>a</b>) half-bridge; (<b>b</b>) with split resonant capacitor; and (<b>c</b>) full-bridge.</p>
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<p>Some types of resonant inverters with energy dosing (<b>a</b>) half-bridge; (<b>b</b>) with split resonant capacitor; and (<b>c</b>) full-bridge.</p>
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<p>Timing diagrams illustrating a fixed operating mode of an inverter with a split resonant capacitor and energy dosing.</p>
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<p>Matlab/Simulink model of an inverter with split resonant capacitor and energy dosing.</p>
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<p>Visualization of the state variables in the studied scheme, obtained through the created model.</p>
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<p>Current <span class="html-italic">i<sub>LR</sub></span> (blue), envelope <span class="html-italic">i<sub>LR</sub></span> (red), and reference envelope <span class="html-italic">i<sub>LR</sub></span> (green).</p>
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<p>Result of the optimization program.</p>
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<p>Current in the AC circuit of the inverter <span class="html-italic">i<sub>LR</sub></span> (blue), envelope of this current (red), and a given reference envelope of the current (green), obtained after optimization.</p>
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13 pages, 3902 KiB  
Article
A Study on the Effect of Plastic Strain on Magnetic Phenomenology and Microstructure
by Mehrija Hasičić, Spyridon Angelopoulos, Aphrodite Ktena and Evangelos Hristoforou
Magnetism 2025, 5(1), 1; https://doi.org/10.3390/magnetism5010001 - 14 Jan 2025
Viewed by 332
Abstract
The present work aspires to contribute to the discussion on the relationship between macroscopic measurements and microstructure, helping establish a methodology that will allow the quantitative assessment of the effect of strain on magnetic properties in the plastic deformation regime. In particular, we [...] Read more.
The present work aspires to contribute to the discussion on the relationship between macroscopic measurements and microstructure, helping establish a methodology that will allow the quantitative assessment of the effect of strain on magnetic properties in the plastic deformation regime. In particular, we study the effect of strain on the magnetization process as a result of varying the anisotropy profile at the grain level. Results on micromagnetic calculations of hysteresis loops for various configurations of magnetic anisotropy are shown and discussed against the interplay between the energy terms involved in the calculations, namely anisotropy, demagnetizing, and exchange. The results are in line with previously obtained results using vector Preisach modeling with the Stoner–Wohlfarth model acting both as a switching and rotation mechanism. The hysteresis loop phenomenology is consistent with the emergence of a hard phase in the form of a boundary around soft grains which is assumed to be the result of the onset of compressive stresses in the plastic region. Future research will be oriented toward the study of the effect of the secondary peak in differential permeability, which is observed experimentally in the plastic deformation region, and its dependence on the angle of misalignment between the hard boundary and the soft grain. Full article
(This article belongs to the Special Issue Mathematical Modelling and Physical Applications of Magnetic Systems)
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<p>(<b>a</b>) Stress–strain curves measured at different rates of applied tensile stress and (<b>b</b>) corresponding magnetic Barkhausen noise counts.</p>
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<p>Hysteresis loops measured at various strain levels.</p>
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<p>Geometry of a sample modeled.</p>
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<p>Magnetization plots for two types of simulated samples: (<b>a</b>) soft grain with hard magnetic boundary and (<b>b</b>) four identical soft grains surrounded by misaligned hard magnetic boundaries. The total volume of hard magnetic material is the same in both samples. For clarity, one arrow represents the magnetization of 19 neighboring cells.</p>
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<p>Major hysteresis loop obtained for the soft homogeneous case #1 (black), case #6 with one soft grain enclosed by a hard grain boundary of <span class="html-italic">K</span><sub>1</sub> = 1000 kJ/m<sup>3</sup> and 100 nm thickness (green), case #7 with one soft grain enclosed by a hard grain boundary with <span class="html-italic">K</span><sub>1</sub> = 1000 kJ/m<sup>3</sup> and 300 nm thickness (red) and case #8 with soft grain enclosed by a hard grain boundary with <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 500 kJ/m<sup>3</sup> and 300 nm thickness (blue).</p>
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<p>Energy terms of the (<b>a</b>) homogeneous case #1, (<b>b</b>) case #6 with hard grain boundary of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> and 100 nm thickness, (<b>c</b>) case #7 with hard grain boundary of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> and 300 nm thickness and (<b>d</b>) case #8 with hard grain boundary of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 500 kJ/m<sup>3</sup> and 300 nm thickness.</p>
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<p>Major hysteresis loop for the case #7 of one hard grain boundary with <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> in <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>y</mi> </mrow> </semantics></math>-direction ±45 degrees and 300 nm thickness (blue) compared against the case #9 of a hard grain boundary with <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> and 300 nm thickness where the top and bottom side are pinned along the y-direction while the left and right side are kept in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>y</mi> </mrow> </semantics></math>-drection ±45 degrees (red).</p>
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<p>Energy terms for the loops shown in <a href="#magnetism-05-00001-f007" class="html-fig">Figure 7</a>: (<b>a</b>) hard grain boundary of 300 nm thickness with <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> in the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>y</mi> </mrow> </semantics></math>-direction ±45 degrees and (<b>b</b>) hard grain boundary of 300 nm thickness with <math display="inline"><semantics> <mrow> <msub> <mrow> <mi>K</mi> </mrow> <mrow> <mn>1</mn> </mrow> </msub> </mrow> </semantics></math> = 1000 kJ/m<sup>3</sup> where the top and bottom sides are oriented along the y-direction while the left and right sides are kept along the <math display="inline"><semantics> <mrow> <mi>x</mi> <mi>y</mi> </mrow> </semantics></math>-drection ±45 degrees).</p>
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<p>Major hysteresis loop obtained for case #7 with one soft grain with hard grain boundary (blue) and case #2 with four soft grains with hard grain boundary (red).</p>
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<p>Energy terms for (<b>a</b>) case #7 with one soft grain with hard grain boundary and (<b>b</b>) case #2 with four soft grains with hard grain boundary.</p>
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17 pages, 8323 KiB  
Article
A Symmetrical Leech-Inspired Soft Crawling Robot Based on Gesture Control
by Jiabiao Li, Ruiheng Liu, Tianyu Zhang and Jianbin Liu
Biomimetics 2025, 10(1), 35; https://doi.org/10.3390/biomimetics10010035 - 8 Jan 2025
Viewed by 511
Abstract
This paper presents a novel soft crawling robot controlled by gesture recognition, aimed at enhancing the operability and adaptability of soft robots through natural human–computer interactions. The Leap Motion sensor is employed to capture hand gesture data, and Unreal Engine is used for [...] Read more.
This paper presents a novel soft crawling robot controlled by gesture recognition, aimed at enhancing the operability and adaptability of soft robots through natural human–computer interactions. The Leap Motion sensor is employed to capture hand gesture data, and Unreal Engine is used for gesture recognition. Using the UE4Duino, gesture semantics are transmitted to an Arduino control system, enabling direct control over the robot’s movements. For accurate and real-time gesture recognition, we propose a threshold-based method for static gestures and a backpropagation (BP) neural network model for dynamic gestures. In terms of design, the robot utilizes cost-effective thermoplastic polyurethane (TPU) film as the primary pneumatic actuator material. Through a positive and negative pressure switching circuit, the robot’s actuators achieve controllable extension and contraction, allowing for basic movements such as linear motion and directional changes. Experimental results demonstrate that the robot can successfully perform diverse motions under gesture control, highlighting the potential of gesture-based interaction in soft robotics. Full article
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<p>Angle schematic diagram.</p>
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<p>Overall system schematic diagram.</p>
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<p>Dynamic gesture features related to palms.</p>
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<p>BP neural network diagram.</p>
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<p>Dynamic gesture recognition process.</p>
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<p>The change diagram of the cost function.</p>
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<p>Structure diagram of TPU soft crawling robot.</p>
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<p>Schematic diagram of positive and negative voltage switching circuits.</p>
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<p>System control.</p>
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<p>Schematic diagrams of positive and negative voltage switching circuits. (<b>a</b>) The curved actuators in the rear are bent. (<b>b</b>) The inflatable actuators of the left and right expandable actuators are inflated and expanded. (<b>c</b>) The inflatable actuators of the front are inflated and bent. (<b>d</b>) The curved actuators in the rear are deflated. (<b>e</b>) The actuators of the left and right expandable are deflated.</p>
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<p>Model of the soft robot.</p>
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<p>The linear motion of the TPU soft crawling robot (the red dotted line represents the initial position). (<b>a</b>) The curved actuators in the rear are bent. (<b>b</b>) The inflatable actuators of the left and right expandable actuators are inflated and expanded. (<b>c</b>) The inflatable actuators of the front are inflated and bent. (<b>d</b>) The curved actuators in the rear are detached from the ground. (<b>e</b>) The actuators of the left and right expandable are deflated under negative pressure. (<b>f</b>) The curved actuators of the front are detached from the ground.</p>
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<p>The TPU soft crawling robot changes direction during movement (the red dotted line represents the state of the two plates during the robot’s movement). (<b>a</b>) The curved actuators in the rear are bent. (<b>b</b>) The inflatable actuators of the left expandable actuators are inflated and expanded. (<b>c</b>) The inflatable actuators of the front are inflated and bent. (<b>d</b>) The curved actuators in the rear are detached from the ground. (<b>e</b>) The actuators of the left expandable are deflated under negative pressure. (<b>f</b>) The curved actuators of the front are detached from the ground.</p>
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26 pages, 16984 KiB  
Article
An Enhanced Solar Battery Charger Using a DC-DC Single-Ended Primary-Inductor Converter and Fuzzy Logic-Based Control for Off-Grid Photovoltaic Applications
by Julio López Seguel, Samuel Zenteno, Crystopher Arancibia, José Rodríguez, Mokhtar Aly, Seleme I. Seleme and Lenin M. F. Morais
Processes 2025, 13(1), 99; https://doi.org/10.3390/pr13010099 - 3 Jan 2025
Viewed by 668
Abstract
Battery charging systems are crucial for energy storage in off-grid photovoltaic (PV) installations. Since the power generated by a PV panel is conditioned by climatic conditions and load characteristics, a maximum power point tracking (MPPT) technique is required to maximize PV power and [...] Read more.
Battery charging systems are crucial for energy storage in off-grid photovoltaic (PV) installations. Since the power generated by a PV panel is conditioned by climatic conditions and load characteristics, a maximum power point tracking (MPPT) technique is required to maximize PV power and accelerate battery charging. On the other hand, a battery must be carefully charged, ensuring that its charging current and voltage limits are not exceeded, thereby preventing premature degradation. However, the voltage generated by the PV panel during MPPT operation fluctuates, which can harm the battery, particularly during periods of intense radiation when overvoltages are likely to occur. To address these issues, the design and construction of an enhanced solar battery charger utilizing a single-ended primary-inductor converter (SEPIC) and soft computing (SC)-based control is presented. A control strategy is employed that integrates voltage stabilization and MPPT functions through two dedicated fuzzy logic controllers (FLCs), which manage battery charging using a three-mode scheme: MPPT, Absorption, and Float. This approach optimizes available PV power while guaranteeing fast and safe battery charging. The developed charger leverages the SEPIC’s notable features for PV applications, including a wide input voltage range, minimal input current ripple, and an easy-to-drive switch. Moreover, unlike most PV charger control strategies in the literature that combine improved traditional MPPT methods with classical proportional integral (PI)-based control loops, the proposed control adopts a fully SC-based strategy, effectively addressing common drawbacks of conventional methods, such as slowness and inaccuracy during sudden atmospheric fluctuations. Simulations in MATLAB/Simulink compared the FLCs’ performance with conventional methods (P&O, IncCond, and PID). Additionally, a low-power hardware prototype using an Arduino Due microcontroller was built to evaluate the battery charger’s behavior under real weather conditions. The simulated and experimental results both demonstrate the robustness and effectiveness of the solar charger. Full article
(This article belongs to the Special Issue Advances in Renewable Energy Systems (2nd Edition))
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<p>Layout of the proposed solar charger.</p>
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<p>Equivalent electrical model of an ideal PV cell.</p>
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<p>P-V characteristics of WANT-M55W solar module at a constant environmental temperature of 25 °C, a NOCT of 47 °C, and varying irradiance levels.</p>
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<p>SEPIC circuit.</p>
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<p>Battery charging phases.</p>
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<p>Flow diagram of the power management strategy.</p>
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<p>Block diagram of an FLC controller.</p>
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<p>Representation of power variations against voltage variations.</p>
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<p>Parameters of the proposed FL-MPPT algorithm: (<b>a</b>) S; (<b>b</b>) ΔP<sub>PV</sub>; (<b>c</b>) ΔD; (<b>d</b>) control surface.</p>
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<p>Parameters of the designed FL-VC: (<b>a</b>) E; (<b>b</b>) ΔE; (<b>c</b>) ΔD; (<b>d</b>) control surface.</p>
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<p>Simulation scheme implemented.</p>
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<p>Behavior of the MPPT methods at a fixed air temperature of 25 °C with various irradiance levels and a load of R = 15 Ω.</p>
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<p>Behavior of the MPPT methods at a fixed air temperature of 25 °C with various irradiance levels and a load of R = 30 Ω.</p>
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<p>Behavior of the MPPT methods for a fixed air temperature of 25 °C and various irradiance levels and a load of R = 60 Ω.</p>
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<p>Comparative performance analysis of the voltage controllers.</p>
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<p>The practical hardware setup.</p>
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<p>MOSFET drain-source signal.</p>
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<p>FL-MPPT transient behavior in response to load disturbance.</p>
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<p>FL-MPPT efficiency at distinct irradiance intensities: (<b>a</b>) low intensity; (<b>b</b>) medium intensity; (<b>c</b>) high intensity.</p>
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<p>Reference change test for the FL-VC.</p>
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<p>Load disturbance test for the FL-VC.</p>
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<p>Experimental results for the three-mode charging strategy.</p>
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15 pages, 5913 KiB  
Article
Research on Self-Excited Inverter Rectification Method of Receiver in Wireless Power Transfer System
by Suqi Liu, Xueying Yan, Gang Wang and Yuping Liu
Processes 2025, 13(1), 89; https://doi.org/10.3390/pr13010089 - 2 Jan 2025
Viewed by 508
Abstract
To decrease the complexity and increase the efficiency of wireless power transfer (WPT) systems, this paper proposes a novel self-excited invert rectification method for the design of the invert rectifier of the receiver (Rx). The self-excited invert rectifier can perform the self-driving and [...] Read more.
To decrease the complexity and increase the efficiency of wireless power transfer (WPT) systems, this paper proposes a novel self-excited invert rectification method for the design of the invert rectifier of the receiver (Rx). The self-excited invert rectifier can perform the self-driving and soft-switching of the MOSFETs as well as the frequency-tracking function without a microcontroller. This allows us to greatly simplify the structure of the invert rectifier and increase the transfer efficiency (TE) of the WPT system. Firstly, a self-excited invert rectifier circuit is designed, and a self-excited invert rectification method is studied. Additionally, the power loss of the self-excited invert rectifier is analyzed. Finally, the self-excited invert rectifier of the WPT experimental system is designed. The self-excited invert rectification method is then verified. The key component parameters of the self-excited invert rectifier are provided and optimized. The TE of the WPT system that includes the self-excited invert rectifier is improved by more than 5% without a microcontroller. The self-excited invert rectifier of the Rx provides a practical solution for decreasing the complexity and increasing the TE of the WPT system. Full article
(This article belongs to the Section Energy Systems)
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<p>WPT system with self-excited invert rectifier.</p>
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<p>Working mode 1 of the self-excited invert rectifier: Current flow is power and control loops; and 0 V is the same voltage level.</p>
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<p>Working mode 2 of the self-excited invert rectifier: Current flow of red dotted lines is the first process section; and Current flow of green dotted lines is the second process section.</p>
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<p>Working mode 3 of the self-excited invert rectifier: Current flow is power and control loops; and 0 V is the same voltage level.</p>
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<p>Working mode 4 of the self-excited invert rectifier: Current flow of red dotted lines is the first process section; and Current flow of green dotted lines is the second process section.</p>
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<p>Steady−state working waveform of the switching tubes in the self-excited invert rectifier.</p>
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<p>Equivalent circuit of the WPT system with self-excited invert rectifier.</p>
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<p>Block diagram of the WPT system that includes a self-excited invert rectifier.</p>
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<p>A self-excited invert rectifier experimental equipment able to perform the SD and SS of the MOSFETs.</p>
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<p>Gate drive waveform (gate−source voltage) of the <span class="html-italic">Q</span><sub>5</sub> and <span class="html-italic">Q</span><sub>6</sub> switching tubes.</p>
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<p>Working waveform (drain−source voltage) of the <span class="html-italic">Q</span><sub>5</sub> and <span class="html-italic">Q</span><sub>6</sub> switching tubes.</p>
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<p>(<b>a</b>) Circuit of the Rx and (<b>b</b>) temperature image of the self-excited invert rectifier.</p>
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<p>OP of the system as a function of the distance.</p>
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<p>TE of the system as a function of the distance.</p>
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<p>OP <span class="html-italic">P</span><sub>2</sub> of the system as a function of the frequency.</p>
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<p>TE <span class="html-italic">η</span><sub>2</sub> of the system as a function of the frequency.</p>
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