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Article

Meta Surface-Based Multiband MIMO Antenna for UAV Communications at mm-Wave and Sub-THz Bands

by
Tale Saeidi
1,2,
Sahar Saleh
1,3,
Nick Timmons
1,
Ahmed Jamal Abdullah Al-Gburi
4,
Saeid Karamzadeh
5,6,
Ayman A. Althuwayb
7,*,
Nasr Rashid
7,
Khaled Kaaniche
7,
Ahmed Ben Atitallah
7 and
Osama I. Elhamrawy
7
1
WiSAR Lab, Atlantic Technological University (ATU), F92 FC93 Letterkenny, Ireland
2
Electrical and Electronics Engineering Department, Faculty of Engineering and Natural Sciences, İstinye University, 34396 Istanbul, Turkey
3
Department of Electronics and Communications Engineering, Faculty of Engineering, Aden University, Aden 5243, Yemen
4
Centre of Telecommunication Research and Innovation (CeTRI), Faculty of Electronics and Computer Technology and Engineering, Universiti Teknikal Malaysia Melaka, Durian Tunggal 76100, Malaysia
5
Millimeter Wave Technologies, Intelligent Wireless System, Silicon Austria Labs (SAL), 4040 Linz, Austria
6
Electrical and Electronics Engineering Department, Faculty of Engineering and Natural Sciences, Bahçeşehir University, 34349 Istanbul, Turkey
7
Electrical Engineering Department, College of Engineering, Jouf University, Sakaka 72388, Saudi Arabia
*
Author to whom correspondence should be addressed.
Drones 2024, 8(8), 403; https://doi.org/10.3390/drones8080403
Submission received: 9 June 2024 / Revised: 4 August 2024 / Accepted: 13 August 2024 / Published: 16 August 2024
Browse Figures
Figure 1
<p>Asymmetry radiation principle: (<b>a</b>) TE<sub>10</sub> mode in even structure; (<b>b</b>) leaky TE<sub>10</sub> mode in the uneven structure; (<b>c</b>) leaky TE<sub>01</sub> mode in even structure (θ is the approximated maximum angle of the radiation and the red arrows show the direction of the field) [<a href="#B36-drones-08-00403" class="html-bibr">36</a>].</p> "> Figure 2
<p>The design configuration of the single antenna and MIMO, including each step of the design and the proposed prototype ((<b>a</b>) step 1, (<b>b</b>) step 2, (<b>c</b>) step 3, (<b>d</b>) proposed, (<b>e</b>) front view of the MIMO, and (<b>f</b>) ground view of the MIMO).</p> "> Figure 3
<p>Coupling mechanisms of coplanar proximity feed ((<b>a</b>) inductive coupling and (<b>b</b>) capacitive coupling).</p> "> Figure 4
<p>The surface current mechanism of the proposed antenna (the blue and red lines are the current and voltage standing wave amplitude, respectively).</p> "> Figure 5
<p>The surface current distribution of the initial design of the antenna at (<b>a</b>) 8.5 GHz, (<b>b</b>) 25 GHz, and (<b>c</b>) 120 GHz.</p> "> Figure 6
<p>The surface current distribution at (<b>a</b>) 8.5 GHz, (<b>b</b>) 25 GHz, and (<b>c</b>) 120 GHz.</p> "> Figure 7
<p>Parametric study of the single element antenna: reflection coefficient results of (<b>a</b>) <span class="html-italic">L<sub>f</sub></span>, (<b>b</b>) <span class="html-italic">L<sub>p</sub></span>, (<b>c</b>) <span class="html-italic">L<sub>g</sub></span>, <span class="html-italic">W<sub>g</sub></span>, and (<b>d</b>) <span class="html-italic">W<sub>p</sub></span>.</p> "> Figure 8
<p>Reflection coefficient results of the proposed single-port antenna for each stage.</p> "> Figure 9
<p>S-parameters results of the MIMO antenna without the neutralization tape and metasurface elements.</p> "> Figure 10
<p>A perspective view of the MIMO antenna showcasing each layer and their materials.</p> "> Figure 11
<p>The electric field of the proposed two ports MIMO antenna at different frequencies (<b>a</b>) 8.5 GHz, (<b>b</b>) 25 GHz, (<b>c</b>) 75 GHz, (<b>d</b>) 115 GHz, and (<b>e</b>) 120 GHz.</p> "> Figure 12
<p>The single-element coffee bean metasurface’s characteristics: (<b>a</b>) S-parameters results; (<b>b</b>,<b>c</b>) permittivity and permeability results; (<b>d</b>) single element of the coffee bean.</p> "> Figure 13
<p>Simulation setup of the proposed antenna: (<b>a</b>) beneath view, (<b>b</b>) upper view, (<b>c</b>) two arrays of the proposed antenna on one drone, (<b>d</b>) two drones and antenna systems with vertical space, and (<b>e</b>) two drones and antenna systems with horizontal space.</p> "> Figure 14
<p>The measurement setup of the proposed antenna in the air at microwave (up to 10 GHz) and mm-wave (up to 30 GHz) bands: (<b>a</b>) without a drone and (<b>b</b>) with a drone.</p> "> Figure 15
<p>The measurement and calibration setup of the antenna in the air at the sub-THz band (77–140 GHz).</p> "> Figure 16
<p>Simulated and measured S-parameter results of the proposed antenna after integrating with metasurface in air and attached to a drone: (<b>a</b>) transmission coefficient and (<b>b</b>) reflection coefficient.</p> "> Figure 17
<p>Reflection and transmission coefficient results when two arrays of the antenna are attached to one drone.</p> "> Figure 18
<p>Transmission coefficients results of the antenna on two drones with different (<b>a</b>) vertical and (<b>b</b>) horizontal distances.</p> "> Figure 19
<p>Radiation pattern measurement of the antenna in the mm-wave chamber.</p> "> Figure 20
<p>The simulated and measured electric and magnetic field of the antenna in free space at (<b>a</b>) 8.5 GHz, (<b>b</b>) 18 GHz, (<b>c</b>) 25 GHz, (<b>d</b>) 66 GHz, and (<b>e</b>) 77 GHz.</p> "> Figure 21
<p>Beam steering of the proposed antenna at (<b>a</b>) 8.5 GHz, (<b>b</b>) 10 GHz, (<b>c</b>) 18 GHz, (<b>d</b>) 25 GHz, (<b>e</b>) 45 GHz, (<b>f</b>) 66 GHz, (<b>g</b>) 77 GHz, and (<b>h</b>) 120 GHz.</p> "> Figure 22
<p>Simulated and measured gain (G), radiation efficiency (eff), and axial ratio (AR) of the proposed antenna over frequencies in free space.</p> "> Figure 23
<p>Diversity gain (DG) and envelope correlation coefficient (ECC) results of MIMO antenna.</p> "> Figure 24
<p>Antenna’s estimated 3D radiation patterns on a commercial drone at (<b>a</b>) 8.5 GHz, (<b>b</b>) 10 GHz, (<b>c</b>) 18 GHz, (<b>d</b>) 25 GHz, (<b>e</b>) 45 GHz, (<b>f</b>) 66 GHz, (<b>g</b>) 77 GHz, and (<b>h</b>) 120 GHz.</p> "> Figure 25
<p>The simulated radiation pattern of the antenna when it is integrated with the drone at (<b>a</b>) 8.5 GHz, (<b>b</b>) 10 GHz, (<b>c</b>) 18 GHz, (<b>d</b>) 25 GHz, (<b>e</b>) 45 GHz, (<b>f</b>) 66 GHz, (<b>g</b>) 77 GHz, and (<b>h</b>) 120 GHz.</p> "> Figure 26
<p>The proposed antenna integrated with the drone: (<b>a</b>) power receiver probe around the transmitter drone in vertical and horizontal spaces, (<b>b</b>) overview of the simulated field test, and (<b>c</b>) horizontal assessment of the transmission and receiving power.</p> "> Figure 27
<p>The simulated air-to-ground and drone-to-drone transmission at different distances, azimuth, and elevation profiles (<b>a</b>) Co-polarization and (<b>b</b>) Cross-polarization.</p> ">
Review Reports Versions Notes

Abstract

:
Unmanned aerial vehicles (UAVs) need high data rate connectivity, which is achievable through mm-waves and sub-THz bands. The proposed two-port leaky wave MIMO antenna, employing a coplanar proximity technique that combines capacitive and inductive loading, addresses this need. Featuring mesh-like slots and a vertical slot to mitigate open-stopband (OSB) issues, the antenna radiates broadside and bidirectionally. H-shaped slots on a strip enhance port isolation, and a coffee bean metasurface (MTS) boosts radiation efficiency and gain. Simulations and experiments considering various realistic scenarios, each at varying vertical and horizontal distances, show steered beam patterns, circular polarization (CP), and high-gain properties, with a maximum gain of 13.8 dBi, an axial ratio (AR) <2.9, a diversity gain (DG) >9.98 dB, and an envelope correlation coefficient (ECC) <0.003. This design supports drones-to-ground (D2G), drone-to-drone (D2D), and drone-to-satellite (D2S) communications.

1. Introduction

The fast progression of the mobile Internet and the Internet of Things (IoT) has caused a significant increase in mobile data traffic. As a result, improving the capacity and data transmission rates of communication systems is crucial. This high-data-rate transmission is increasingly essential for unmanned aerial vehicles (UAVs) to enable real-time sensor and camera data transmission, remote control, and situations where UAVs function as aerial base stations. Within fifth-generation (5G) wireless technology, the adoption of the millimeter-wave (mm-wave) frequency range holds promise for enabling rapid data transfer and extensive throughput for large-scale IoT devices and other user equipment [1,2,3]. Furthermore, mm-wave signals (up to sub-THz) are widely anticipated for their exceptional capabilities, such as high-angle resolution, sensing, and robust resistance to interference, leading to widespread application in various scenarios [4]. However, the propagation of mm-wave signals encounters significant hurdles, such as substantial path loss and limited transmission range [5]. These challenges could potentially hinder the progress of mm-wave wireless communications.
The adaptable nature of communication systems supported by UAVs holds promise in addressing many limitations of mm-wave signals, offering wireless connectivity for radiofrequency (RF) devices in areas with signal shadows [6]. Aside from providing ample bandwidth, mm-wave and sub-THz frequency bands possess the potential to enable high data rates with minimal latency [7]. Moreover, connections to UAVs often operate in a line-of-sight (LOS) manner, bypassing common obstruction issues encountered in mm-wave deployments for cellular and local area networks (LANs). However, mm-wave communication with UAVs presents several technical challenges. These concerns include range limitations, power usage, and the capability to maintain directional alignment, particularly in high-mobility situations.
The design and placement of antennas play a pivotal role in determining communication range, with multi-array configurations offering significant advantages in long-range applications. As UAVs prioritize a flat and compact design to reduce wind resistance, they favor planar antenna structures for their lightweight and small-scale wireless systems. Apart from the miniaturizations, other factors such as resonant frequency, impedance bandwidth, polarization, and radiation pattern affect antenna performance in UAV-assisted communications. Studies have shown that lowering antenna weight by 25% can save around 10% of a UAV’s energy consumption [8].
Research on mm-wave UAV communication is relatively nascent, and the existing articles consider either the analysis and investigation of UAV communication networks or the required antennas for UAV communications at mm-wave band up to sub-THz. For example, the authors of [9] conducted laboratory tests to gauge the efficacy of beam tracking, while those of [10] employed ray tracing and channel sounding methods to evaluate directional antennas’ capabilities for drone-to-ground UAV communications operating at 28 GHz and 60 GHz. However, since this article focuses on antenna design and its analysis for UAV communications (D2D and D2S) at mm-wave and sub-THz, the state of the art will move further towards antennas designed for these UAV communications.

The State of the Art and Originality of Study

This section presents recent studies on antennas designed for UAV communications, including D2G, D2D, and D2S. It also considers the long- (low-frequency) and short-range (mm-wave and sub-THz) UAV communications. The researchers who wrote [11] employed reconfigurable radiation beams to guarantee extensive signal coverage for air-to-ground communications. Despite obtaining a high gain of 10.2 dBic, adjusting these beams required supplementary feed networks, which added complexity to the UAV design. Furthermore, it also offered size increments and thicknesses of up to 13 mm, escalating energy consumption. The authors of [12] introduced wide-band mm-wave antennas tailored for high-speed UAV communications. However, their designs encountered challenges such as low gain, increased thickness and larger size compared to the center frequency of 28.3 GHz, and a single-band dipole designed for D2G communication. It offered a deficient gain of 1.76 dBi, having a length of 110 mm. It could be considered for long-range communication, but obtaining a meager gain affects signal transmission, brings a high chance of signal interference, and results in poor link quality [13]. The dual-band MIMO antenna was designed for long-range and D2G communications. Using this, they obtained a peak gain of 8.7 dBi and an impedance BW of 21% at 2.4 GHz. The antenna also showed high isolation between the two ports and had a low profile. However, the BW was limited [14]. Another dual-band antenna was designed for D2G communication with a maximum gain of 9.55 dBi and a diameter of 76 mm. However, narrow BW and polarization assessment were not considered [15]. Last but not least, a single-band horn antenna was designed for the C-band at 5.4 GHz, offering a high gain of up to 16.25 dBi. However, the antenna was bulky (0.5 kg), nonplanar, and had linear polarization [16]. All the long-range antennas presented for D2G communication showed limitations such as narrow BW, oversized and heavy structure, lower gain compared to their size, single/dual bands, and linear polarization [11,12,13,14,15,16]. Researchers used slightly higher bands, mm-wave and sub-THz, to address these limitations. Alternatively, a separate strategy involves using dual-band leaky-wave antennas, offering frequency-scanning beams for prospective micro- and mm-wave mobile communications, covering both low and higher bands. They offer high gain but with high complexity and an extended length of more than 25 cm [17]. Additionally, the antenna design presented in [18] operates at 2.4/5.2 GHz and 60 GHz bands, catering to existing WiFi connections and anticipated high-data-rate Wi-Gig applications. The antenna offered high gain, having large dimensions and linear polarizations. Nonetheless, many of these designs encounter issues like elevated profiles and limited bandwidths in the mm-wave spectrum, rendering them capable of only a restricted performance for UAV applications. Moreover, their radiation directions could not be flexibly adjusted to the upper and lower half-spaces based on UAV mission requirements. In addition, some studies have delved beyond the typical frequency range, exploring communication frequencies above 100 GHz, where the available bandwidths surpass the lower mm-wave frequencies [19]. For instance, some worked on communication protocols and cooperative communication for UAVs at sub-THz and THz bands [20,21]. Not much work was performed considering the antenna design for both mm-wave and sub-THz bands. A single study examined two distinct systems: (a) a 28 GHz system resembling existing 5G new radio (NR) mm-wave deployments; (b) a theoretical 140 GHz system employing a connection akin to NR. The 140 GHz band represents a prospective sub-terahertz frequency spectrum for forthcoming 6G systems. They simulated and proved that the antenna works at these two bands when the drone is far from the base station, with a vertical distance of 25 m and a horizontal distance of 700 m [22].
Several techniques, like metamaterials and metasurfaces, were utilized to enhance the performance of antennas at microwave and mm-wave bands. Researchers have introduced MTSs and metamaterials into the realm of study to develop broadband antennas that offer advantages like low profile, affordability, and seamless integration. Metamaterials’ three-dimensional (3D) patterns can be transformed into two-dimensional (2D) patterns on a surface or interface by arranging multiple electrically small elements or apertures. This 2D version of a metamaterial is known as a metasurface (MTS). Unlike bulky 3D metamaterials, MTSs take up less physical space, resulting in lower losses. They hold great potential for revolutionizing various photonic and electronic device technologies. MTSs are engineered structures that exhibit unusual electromagnetic characteristics, such as negative permittivity and permeability. These MTSs, along with metamaterials, have garnered significant attention due to their unique properties, making them a promising technology for enhancing the performance of microwave components and overcoming limitations, particularly in antennas [23,24]. Initial investigations employed composite periodic structures such as EBG (electromagnetic bandgap) and AMC (artificial magnetic conductor) to create artificial-impedance-controlled surfaces. These structures aimed to reduce size, achieve wide bandwidth, and minimize back radiation. A cut circular ring MTS structure with a diameter of 34.7 mm was utilized to enhance an antenna’s gain up to 3 dB for the Sub-6 GHz band and reduce the radar cross-section of the antenna. The antenna obtained a maximum gain of 11.4 dBi. However, the complex structure operated at only a single broadband [25]. The unique shape, a slotted cutting edge, rectangular shape, that was used for the MTS structure, obtained a 5.66 dBi gain increment for the Sub-6 GHz band with the dimensions of 80 × 80 × 1.5 mm3. Cutting the slot from the MTS gave a CP feature but only at a single band [26]. The authors of [27,28] presented rectangular and slotted mushroom MTS structures for antenna performance enhancement for mm-wave and higher 5G bands, respectively. Despite obtaining high gains of 11 and 19 dBi, the structure was complex and multilayered, with higher dimensions related to the designing frequency.
Furthermore, their MTS structures’ unique multimode resonance characteristics and rectangular ring and split ring resonators have proven advantageous in broadband, multiband, and omnidirectional radiation applications. Nevertheless, they could offer only a single wide band [29,30]. MTS antennas maintain characteristics like low profile and ease of integration and usually exhibit a structure comprising multiple patch resonators. Having multiple arrays of resonators enables them to achieve broadside radiation gain and broadband bandwidth. It only had a single wide band of 3.8–8.5 GHz with a gain of almost eight dBi, but the antenna had a complex structure and a thickness of more than 4 mm [31]. Using characteristic mode analysis (CMA) may alter the structures of MTS, which significantly tightens the coupling between MTS structures and inevitably affects the desired modes.
Moreover, segmented small-size patches uniformly impact the desired radiation pattern, making it challenging to adjust a specific mode. Complex feeding structures can also influence the MTS’s characteristic mode, complicating the MTS antenna design. For example, dipole feeding and slot coupling feeding introduce new modes that expand antenna bandwidth. However, this demands a tighter correlation between the design of the MTS and the feeding structure, potentially increasing the antenna’s profile [32]. Table 1 indicates the performance comparison of some recent, related, state-of-the-art UAV communications at microwave and mm-wave bands and several works utilizing MTS structures for both bands.
To address the challenges and limitations of UAV communications and the applied antennas for these communications, a particular type of antenna should be designed to face all those challenges and obtain promising performances; these should exceed those of the currently available, similarly designed antennas for UAV communications and applied MTS structures, such as those presented in Table 1.
Contributions: This paper introduces a novel two-port leaky wave MIMO antenna designed to encompass a wide frequency range, catering to various UAV-assisted communication bands. The utilization of mm-wave networks is crucial for supporting high-speed, real-time communication links, while the UAV operates in-flight, offering essential wireless connectivity in hotspot areas. In addition to mm-wave networks, the antenna addresses communication sub-systems spanning multiple frequency bands, including the X-band (frequently used for applications above the atmosphere, such as radar and fixed and mobile satellite communication), K-band (for satellite receiver communication), and bandwidths operating within the 5G mm-wave and future 6G systems. This extensive frequency coverage supports various communication needs, including down-and-up link communications (drone communications to ground and satellite) and communications among drones. This comprehensive coverage facilitates UAV-assisted communication links, which is crucial for a broad range of applications, such as the most probable mm-wave and sub-THz frequency range for the future 6G systems. To sum up, the contributions of this work are summarized as follows:
  • A significant feature of the proposed antenna is its use of a coplanar proximity technique combining capacitive and inductive loading, enhanced by mesh-like slots and a vertical slot to mitigate open-stopband (OSB) issues. These design elements enable the antenna to radiate broadside and bidirectionally. Additionally, H-shaped slots on a strip enhance port isolation, and a coffee bean metasurface (MTS) boosts radiation efficiency and gain. Compared to other shapes like rectangles and rings, this unique structure can support multiple resonant modes, potentially enhancing the operational bandwidth due to its asymmetrical geometry. It also helps distribute the electromagnetic fields, reduces losses, enhances radiation efficiency, and offers effective CP.
  • This approach enables a compact, planar configuration suitable for installing unmanned aerial vehicles (UAVs). The antenna provides adaptable radiation patterns and the capability to steer beams across various frequency bands, catering to the signal coverage needs in both the upper and the lower half-spaces, meeting the demands of UAV missions. A significant achievement of this design lies in its attainment of a broad bandwidth within the mm-wave band and sub-THz, making it highly appealing for facilitating high-data-rate communications in UAV-assisted mm-wave scenarios.
  • It can offer high data rates to D2G base stations and allows for high data rate transmissions, enabling the transfer of high-definition video, real-time telemetry, and other data-intensive applications. In addition, higher-frequency bands are less crowded, reducing the likelihood of interference and improving the reliability and quality of communication. The high directional gain of the antenna enhances the effective communication range and signal quality by focusing the signal that is going toward the base station, reducing power requirements, and increasing the signal-to-noise ratio. For D2D communications, it can offer efficient swarm communication (high data rates at mm-wave and sub-THz), low latency (crucial for time-sensitive operations such as formation flying and collision avoidance), and directional communication (high directional gain helps in maintaining robust links between specific drones in a swarm). Last but not least, in D2S communications, the proposed antenna can be beneficial for long-range communication due to the high directional gain, higher BW, and the X-band, higher-frequency bands, which can support high bandwidth communication, and polarization flexibility (circular polarization helps reduce signal degradation due to atmospheric conditions and maintains consistent communication quality with satellites).
The paper’s structure is as follows: Section 2 elucidates the proposed multiband antenna’s geometric arrangement. Section 3 expounds the antenna’s operational principles. Subsequently, Section 4 furnishes performance metrics of the antenna, facilitating comparative analysis. Lastly, the paper is concluded by summarizing the study’s findings.

2. Materials and Methods

This section outlines the antenna configuration and design procedures. The MIMO LWA is initially constructed for specific frequency bands before integrating the MTSs. Incorporating the transversal and longitudinal slots and the MTS arrays enhanced the LWA performances. Moreover, the periodic H-shaped slots cut from a wide strip between two antennas improved the isolation. The radiation characteristics of the one-port configuration are examined and then extended to a two-port setup. Current- and electric-field concentrations were analyzed, and critical parameters were optimized based on actual values obtained from principle equations before detailing the design process.

2.1. Simulation

2.1.1. Design Process of the Antenna with a Single Port

The proposed LWA incorporates two rectangular patches capacitively fed through the coplanar waveguide (CPW) technique, incorporating chamfered edges. Initially, fundamental equations, as provided by [33], are employed to determine the dimensions of the rectangular patch and transmission line, enabling the antenna’s design at 8.5 GHz (where λ0/2 ≤ 18 mm). Despite well-established principles, these equations are presented again for more information.
W = c 2 f 0 ( ε r   +   1 ) 2 ,   ε e f f   = ( ε r + 1 ) 2 + ( ε r − 1 ) 2 ( 1 + ( 12 h w ) ) ,
∆ L = 0.412   h ( ε e f f + 0.3 ) ( w h + 0.264 ) ( ε e f f − 0.258 ) ( w h + 0.8 ) ,   L e f f = c 2 f 0 ε r e f f ,   L = L e f f − 2 ∆ ,   L g = L + ( 6 h )
W g = W + ( 6 h )
where W is the width of the patch, c is the velocity of light, f 0 is resonance frequency, ε r is the dielectric constant of the substrate, ε r e f f is effective dielectric constant, ∆ L is the extension length, L e f f is the effective length, h is the substrate thickness, L g is the length of the ground, and W g is the width of the ground. The calculated actual dimensions for the initial dimensions of the antenna are W = 12.17 mm, L = 9.8 mm, L g = 10.7, and W g = 12.94.
Subsequently, the design process for the proposed LWA begins, adhering to the following rules, equations, and standards, which are essential for every LWA. Altering an antenna’s length significantly affects its resonant frequency. Generally, the resonant frequency is inversely proportional to the antenna’s length: increasing the length decreases the resonant frequency, and decreasing the length increases it. This relationship is governed by the principles of electromagnetic waves and the antenna’s construction.
The design of the leaky wave structure prioritizes high directivity and gain, low loss, and structural simplicity. LWAs belong to a class of traveling-wave antennas, characterized by a wave propagating along a long structure compared to the wavelength. They generate currents propagating along its longitudinal direction, resulting in end-fire and back-fire radiation patterns due to the open-stopband (OSB) phenomenon, as described in [34,35]. These antennas can be effectively modeled using the balanced composite right-/left-handed (CRLH) transmission lines model, which combines the effects of capacitors and inductors. Enhancements are necessary to enable outward radiation from the antenna and the drone in a broadside direction.
Before delving into the design process of the antenna, it is essential to understand the principles of the LWA antenna and how slots and stubs can influence its characteristics, such as impedance bandwidth and radiation pattern. The proposed LWA incorporates asymmetric elements. It includes longitudinal slots to enhance OSB suppression and transverse slots to resolve OSB issues. These slots function similarly to a series of left-handed capacitances (CLs). As a result, they alter the radiation pattern, favoring the broadside direction over the end-fire or back-fire directions. Additionally, these slots help mitigate the surface waves produced by LWAs. The characteristic mode method is utilized to analyze and design these slots, evaluating their impact on the radiation pattern throughout the design process while considering the principles of asymmetry radiation, as illustrated in Figure 1 [36].
The periodic slots do not cover the entire width of the substrate (Ws1) and are positioned with an offset from one side (see Figure 1 and Figure 2). Moreover, the waveguide port should be excited with the TE10 mode instead of the TE01 mode used in the dielectric inset. With the proposed antenna shape and the enhancement of OSB suppression through an asymmetrical structure, the radiation mechanism of the new antenna is based on the asymmetry principle depicted in Figure 1. When the periodic slots are symmetrically positioned relative to the side walls, the vertically polarized TE10 mode does not induce any horizontal field in the antenna aperture, as shown in Figure 1a. This principle is utilized by the transverse parts of the mesh-like slots in the proposed design, meaning the electromagnetic waves emitted from the port have polarizations perpendicular to the longitudinal planes of both the antenna and the transmission line.
In the case of the stub-loaded LWA, the parallel plates (patch and ground layers) act as a filtering mechanism, permitting only the horizontal field to reach the waveguide aperture, positioned at height, m, from the dielectric interface in space. Thus, the symmetrical periodic structure makes the TE10 mode nonradiative when m is sufficiently high. The slots must be asymmetrically positioned to induce a potential difference in the parallel plates. This potential difference generates a horizontally polarized TEM wave, which can propagate to the aperture and radiate into the free space region, as shown in Figure 1b. The longitudinal parts of the mesh-like slots in the proposed design operate based on this principle. Therefore, by adjusting the slot asymmetry, one can control the excitation level of this radiating horizontal wave, thus altering the leakage rate of the TE10 leaky mode.
In contrast, the dielectric inset operates by exciting the TE10 mode, which is itself horizontally polarized. The TE10 mode alone is responsible for the antenna’s radiation, eliminating the need for introducing asymmetry to induce radiation in the structure, as depicted in Figure 1c. Additionally, altering the slot aperture is the only way to control the inset’s leakage rate. However, in this novel design and combination of slots, the leakage rate can be controlled by adjusting the slot asymmetry (spacing between the mesh-like slots and the longitudinal slot) while keeping the inset space value unchanged. After the antenna’s characteristic modes and the associated LWA principles were explained and discussed, the design process of the antenna started following the rules and equations presented in the above sections. Figure 2 displays the simulated prototype and antenna dimensions, highlighting each component’s crucial parameter in antenna design. As presented in Figure 2, the design process begins by designing one port antenna following each design step shown in Figure 2a–f. When the proposed antenna’s characteristics with one port are assessed, it is expanded to a two-port antenna by adding the antenna next to it and applying the neutralization network between them. The design steps are explained as follows, based on what is depicted in Figure 2. In addition, Table 2 shows the dimensions and optimized values of the proposed antenna parts. Each value was optimized after the actual values were obtained using the related equation, and the parametric studies were performed.
The proposed antenna is fabricated on a layer of Rogers 4830 substrate, with a dielectric constant of 3.2, a loss tangent of 0.0006, a dielectric thickness of 0.127 mm, and a copper thickness of 18 μm. The primary concept behind this design is to create a semi-flexible antenna that seamlessly integrates into mm-wave sub-THz circuitry or packaging for drones. In addition, the proposed LWA design is fundamentally fed by a coplanar proximity technique [37], which employs an open-ended microstrip line. This feeding technique utilizes two distinct coupling mechanisms: capacitive and inductive coupling. In the configuration employing capacitive coupling, a square patch is positioned such that the central line of the patch aligns at a half wavelength from the tip of the open-ended feed line. This arrangement results in the maximum magnitude of the voltage standing wave on the feed line, coinciding at the center of the patch. As a result, a fundamental resonant current pattern (illustrated with arrows in Figure 3b) perpendicular to the feed line is excited through coupling across the gap.
Consequently, this setup produces linearly polarized radiation in the distant area. Conversely, with inductive coupling (as depicted in Figure 3a), the edge of the square patch lines up with the tip of the feed line. In this scenario, the patch is powered by the highest amplitude of the current standing wave, which is 90° out of phase with the voltage standing wave. In contrast to capacitive coupling, this configuration’s surface current and radiation polarization are perpendicular to their counterparts. Circular polarization (CP), essential for UAV communication compared to linear polarization, can be understood as a combination of two linear polarizations: spatially orthogonal and phase quadrature.
By combining Figure 3a,b, the patch antennas can emit orthogonal linear polarizations as in step one of the designs shown in Figure 2a. In addition, this configuration requires two-element patches to resonate at slightly different frequencies.
In this particular design, the shared feed line enables the application of two distinct coupling methods to the two patches, as previously described. As the voltage rises and the current diminishes along the open-ended feed line, depicted conceptually in Figure 4, a process ensues wherein the decreasing current along the feed line induces diminishing currents in the patch below it, utilizing inductive coupling (indicated by black arrows in Figure 4). Simultaneously, the voltage on the feed line prompts decreasing surface currents, facilitated by capacitive coupling, to flow from the gap side towards the opposite end of the other patch (also indicated by black arrows in Figure 4). Consequently, if the two patches resonate at slightly different frequencies, then their currents appear in phase. Therefore, meticulous adjustment of the patch sizes is essential to achieve CP through this antenna configuration. This adjustment generates the necessary 90° phase difference between the resonant frequencies of the two patches. In Figure 4’s arrangement, left-handed circular polarization (LHCP) is achievable by ensuring the capacitively coupled patch resonates at a higher frequency (25 GHz). On the contrary, to achieve right-handed circular polarization (RHCP), the inductively coupled patch must resonate at a lower frequency (8.5 GHz). It should be mentioned that the black arrows in Figure 4 indicate the current direction of the surface due to the capacitive and inductive effects.
Following the previous equations used to define the overall dimensions of the patch, substrate, and ground, the following equations are utilized to obtain the dimensions of the capacitive and inductive patches after using this feeding technique. According to the voltage and current arrows and the previous explanations, two types of coupling occur in this structure at two different frequencies. These are described here.
(1) At a frequency, f1, with guided wavelength λ g 1 , l = (n λ g 1 )/2, where l is the distance from the red dashed line (Figure 3) and n is an integer. The patch will resonate having this distance and will do so when the W p (in Figure 2) is half of the first effective wavelength (vertical polarization). (2) At a different frequency, f2, with guided wavelength λ g 2 , l = ( n λ g 1 )/2 + λ g 2 /4, where n is also an integer, and the patch will resonate when L p is almost half of the second effective wavelength (horizontal polarization). In these two equations, f1 and f2 are 8.5 GHz and 25 GHz, respectively. After calculating, the actual values of l for cases number 1 and 2 are 4.9 mm and 5.8 mm, considering n = 1.
After achieving resonances at frequencies of 8.5 GHz and 25 GHz and confirming the functionality of the CP concept, both rectangular patches undergo modification by incorporating meshed slots. These slots, comprising transverse and longitudinal slots, serve specific purposes within the antenna system. Following the LWA principles in Figure 1, the transverse slots (Figure 1a and Figure 2b) primarily act as left-handed capacitances in series, while the longitudinal slots (Figure 1b and Figure 2c) alter currents and introduce an additional frequency band spanning 70–75 GHz. Furthermore, these transverse and longitudinal slots improve the impedance BW of the LWA and address inherent OSB issues in leaky-wave antennas.
These slots play a pivotal role in guiding and facilitating wave propagation, promoting a constructive radiation pattern in the central region of the patch, and enhancing circular polarization. However, incorporating these slots and cuts into the patch has a drawback, resulting in decreased radiation efficiency and gain. Additionally, the radiation pattern tends to lean towards an end-fire orientation, a common characteristic in LWAs. Therefore, transforming from end-fire to broadside radiation directed outward from the drone is vital for D2D, D2G, and D2S. One method to achieve this transformation involves introducing a matching element in the feeding section. A three-step impedance-matching approach is utilized to mitigate the OSB issue. This process converts the 50 Ω feeding to a 110 Ω impedance observed from the patch (Figure 2d). Subsequently, another longitudinal slot is introduced adjacent to the feed line, altering the transmission mode and redirecting surface currents. This alteration aims to create an additional resonance around 120 GHz, further enhancing the antenna’s bandwidth. Conceptually, the vertical slots can be envisioned as two impedances connected by a transmission line within the equivalent circuit model, contributing to the antenna’s overall performance and characteristics.
The structural changes applied to the capacitive coupling mechanism of the antenna play a crucial role in widening the bandwidth for both resonance modes of the antenna across the millimeter-wave and sub-THz frequency bands, extending BW up to 5 GHz. Moreover, incorporating vertical slots and an additional slot, which is capacitively loaded and positioned adjacent to the meshed slots but distanced from the transmission line, serves the purpose of suppressing surface waves around the patches. This suppression significantly contributes to improving the antenna’s impedance bandwidth. Notably, the edges neighboring the transmission line are curved to minimize surface waves in the proximity, thereby improving bandwidth at lower frequencies and within the 5G mm-wave spectrum.
The principles derived from LWA theories and inductive coupling feeding, explained thoroughly in the previous paragraphs in this section, provided a baseline for establishing the dimensions of patches, slots, and gaps [33,38]. Since the proposed antenna facilitates multiple operational radiation modes with distinct radiation requirements for UAV-assisted communications, considering the influence of fringing fields and surface current distribution around the patch’s perimeter is essential. In addition, a parametric study of the vital parameters of the design provides us with fine-tuning tools to achieve the desired resonant frequency and working BW. (The surface current assessment was performed by CST studio).
As can be observed from Figure 5 for the proposed antenna, a strong current density is mainly concentrated on the inner edge of the square patches with the direction along the x-axis, verifying that the fundamental mode (TM10) of the inner patch is activated at 8.5 GHz. Furthermore, the current solid density can be observed around the gap between the patches and the transmission line, showing practical resonance around 8.5 GHz, and the antenna’s omnidirectional radiation at its early stage is slightly changing to end-fire radiation when it is steered for frequencies higher than 80 GHz (exp. Figure 5c at 120 GHz).
According to the transmission line and coupling proximity model, the antenna’s total radiation pattern exhibits a radiation null at the boresight in the vertical plane. This null arises from the self-cancellation of E-field radiation from distinct radiating apertures in the far field. Consequently, an almost omnidirectional radiation pattern can be achieved in the azimuth plane around the 8.5 GHz frequency. As depicted in Figure 6, the current intensity on the outer edges of the patch at 8.5 GHz, 25 GHz, and 120 GHz is considerably strong, and each edge has almost similar amplitude distribution, especially at two higher modes. The orientation of the current direction is very similar to that of the TM11 mode of a square patch. The antenna also employs the TM11 mode of the patch antenna for radiation to achieve omnidirectional at the lower bands. Introducing parasitic slots on the capacitive patch on the transmission line’s right side improves this mode’s characteristics. The operational mechanism of the TM11 mode is extensively discussed in [39] and is only briefly interpreted here. In addition, it is radiated towards the front and makes an end-fire radiation (90°) at a higher frequency of around 120 GHz.
To achieve CP and broadside radiation operating around the 22–25 GHz band for mm-wave 5G and in the 110–130 GHz band for sub-THz communication, two orthogonal modes, TM10 and TM01, created by employing transversal and longitudinal slots (with this knowledge that slots are mode perturbation), are incorporated into the antenna design. By adjusting the size parameter of these slots (in terms of length and width), both modes are aligned to possess equal amplitudes and maintain a phase difference of 90°, resulting in favorable CP performance. Explicitly referring to Figure 6b,c, the observed anticlockwise rotation of currents on the patches signifies the realization of the CP wave propagating in the broadside direction (the red and orange color of the current figures.
Some of the most effective design parameters are presented in Figure 7 to help understand how the proposed antenna can be tuned to different frequencies. Figure 7a presents the tuning of the feed line length (Lf = Lf1 + Lf2 + Lf3). It shows that, when the length is reduced from 12 mm, all three working bands shift to lower frequencies. The patch length (Lp) increment follows the same trend as the feed line. Conversely, when reduced to less than 4 mm, the other two working bands at mm-wave and THz frequencies are significantly suppressed. The optimization of the ground dimensions (Lg, Wg) is depicted in Figure 7c. It also indicates that the ground length follows the same tendency as the feed line. However, reducing the ground width significantly affects impedance matching and creates more unwanted bands when reduced to 2 mm.
Last but not least, increasing the patch’s width increases the frequency band and shifts them slightly to a lower band, while the opposite happens in the case of a reduction in width. After a parametric study of the single-port antenna, the impedance bandwidth results of the antenna for each step of the design (shown in Figure 2) are depicted in Figure 8. The variation of the reflection coefficient with the frequency illustrates that three modes of resonance are obtained for each stage, and they were enhanced and widened by adding the transversal and longitudinal slots and the three-step feed line. The figure elucidates the wide-band mm-wave antenna’s operational mechanism and proves that following the operational principles could offer us the required resonation. The curve exhibited in Figure 8 illustrates the excitation of three resonant modes occurring at approximately 8.5 GHz, 25 GHz, and 120 GHz (black dashed line for the final stage). It should be mentioned that a bit of optimization can achieve the exact resonances. Figure 9 shows that the proposed MIMO antenna provides an isolation of more than 20 dB for almost all the working bands when no metasurface is integrated with the antenna.

2.1.2. The Proposed MIMO LWA and Incorporation of Neutralization Network and MTS Cells

Upon achieving the results from the initial single-port antenna setup, the design expanded into an MIMO antenna system with two ports. The objective was to broaden the overall bandwidth of the antenna, improve the bit rate transmission, and alleviate the path loss and fading effects prevalent in MIMO communication, especially in D2D and D2G communications. However, introducing an additional antenna increased mutual coupling between the two ports, undermining the desired isolation in MIMO communications. Various techniques were explored to enhance isolation between the ports. These included implementing a neutralization network, utilizing shorting posts, or increasing the physical separation between the ports, which unavoidably increases the antenna’s profile. A narrow copper tape was introduced between the antennas to tackle this challenge. Subsequently, sixteen periodic H-slots (comprising three different scaling factors and sizes) were carved from this copper tape. This alteration served multiple purposes: enhancing isolation between the ports, diminishing surface waves around the CPW feeding grounds, and improving the antenna’s impedance bandwidth. Introducing these H-slots inherently endowed the antenna with multiband characteristics. It also enhances the impedance BW level at the working bands and incorporates more poles across the broad operating bandwidth encompassing the millimeter-wave and sub-THz regions. Figure 10 shows a perspective view of the MIMO antenna, showing each layer and its associated materials.
For further investigation, electric field distribution on the radiating elements in the three main desired resonance frequencies and modes and other poles in the working BW are illustrated in Figure 11. Figure 11 also shows the electric field variation of the antenna without the MTS cells at the back. It also indicates how the antenna works in the desired modes and frequency bands when port one is active. It can be seen that the two rectangular slots and feed lines at their centers are responsible for 8.5 GHz resonance. In contrast, in the 25 and 75 GHz, mostly the transversal and longitudinal slots at the corners and the gap between the feed line and patches create resonances, and almost no surface wave is concentrated around the slots. Therefore, according to the electric field magnitude, when one port is excited, the other one is completely isolated; this means that the mutual coupling between two elements is low, which is one of the critical challenges in MIMO antennas.
Furthermore, the dominant modes can be seen with the determined antenna length and width, and other lower modes can be achieved by using these slots and the three-step matching feed line. In addition, strong fields are observed around the edges of the patches and in proximity to the substrate’s edge, indicating that this resonance is associated with the 8.5 GHz and its mode of TM11. Additionally, the intensity of the electric field resembles that of a patch-loaded monopolar antenna, suggesting that the patch TM11 mode introduces the second resonance. Lastly, examining the electric field in Figure 11d,e reveals that the mode related to the 115 GHz and 120 GHz resonances corresponds to the leaky wave mode. It is distributed throughout the leaky wave’s transmission line, capacitive patch, and inductive patch based on the characteristic of a higher mode. This observation confirms that the resonant mode at 120 GHz primarily involves the TM22 mode.
Following the characterization of the antenna across three antenna modes, it undergoes loading with MTS elements. An MTS often comprises a thin layer containing regularly spaced scatterers within a base material. It exhibits remarkable electromagnetic characteristics like elevated surface impedance, coherent reflection, negative refractive indices, and propagation constants with spatial gradients. When dealing with a finite-sized MTS connected closely to a conductive ground and a primary source, using characteristic mode analysis offers advantages in system assessment and facilitating the development of metasurface antennas (MAs) [40,41]. The resonant modes of the proposed antenna and the resonant mode of the coffee bean unit cell are amalgamated, followed by an investigation into their characteristics. In the design phase of the proposed unit cell, emphasis is placed on ensuring that the lattice period is approximately 0.2 of the wavelengths at the resonance of the operating mode, thus inducing a local TM01 resonance for each unit cell. Moreover, integrating the MTS and antenna modes enhances the antenna’s performance, resulting in wider bandwidth, higher gain, and improved radiation efficiency.
Initially, the radiation and impedance characteristics of a single-element coffee bean metasurface (depicted in Figure 12d) are evaluated. The reflection and transmission coefficient results are presented in Figure 12a, showcasing passbands at the desired frequencies. Additionally, Figure 12b,c depict the permittivity and permeability results of the single-element metasurface. The negative index results of the metasurface element shown in Figure 12 demonstrate its capability to enhance the proposed antenna’s radiation characteristics. The results of integrating the metasurface cells with the proposed antenna will be discussed later in the text.

3. Fabrication, Testing, and Comparison of Simulated and Measured Results

3.1. Antenna Characteristics in Air and Integrated with Drones

Figure 13, Figure 14 and Figure 15 depict the simulated setup, the measurement setup in MW/mm-wave, and the measurement setup at sub-THz of the proposed antenna attached to drones (similar to industrial drone: an accurate 3D model for a commercial drone downloaded from [42]). It is a typical dual-copter with a diagonal dimension of 18.50 × 9 cm2. It is suitable for various conditions, such as horizontal and vertical movements between two drones and having two antenna arrays on the drone. The impact of the drone’s structure on the antenna parameters is subsequently assessed. Typically, larger drones are constructed from carbon fiber. However, in simulations, we employ PEC material for the conductors and acrylonitrile butadiene styrene (ABS) for the drone section to streamline the model and maintain the comparable characteristics. This approach also allows us to investigate whether the influence of the drone’s frame becomes more severe at higher frequencies. The white compartment beneath the black drone and the thinner layer beneath the drone are made of Zetamix. An additional dielectric layer of Zetamix is considered in the measurement assessment. This layer is 3D printed using Zetamix filament (the Zetamix Epsilon range of products contains three filaments, each with different permittivity going from 2.2 for the lowest to 7.5 for the highest; these products are characterized by their low dielectric losses and high heat resistance for enduring up to 110 degrees Celsius HDT). It was mentioned that the drone was 3D printed using ABS material with a dielectric constant of 2.2. After 3D printing, the lowest layer of the drone faces the antenna, which is 3D-printed using Zetamix. The reason for choosing Zetamix is its dielectric constant of 7.5, which is in a similar range to that of polyvinylidene fluoride (PVDF) ( ε r = 6–9). PVDF is one of the most used materials for building a random body.
The simulated and measured reflection and transmission coefficient results of the antenna for the conditions presented in Figure 13, Figure 14 and Figure 15 are depicted in Figure 16 both in the air and when attached to a commercial-sized drone. A reliable and excellent agreement exists between the simulated and measured results. Furthermore, as explained before, all the antenna resonances and modes were obtained in the simulated and measured results. Wide BW is achieved for the lower and higher antenna modes and high isolation of more than 25 dB is also achieved with and without drone integration. Figure 17 illustrates the reflection and transmission coefficient results obtained when two arrays are attached to a single drone. The reflection coefficient results did not change dramatically, maintaining a high isolation of more than 30 dB between the ports. The transmission coefficient results for the antennas on two drones, considering various vertical and horizontal distances between them, are presented in Figure 18. This figure demonstrates that the transmission coefficient level decreases as the distance between the drones increases. For instance, at a horizontal distance of 25 m, the transmission coefficient drops to −90 dB. Moreover, a similar trend is observed for vertical distances up to −85 dB, albeit with a slightly higher transmission coefficient.
After investigating the S-parameter results of the antenna both in free space and attached to the commercial drone, the radiation characteristics of the antenna, such as radiation pattern, gain, and axial ratio (AR), should be evaluated. The radiation pattern measurement of the antenna is carried out in a mm-wave chamber, as shown in Figure 19. Figure 20 indicates the simulated and fabricated radiation pattern for Co. and cross-polarization in free space. A good agreement exists between the simulated and measured patterns of the antenna. The antenna’s radiation pattern varies between 30° and 60° at lower frequencies and is predominantly around 180° at higher frequencies. The half-power beam widths (HPBW) are approximately 50 and 46, respectively.
Furthermore, the steering angle ability of the proposed leaky wave antenna is investigated, as indicated in Figure 21. It also demonstrates that the antenna can offer a wide range of steering angles with frequencies up to 180° for D2D and D2G communication. The simulated cross-polarization (X-pol) levels in the vertical plane will not be visible due to their relatively low magnitude, less than −30 dB. It is important to note that, given the random distribution of most wireless devices in the environment, antennas with omnidirectional characteristics, multi-beam capabilities, or those capable of beam steering offer more extensive coverage. Hence, the proposed MIMO LWA is a promising choice for such requirements.
Figure 22 showcases the simulated and measured outcomes regarding the proposed antenna’s gain, radiation efficiency, and axial ratio (AR). The results indicate a consistent AR level below three across the operational frequency range, both in simulation and measurement. Notably, it reveals a maximum gain of 13.8 dBi, consistently exceeding 5.5 dBi across most frequency bands. Furthermore, the antenna demonstrates notable radiation efficiency, surpassing 77% in higher bands and exceeding 85% for frequencies below 80 GHz. Moreover, it illustrates minimal variations in gain for LHCP and RHCP within the operational band, staying below 1.1 dB and 0.5 dB, respectively (although not depicted in figure). However, a marginal inconsistency between the simulated and measured outcomes is discernible in Figure 22.
After assessing the antenna’s performance in airborne conditions and its successful integration with a commercial drone, the next phase involves evaluating its MIMO antenna capabilities and diversity. Figure 23 displays the variation of the proposed MIMO leaky wave antenna in terms of diversity gain (DG) and envelope correlation coefficient (ECC). As depicted in Figure 23, the proposed MIMO antenna exhibits a notable diversity gain, measuring less than 9.98, indicating its ability to provide improved signal diversity. Additionally, it demonstrates an envelope correlation coefficient (ECC) of less than 0.003 within the operational frequency band of the antenna. These results suggest that the proposed MIMO antenna offers a high diversity gain and maintains a low ECC level, emphasizing its potential for practical MIMO applications.

3.2. Antenna System Positioning on Drones and Its Assessment for UAV Communications

Two practical array configurations are considered for assessment: (1) arrays positioned beneath the drone to ensure coverage of the lower hemisphere; (2) antenna arrays located on the drone’s sides and tilted at a −45° angle relative to the drone’s plane. Each array covers sectors of approximately 90°, both accumulating 180°. Before assessing these two configurations, it is better to elaborate on the antennas’ 3D radiation pattern at each radiation mode. After evaluating the antenna’s radiation in free space, the radiation patterns are defined when the antenna is attached to the drone. The 3D radiation patterns of the proposed antenna in all three modes and different bands are simulated, as shown in Figure 24. The findings displayed in Figure 24 and Figure 25 suggest the presence of conical radiation patterns in the vertical plane (XOZ plane and towards the ground) at the lower bands and dual-directional (both bidirectional and end-fire radiations) beams in the horizontal plane (XOY plane) at higher bands.
Figure 26 shows the simulation setup for the power received by the receiver drone and tests the antenna performance when it is attached to the drone and the electronic circuitry module. Figure 26a depicts the simulation setup to obtain the received power from the transmitter drone at various distances. The transmitter drone is located at the center of the field. Then, several power receiver probes (shown by blue and green markers in CST software, version 2023) are situated at vertical and horizontal distances from the transmitter drone. They are also located at 45 degrees between each axis. After the simulated calculation, the antenna is attached to the communication module, and the 3D-printed drone is used to see the antenna performance in real scenarios.
Figure 27 shows the maximum received power at different locations (both horizontal and vertical distances and various angles) around the proposed antenna at the center. As expected, the received power decreases when the distances increase. However, since the proposed antenna offers circular polarization, it can provide consistent radiation for both left- and right-hand polarization (LHCP and RHCP). The received power is assessed at up to 1 km distances around the transmitter drone. However, we focus more on the drone for indoor communication, such as performing in a large factory. The simulated evaluation of the received power shows that the proposed MIMO antenna can perform perfectly for indoor communication for both ground-to-drone and drone-to-drone communications.
Table 3 shows the comparative performance of several MIMO antennas used for UAV communication. The comparison is made regarding dimensions, operating bandwidth, radiation characteristics (maximum gain and efficiency), coverage, and diversity performance of the proposed MIMO antenna (DG, ECC, and isolation). Table 3 also demonstrates that the proposed antenna system surpasses these recent similar works and has excellent potential as a multiband operating system for both mm-wave and sub-THz communications in UAV applications.

4. Conclusions

The article evaluates the potential of millimeter-wave (mm-wave) communication using unmanned aerial vehicles (UAVs) by developing and testing an MIMO antenna system with circular polarization. It is a conformal antenna which is designed on a flat, broad space and consists of two individual antennas which work together as a single antenna to transmit or receive radio waves. This system utilizes a coffee bean metasurface structure to achieve low loss, high efficiency, and easy integration with mm-wave circuits. The antenna’s radiation properties were analyzed, considering the drone frame’s impact on signal coverage. Testing covered multiple frequency bands: X-band, K-band, Ku-band (8.1–24.7 GHz), 5G mm-wave bands (66–68 GHz, 74–78 GHz), and sub-THz bands (80–140 GHz). Results showed an axial ratio below 3, a maximum gain of 13.8 dBi, and radiation efficiency over 85% for frequencies below 80 GHz, with low cross-polarization levels. The antenna demonstrated significant diversity gain and a low envelope correlation coefficient (ECC), which is crucial for MIMO applications. Both simulations and measurements were consistent, confirming the antenna’s wide bandwidth, high gain, and flexible radiation patterns, making it suitable for UAV-assisted 5G mm-wave and sub-THz wireless communications.
It also included practical evaluations on a 3D-printed drone model, with future plans to test the antenna on an actual Parrot drone to further validate our design methodology. The proposed antenna setup facilitated the investigation of two potential arrangements for positioning antenna arrays on a UAV, providing insights into hemispheric coverage, gain distribution, and received power across different frequency bands. Leveraging UAVs as aerial wireless communication platforms presents a promising alternative to traditional cellular systems, though energy constraints often limit their operational duration. Addressing these challenges, our antenna design emphasizes miniaturization and lightweight construction, which are crucial for reducing energy consumption and improving UAV endurance. Our results underscore the advantages of employing multi-array systems, particularly over extended distances, and highlight the potential for indoor communication and tracking within large structures like factories. This work contributes to implementing multiband antennas with adaptable radiation patterns. It provides valuable insights into the landscape beyond 5G and sub-THz wireless systems for UAV-enabled communications.
The proposed high-performance multi-functional antenna was designed for UAV-assisted wireless communication, enhancing D2G, D2D, and D2S applications. The associated features and benefits are as follows. (1) D2G communications: These operate at X-band (8–12 GHz) and mm-wave (30–300 GHz) frequencies, providing high data rates for HD video streaming, real-time telemetry, and sensor data transfer. The obtained operating bands can reduce interference, improve reliability, and enhance communication range and signal quality through their high directional gain. The related applications for this type of UAV communication include surveillance, high-speed sensor data transfer, and reliable command/control links. (2) D2D communications: These can support efficient swarm communication with high data rates at mm-wave and sub-THz frequencies. This ensures low latency for time-sensitive operations and robust directional links between drones. The related applications are synchronized search-and-rescue operations, environmental monitoring, and networking. (3) D2S communications: These can provide long-range communication with high bandwidth and directional gain. X-band and higher frequencies enable high bandwidth communication, and circular polarization improves satellite signal quality. The related applications are beyond visual line of sight (BVLOS) operations, data relays in remote areas, and global positioning/navigation support. All these advancements allow for more reliable, efficient, and versatile UAV communications across various applications.

Author Contributions

Methodology, T.S., N.T. and K.K.; Software, S.S., A.J.A.A.-G. and O.I.E.; Validation, S.S., N.T., A.J.A.A.-G. and S.K.; Formal analysis, T.S., N.T., N.R., A.B.A. and K.K.; Investigation, T.S., A.J.A.A.-G., N.R. and A.B.A.; Resources, A.J.A.A.-G. and A.B.A.; Data curation, T.S., S.S., S.K. and A.A.A.; Writing—original draft, T.S.; Writing—review &editing, S.S., N.T., N.R. and S.K.; Visualization, S.S., N.T., A.A.A.; Supervision, N.T. and S.K.; Project administration, N.T.; Funding acquisition, A.A.A., N.R., K.K., A.B.A. and O.I.E. All authors have read and agreed to the published version of the manuscript.

Funding

This work was funded by the Deanship of Scientific Research at Jouf University under Grant Number (DSR2022-RG-0110).

Data Availability Statement

The original contributions presented in the study are included in the article, further inquiries can be directed to the corresponding author.

Acknowledgments

The author thanks the support of the Atlantic Technological University (ATU) and Jouf University.

Conflicts of Interest

The authors declare no conflicts of interest.

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Figure 1. Asymmetry radiation principle: (a) TE10 mode in even structure; (b) leaky TE10 mode in the uneven structure; (c) leaky TE01 mode in even structure (θ is the approximated maximum angle of the radiation and the red arrows show the direction of the field) [36].
Figure 1. Asymmetry radiation principle: (a) TE10 mode in even structure; (b) leaky TE10 mode in the uneven structure; (c) leaky TE01 mode in even structure (θ is the approximated maximum angle of the radiation and the red arrows show the direction of the field) [36].
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Figure 2. The design configuration of the single antenna and MIMO, including each step of the design and the proposed prototype ((a) step 1, (b) step 2, (c) step 3, (d) proposed, (e) front view of the MIMO, and (f) ground view of the MIMO).
Figure 2. The design configuration of the single antenna and MIMO, including each step of the design and the proposed prototype ((a) step 1, (b) step 2, (c) step 3, (d) proposed, (e) front view of the MIMO, and (f) ground view of the MIMO).
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Figure 3. Coupling mechanisms of coplanar proximity feed ((a) inductive coupling and (b) capacitive coupling).
Figure 3. Coupling mechanisms of coplanar proximity feed ((a) inductive coupling and (b) capacitive coupling).
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Figure 4. The surface current mechanism of the proposed antenna (the blue and red lines are the current and voltage standing wave amplitude, respectively).
Figure 4. The surface current mechanism of the proposed antenna (the blue and red lines are the current and voltage standing wave amplitude, respectively).
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Figure 5. The surface current distribution of the initial design of the antenna at (a) 8.5 GHz, (b) 25 GHz, and (c) 120 GHz.
Figure 5. The surface current distribution of the initial design of the antenna at (a) 8.5 GHz, (b) 25 GHz, and (c) 120 GHz.
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Figure 6. The surface current distribution at (a) 8.5 GHz, (b) 25 GHz, and (c) 120 GHz.
Figure 6. The surface current distribution at (a) 8.5 GHz, (b) 25 GHz, and (c) 120 GHz.
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Figure 7. Parametric study of the single element antenna: reflection coefficient results of (a) Lf, (b) Lp, (c) Lg, Wg, and (d) Wp.
Figure 7. Parametric study of the single element antenna: reflection coefficient results of (a) Lf, (b) Lp, (c) Lg, Wg, and (d) Wp.
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Figure 8. Reflection coefficient results of the proposed single-port antenna for each stage.
Figure 8. Reflection coefficient results of the proposed single-port antenna for each stage.
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Figure 9. S-parameters results of the MIMO antenna without the neutralization tape and metasurface elements.
Figure 9. S-parameters results of the MIMO antenna without the neutralization tape and metasurface elements.
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Figure 10. A perspective view of the MIMO antenna showcasing each layer and their materials.
Figure 10. A perspective view of the MIMO antenna showcasing each layer and their materials.
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Figure 11. The electric field of the proposed two ports MIMO antenna at different frequencies (a) 8.5 GHz, (b) 25 GHz, (c) 75 GHz, (d) 115 GHz, and (e) 120 GHz.
Figure 11. The electric field of the proposed two ports MIMO antenna at different frequencies (a) 8.5 GHz, (b) 25 GHz, (c) 75 GHz, (d) 115 GHz, and (e) 120 GHz.
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Figure 12. The single-element coffee bean metasurface’s characteristics: (a) S-parameters results; (b,c) permittivity and permeability results; (d) single element of the coffee bean.
Figure 12. The single-element coffee bean metasurface’s characteristics: (a) S-parameters results; (b,c) permittivity and permeability results; (d) single element of the coffee bean.
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Figure 13. Simulation setup of the proposed antenna: (a) beneath view, (b) upper view, (c) two arrays of the proposed antenna on one drone, (d) two drones and antenna systems with vertical space, and (e) two drones and antenna systems with horizontal space.
Figure 13. Simulation setup of the proposed antenna: (a) beneath view, (b) upper view, (c) two arrays of the proposed antenna on one drone, (d) two drones and antenna systems with vertical space, and (e) two drones and antenna systems with horizontal space.
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Figure 14. The measurement setup of the proposed antenna in the air at microwave (up to 10 GHz) and mm-wave (up to 30 GHz) bands: (a) without a drone and (b) with a drone.
Figure 14. The measurement setup of the proposed antenna in the air at microwave (up to 10 GHz) and mm-wave (up to 30 GHz) bands: (a) without a drone and (b) with a drone.
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Figure 15. The measurement and calibration setup of the antenna in the air at the sub-THz band (77–140 GHz).
Figure 15. The measurement and calibration setup of the antenna in the air at the sub-THz band (77–140 GHz).
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Figure 16. Simulated and measured S-parameter results of the proposed antenna after integrating with metasurface in air and attached to a drone: (a) transmission coefficient and (b) reflection coefficient.
Figure 16. Simulated and measured S-parameter results of the proposed antenna after integrating with metasurface in air and attached to a drone: (a) transmission coefficient and (b) reflection coefficient.
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Figure 17. Reflection and transmission coefficient results when two arrays of the antenna are attached to one drone.
Figure 17. Reflection and transmission coefficient results when two arrays of the antenna are attached to one drone.
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Figure 18. Transmission coefficients results of the antenna on two drones with different (a) vertical and (b) horizontal distances.
Figure 18. Transmission coefficients results of the antenna on two drones with different (a) vertical and (b) horizontal distances.
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Figure 19. Radiation pattern measurement of the antenna in the mm-wave chamber.
Figure 19. Radiation pattern measurement of the antenna in the mm-wave chamber.
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Figure 20. The simulated and measured electric and magnetic field of the antenna in free space at (a) 8.5 GHz, (b) 18 GHz, (c) 25 GHz, (d) 66 GHz, and (e) 77 GHz.
Figure 20. The simulated and measured electric and magnetic field of the antenna in free space at (a) 8.5 GHz, (b) 18 GHz, (c) 25 GHz, (d) 66 GHz, and (e) 77 GHz.
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Figure 21. Beam steering of the proposed antenna at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
Figure 21. Beam steering of the proposed antenna at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
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Figure 22. Simulated and measured gain (G), radiation efficiency (eff), and axial ratio (AR) of the proposed antenna over frequencies in free space.
Figure 22. Simulated and measured gain (G), radiation efficiency (eff), and axial ratio (AR) of the proposed antenna over frequencies in free space.
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Figure 23. Diversity gain (DG) and envelope correlation coefficient (ECC) results of MIMO antenna.
Figure 23. Diversity gain (DG) and envelope correlation coefficient (ECC) results of MIMO antenna.
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Figure 24. Antenna’s estimated 3D radiation patterns on a commercial drone at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
Figure 24. Antenna’s estimated 3D radiation patterns on a commercial drone at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
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Figure 25. The simulated radiation pattern of the antenna when it is integrated with the drone at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
Figure 25. The simulated radiation pattern of the antenna when it is integrated with the drone at (a) 8.5 GHz, (b) 10 GHz, (c) 18 GHz, (d) 25 GHz, (e) 45 GHz, (f) 66 GHz, (g) 77 GHz, and (h) 120 GHz.
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Figure 26. The proposed antenna integrated with the drone: (a) power receiver probe around the transmitter drone in vertical and horizontal spaces, (b) overview of the simulated field test, and (c) horizontal assessment of the transmission and receiving power.
Figure 26. The proposed antenna integrated with the drone: (a) power receiver probe around the transmitter drone in vertical and horizontal spaces, (b) overview of the simulated field test, and (c) horizontal assessment of the transmission and receiving power.
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Figure 27. The simulated air-to-ground and drone-to-drone transmission at different distances, azimuth, and elevation profiles (a) Co-polarization and (b) Cross-polarization.
Figure 27. The simulated air-to-ground and drone-to-drone transmission at different distances, azimuth, and elevation profiles (a) Co-polarization and (b) Cross-polarization.
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Table 1. Performance comparison of recent related state-of-the-art works for UAV communications at lower and higher bands.
Table 1. Performance comparison of recent related state-of-the-art works for UAV communications at lower and higher bands.
RefAntenna Type f r (GHz)Dims. (mm × mm × mm)Peak Gain (dBi)MIMOPlanar/NonplanarSingle/MultibandUAV Comm. Type
Antennas designed for UAV communications at different frequency bands
[11]Conical beam28.348 × 48 × 58.51noPlanarSingle bandmm-wave drone
[12]Reconfigurable 5.784 × 84 × 1310.2yesNonplanarSingle bandD2G
[13]Dipole1.25110 × 13 × 11.76noPlanar Single bandD2G
[14]Dipole2.4, 4.458 × 58 × 0.768.7yesPlanarDual bandD2G
[15]Dual-mode circular patch antenna 2.4, 5.2769.55noPlanarDual bandD2G
[16]Horn antenna 5.4>30016.5noNonplanar Single band D2G
[17]Dual-mode composite microstrip line5.8–8.5, 35–41.5>25020.6yesPlanar Dual band mm-wave
[18]Patch, magnetic–electric dipole2.40–2.48, 5.15–5.85, 57–64140 × 1409.8yesPlanar Triple bandmm-wave
Antennas utilizing MTS to enhance their performances.
Ref MTSs shape f r (GHz)Dims. (mm × mm × mm)Peak gain (dBi)BW (GHz)Planar/nonplanarSingle-/multibandApplications
[25]Cut circular rings 5.834.7 11.44–8.5Planar Dual bandLTE, Sub-6 GHz
[26]Slotted cutting-edge rectangular 5.580 × 80 × 1.55.665.25–6.35PlanarSingle bandSub-6 GHz
[27]Rectangular2765 × 55 × 1.311.123.89–31.23Planar Single band mm-wave, higher 5G
[28]Slotted mushroom 2837.7 × 37.7 × 1.719.224.4–30.5Planar Single bandmm-wave higher 5G
[29]Rectangular ring 2719.7 × 19.7 × 0.59.423.9–30.7PlanarSingle bandmm-wave, higher 5G
[30]Split-ring resonator2820.6 × 20.5 × 0.512.727.38–33.34PlanarSingle bandmm-wave, higher 5G
[32]Rectangular3412.5 × 12.511.330–38.58PlanarSingle band mm-wave
Table 2. The optimized parameters of the proposed antenna.
Table 2. The optimized parameters of the proposed antenna.
Parameters Values (mm)Parameters Values (mm)Parameters Values (mm)Parameters Values (mm)
L p 6 L 4 15 W f 2 0.75 g 3 0.12
L s 15 L 5 4.5 W f 3 1.25 g 4 0.12
L g 1.7 L 6 1.25 W g 1.75 g 5 0.12
L f 1 1.5 L 7 6.5 W 1 0.20 g 6 0.25
L f 2 1.25 L 8 2.5 g 7 0.125 g 8 0.40
L f 3 6.25 L 9 2 s 1 0.20 s 2 1.25
L 1 4.5 L 10 1.75 s 3 1.75 s 4 1.85
L 2 1.25 W 2 1.5 s 5 0.50 s 6 0.25
L 3 5.5 g 1 0.12a0.40 W s 16.5
W f 1 0.5 g 2 0.12b0.65
Table 3. Performance comparison between the proposed and recent similar reported antennas.
Table 3. Performance comparison between the proposed and recent similar reported antennas.
RefSize (mm2)PortsOperating
Frequency (GHz)		Max Gain (dBi)Radiation
Efficiency (%)		Coverage Isolation (dB)ECC
[10]114 × 11431.58/2.4/267.8/3.9/8.1-Upper/lower space<50-
[43]70 × 2932.4/5.2/609.8/7.9/8.4<90Half space<−25-
[44]196 × 1510.9154.5<78---
[45]30 × 3012.45/5.8/255.85----
[46]30 × 304286.192Half space<−29<0.16
[47]61.4 × 37.211.06/3.6<7.5-Half space--
[48]53 × 5342.5, 3.5, 5.5, 7.528<8.5<95-<20-
This work15 × 16.528.1–24.7/
66–68/74–78/80–140 		<14<90Upper/lower space<35<0.003
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Saeidi, T.; Saleh, S.; Timmons, N.; Al-Gburi, A.J.A.; Karamzadeh, S.; Althuwayb, A.A.; Rashid, N.; Kaaniche, K.; Ben Atitallah, A.; Elhamrawy, O.I. Meta Surface-Based Multiband MIMO Antenna for UAV Communications at mm-Wave and Sub-THz Bands. Drones 2024, 8, 403. https://doi.org/10.3390/drones8080403

AMA Style

Saeidi T, Saleh S, Timmons N, Al-Gburi AJA, Karamzadeh S, Althuwayb AA, Rashid N, Kaaniche K, Ben Atitallah A, Elhamrawy OI. Meta Surface-Based Multiband MIMO Antenna for UAV Communications at mm-Wave and Sub-THz Bands. Drones. 2024; 8(8):403. https://doi.org/10.3390/drones8080403

Chicago/Turabian Style

Saeidi, Tale, Sahar Saleh, Nick Timmons, Ahmed Jamal Abdullah Al-Gburi, Saeid Karamzadeh, Ayman A. Althuwayb, Nasr Rashid, Khaled Kaaniche, Ahmed Ben Atitallah, and Osama I. Elhamrawy. 2024. "Meta Surface-Based Multiband MIMO Antenna for UAV Communications at mm-Wave and Sub-THz Bands" Drones 8, no. 8: 403. https://doi.org/10.3390/drones8080403

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