RU2467346C1 - Method of calibrating active phased antenna array - Google Patents

Abstract

FIELD: physics, radio.

SUBSTANCE: invention relates to antenna equipment and is meant for calibrating active phased antenna arrays. The method of calibrating an active phased antenna array, in which in order to calibrate the receiving part of the transmit-receive channel, a control signal is transmitted to the input of the receiving part of each channel; parameters of the received signal are measured; adjusting coefficients which are used to adjust parameters of the receiving part of the channel are formed based on the measurements; to calibrate the transmitting part of the transmit-receive channel, a control signal is transmitted to the input of the transmitting part of each channel; parameters of the transmitted signal are measured; adjusting coefficients which are then used to adjust parameters of the transmitting channel are formed based on the measurements; calibration of the receiving part of the transmit-receive channels is carried out in pairs in receive mode, wherein the control signal is picked up from the output of the receiving part of the transmit-receive channels; calibration of the transmitting part of the transmit-receive channels is carried out in pairs in transmit mode, wherein part of the signal power tapped from the output of the corresponding transmit-receive channel and passing through the receiving part of that channel is used for calibration; during calibration, phase shift and amplitude difference of the signal from the output of the calibrated channel relative the reference channel are determined; the same reference channel is used for calibrating all channels.

EFFECT: wider field of application and high accuracy of calibration.

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Description

The invention relates to the field of phased antenna arrays (PAR), providing the formation of a radiation pattern (NF) of a given shape, electronically variable in space, and used in communication systems and radar technology.

Currently, active phased arrays (AFARs) are widely used, in which a transmitter and a receiver are connected to each emitter (antenna element) through decoupling devices. In the AFAR in the transmission mode, the emitted signal is transmitted through a multichannel transceiver path, each channel of which includes a receiving and transmitting part, while the transmitting part includes such nodes as a power divider, phase shifter, power amplifier, matching circuits, emitter, to the receiving the part includes such nodes as: low-noise amplifier (LNA), phase shifter, matching circuits, etc. As a decoupling device, a circulator is used.

The elements that make up each node are not completely identical. They have deviations of parameters from the nominal value caused by manufacturing errors, temperature effects, aging, etc. As a result, the amplitudes and phases of the signals at the outputs of different channels of the multichannel system will differ from the calculated values, this causes errors in the amplitude-phase distribution along the AFAR aperture relative to the calculated values and, ultimately, causes the deterioration of such important parameters as directional coefficient (LPC) ), the efficiency coefficient (COP) of the AFAR and the level of the side lobes (UBL). For example, the level of side lobes, depending on the amplitude and phase errors in the channels of the AFAR, can increase by tens of dB [Design of phased antenna arrays. Ed. D.I. Voskresensky. M .: Radio engineering, 2003, pp. 466-447].

To maintain the calculated parameters of the AFAR during operation, it is necessary to periodically monitor the parameters of the multi-channel transmit-receive path of the AFAR and calibrate it to restore their identity of all channels.

A known method of monitoring the diagram parameters of the antenna of the direction finder, which consists in the fact that they emit an auxiliary antenna signal, take it with a controlled antenna, measure the signal power at the output of a controlled antenna and compare it with a given value [USSR Author's Certificate No. 179800, Cl. G01R 29/10].

The main disadvantage of this method is the need to use an auxiliary antenna, spaced from the controlled AFAR to a distance of the far zone, which is not always possible when installing the AFAR on moving media, such as an airplane or a car.

There is a method of integrated monitoring of the characteristics of a phased array, described in [Bubnov GG and other Switching method for measuring the characteristics of the PAR. M .: Radio and communication. 1988], based on the serial switching of the phase shifters and the reception of the emitted signal by an auxiliary antenna. The main disadvantage of this method is the need to use an auxiliary antenna, spaced from the controlled AFAR to a distance of the far zone, which is not always possible when installing the AFAR on moving media, such as an airplane or a car. The method of integrated monitoring of the characteristics of a phased antenna array described in [RF Patent 2169376, MKI G01R 29/10], which also uses serial switching of phase shifters and receiving the emitted signal by an auxiliary antenna, also has the same drawback.

A known method of calibrating an antenna array [RF Patent 2147753, Cl. G01S 7/40, 04/20/2000], closest to the proposed method and adopted as a prototype, in which, for calibration of the transmitting path, an input signal is fed alternately to each channel of the transmitting path of the PAR, the signal transmitted by each channel of the transmitting path is measured, and correction coefficients are generated for each channel of the transmitting path based on the measurement results, correction factors are used to adjust the phases and / or amplitudes of the transmitted signals.

To calibrate the receive path, an input signal is fed alternately to each channel of the receive path of the PAR, measure the signal received by each channel of the receive path, generate correction coefficients for each channel of the receive path based on the measurement results, correction coefficients are used to adjust the phases and / or amplitudes of the receive signals.

The disadvantages of the prototype are:

1) the limited scope of the prototype only for use in AFAR with separate receiving and transmitting paths: the prototype considers separate receiving and transmitting antenna arrays. Separate receiving and transmitting paths implies different frequency ranges of transmitted signals and received signals, therefore, for calibration of the prototype, it is necessary to use signals with different frequencies. The prototype cannot be used for AFAR with transceiver channels, each of which includes transmitting and receiving parts and operates on its own emitter.

2) The use of a measuring device of excessive complexity to ensure high accuracy of measurements, since the prototype uses measurement in the process of calibration of the absolute parameters of the signal at the output of one calibrated channel. In accordance with the formula and description of the device that implements the prototype, the measuring receiver has no connections with other devices of the prototype, it receives only the measured signal. Thus, the measuring receiver must have a built-in highly stable reference signal source, using which the changes in the parameters of the received signal in time will be measured. The same highly stable reference signal source must be present in the prototype pilot signal driver. Let us estimate the permissible instability of these two sources of reference signals when measuring the phase of the signal, for example, at a frequency fc = 1800 MHz.

, и в соответствии с [Аппаратура для частотных и временных измерений. Под. ред. А.П.Горшкова. М.: Сов. радио, 1971, стр.225] можем найти допустимую относительную нестабильность частоты за время измерений

. Если требуются более высокая точность измерения сдвига фаз, например, 3°, то

требования к стабильности встроенных генераторов будут еще выше, а именно

. Такие высокие требования к допустимой нестабильности частоты можно обеспечить только применением дорогих, сложных и имеющих большие габаритные размеры стандартов частоты - рубидиевых или цезиевых.If you set the permissible phase shift between these signals during the calibration of all channels Δt = 1c not more than 10 ° due to frequency instability, then this corresponds to a time delay

, and in accordance with [Equipment for frequency and time measurements. Under. ed. A.P. Gorshkova. M .: Sov. radio, 1971, p. 225] we can find the permissible relative frequency instability during the measurement

. If higher accuracy in phase shift measurements, e.g. 3 °, is required, then

stability requirements for built-in generators will be even higher, namely

. Such high requirements for permissible frequency instability can be achieved only by the use of expensive, complex and large-sized frequency standards - rubidium or cesium.

Thus, the method adopted in the prototype for measuring a single signal significantly complicates and increases the cost of the prototype.

3) Use for control of an arbitrary waveform, which reduces the measurement accuracy compared to using a sinusoidal signal. The arbitrary waveform spectrum is always enriched with harmonic components. In this case, the presence of harmonics causes an increase in the measurement error. So, in accordance with [Glinchenko A.S., Kuznetskiy S.S. et al. Digital methods for measuring phase shift. Novosibirsk .: Nauka, 1979, pp. 53-59] the presence of harmonics in the signal causes significant phase measurement errors, the magnitude of which reaches tens of degrees.

The aim of the invention is to expand the scope and increase the accuracy of calibration.

To achieve this goal, a method for calibrating the AFAR is proposed, in which to calibrate the receiving part of the transmitting and transmitting channel, a control signal is supplied to the input of the receiving part of each channel, the parameters of the received signal are measured, and correction factors are generated based on the measurements, which are then used to adjust the parameters of the channel receiving part , for calibration of the transmitting part of the transceiver channel, a control signal is supplied to the input of the transmitting part of each channel, the parameters of the transmitted signal are measured Nala, is formed on the basis of measurements of correction coefficients, which are then used to adjust the parameters of the channel transmission path.

According to the invention, the calibration of the receiving part of the transmitting and transmitting channels is performed in pairs in the receiving mode, while the control signal is removed from the output of the receiving part of the receiving and transmitting channels, the calibration of the transmitting part of the transmitting and transmitting channels is made in pairs in the transmission mode, while part of the power is used for calibration of a signal branched from the output of the corresponding transceiver channel and passed through the receiving part of this channel, the phase shift and the difference of signal amplitudes with Exit calibrated relative to the reference channel as a reference to calibrate all channels using the same channel, for example the first, is used to calibrate the sinusoidal signal with a frequency in the operating frequency range APAA.

A comparative analysis of the claimed method and prototype shows that the claimed method is characterized in that:

- in the prototype, the absolute values of the phase and amplitudes of the signals at the channel outputs are measured, in the proposed method, the conditions for measuring the parameters of the calibrated channel are changed - not the absolute values of the parameters of the control signal are measured, but the difference parameters (phase and amplitude difference) of the signals at the output of two channels - calibrated and reference, which significantly reduces the cost of measuring equipment and increases the accuracy of measurements;

- the calibration sequence of the transmitting and receiving parts of the channels has been changed: if the prototype first calibrates the transmitting, and then the receiving parts of the AFAR channels, then in the proposed method, the receiving part is calibrated first, and then the transmitting part. This is due to the fact that in the proposed method, the receiving part is used in the process of calibrating the transmitting part, therefore, it must be calibrated first;

- the prototype uses an arbitrary waveform, in the proposed method uses a sinusoidal signal, which increases the accuracy of the calibration compared to the protopope due to the fact that eliminates the influence of frequency instability and the shape of the control signal on the measurement accuracy;

- in the proposed method, the calibration of the receiving part of the transceiver channel is performed in the reception mode, and the calibration of the transmitting part of the transceiver channel is performed in the transmission mode.

The combination of distinguishing features and properties of the proposed method from the literature are unknown, therefore, it meets the criteria of novelty and inventive step.

Figure 1 shows the structural diagram of a device that implements the calibration of AFAR in accordance with the proposed method;

figure 2 shows the structural diagram of the control unit (BU);

figure 3 shows the structural diagram of the meter;

figure 4 shows the structural diagram of the connection of the AFAR channels to the CAN bus;

figure 5 shows the structural diagram of the algorithm of the device.

The proposed method performs the following sequence of operations:

- for calibration of the receiving part, the control signal is supplied in the receiving mode to the input of the receiving part of each channel;

- compare in turn the parameters of the signal from the output of all monitored channels with the signal of the reference channel, while the phase difference and amplitudes are measured, correction coefficients for each channel are generated based on the measurements, which are then used to adjust the parameters of the receiving part of the transceiver channel;

- for calibration of the transmitting part of the channel, a control signal is supplied to the input of the transmitting part of each channel in the transmission mode;

- they compare in turn the signal parameters from the output of all monitored channels with the signal of the reference channel, while the phase and amplitude differences are measured, and correction coefficients for each channel are generated based on the measurements, which are then used to adjust the parameters of the transmitting part of the transmitting and transmitting channel.

A device that implements the proposed method for calibrating the PAR, contains (Fig. 1) an N-channel transceiver path, each channel of which contains (for example, Fig. 1 shows the contents of the first channel) a power divider 1 connected in series, the input of which is the channel input, first phase shifter 2, first attenuator 3, power amplifier 4, circulator 5, directional coupler (HO) 6 and emitter 7. The second output of the circulator 5 is connected to a low-noise amplifier in series with a protective device (LNA) 8, the first to the mutator 9, the second attenuator 10, the second phase shifter 11, the output of which is the output of the channel. The control bidirectional output HO 6 is connected to the first bi-directional input of the second switch 12, the second input and output of which are connected respectively to the second output of the power divider 1 and the second input of the first switch 9.

The first, second, third, fourth, fifth, sixth and seventh outputs of the controllers 13 in each channel are connected to the control inputs of the second phase shifter 11, the second attenuator 10, the first switch 9, the second switch 12, the power amplifier 4, the first attenuator 3 and the first phase shifter 2 respectively. The input and output of the control interface of the controller 13 are connected via a transceiver 14 to the AFAR control line.

The first and second outputs, as well as the control input of the switch 15 are connected to the first and second measuring inputs and the first control output of the control unit 16, and the inputs of the switch 15 from the first to N are connected to the outputs of the second phase shifters 11, which are outputs from the first to N channels, respectively .

The second output of the control unit 16 is connected to the inputs of the transceivers 14 of each channel, the inputs of which are the control inputs of each channel AFAR. The third output of the control unit 16 is connected to the input of the power divider 17.

The outputs of the power divider 17 from the first to N are connected to the input of the power divider 1 of each channel, the input of the power divider 1 is the input in each channel of the AFAR.

As part of a device that implements the proposed calibration method, only blocks are shown that ensure the operation of the described algorithm.

One of the embodiments of the control unit 16 contains (Fig. 2) a meter 18, the first and second inputs of which are the measuring inputs of the control unit 16, and the three outputs are connected to the measuring inputs of the controller 19, the first output of which is connected to the control signal generator 20, the output of which is the third output of the control unit 16. The input and output of the control interface of the controller 19 is connected via a transceiver 21 to the control line, which is the second output of the control unit 16, the third output of the controller 19 is the first control output of the control unit 16.

Controllers 13 and 19 can be made on the basis of a microcontroller type M16C [Single-chip 16-bit CMOS microcomputer M16C / 6N Group. Users manual. Mitsubishi Electric. 1999], which contains in one chip 26 measuring inputs, switched to the built-in 10-bit analog-to-digital converter (ADC), two outputs of the built-in 8-bit digital-to-analog converter (DAC), 10 built-in parallel I / O ports, RAM, ROM, integrated processor, two integrated serial CAN interfaces.

The AFAR channel control highway is a serial data bus to which all control channel inputs are connected in parallel. This can be, for example, a two-wire CAN bus, the beginning and end of which are loaded with 120 Ohm loads. The CAN protocol is described, for example, in [CAN specification. Version 2.0., Robert Bosch GmbH, 1991; Kustarev P. CAN protocol in microcontrollers. Electronic components. 2002. No. 5].

The circuit for connecting channels to the CAN bus is shown in Fig. 4 in accordance with [MAX3057. ± 80V Fault-Protected, 2Mbps, Low Supply Current CAN Transceivers // Maxim Integrated Products. 2002]. In controllers 13 and 19, a built-in CAN interface is used for data transfer.

Transceiver 21 can be made on the basis of the MAX3050 microcircuit [Website of Maxim Integrated Products Inc. http://www.maxim-ic.com], which ensures the operation of controllers 13 and 19 along the CAN bus, through which control signals are transmitted to all transceiver channels.

The meter 18 contains (Fig. 3) a first 22 and a second 23 amplitude detector, the inputs of which are the first and second inputs of the meter, and the outputs are connected to the inputs of the first 24 and second 25 low-pass filters (LPFs), respectively. The outputs of the first 24 and second 25 low-pass filters are the first and second outputs of the meter, respectively. The first and second inputs of the phase detector (PD) 26 are combined with the inputs of the first 22 and second 23 amplitude detectors, and the output is connected to the input of the third low-pass filter 27, the output of which is the third output of the meter 18.

The first 22 and second 23 amplitude detectors can be made on the basis of detectors type ZX47 manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The first 24, second 25 and third 27 low-pass filters can be performed based on SCLF filters manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The phase detector 26 can be made on the basis of phase detectors type MPD manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The devices in each i-th receiving and transmitting channel of the HEADLIGHTS are controlled by the i-th controller 13: the first switch 9, the second switch 12, the power amplifier 4, the first 3 and the second 10 attenuators are carried out by parallel codes. The first 2 and second 11 phase shifters are controlled by an analog signal from the outputs of the DACs built into the controller 13. Communication with the control unit 16 is carried out on a serial CAN bus through the transceiver 14.

The power divider 17 provides the separation of the control signal from the output of the control unit 16 into N outputs. It can be performed, for example, on power dividers manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007]. Since the company produces dividers with a different number of channels from 2 to 48, the specific types of elements used are determined by the number of channels N in the AFAR. If N exceeds 48, then it is necessary to use a serial branching of the input signal to achieve the required number of channels. The implementation of multi-channel DM on 64 channels is also described in [Design of phased antenna arrays. Ed. D.I. Voskresensky. M .: Radio engineering. 2003, pp. 550-551].

The switch 15 provides switching signals from N inputs to two outputs. It can be performed, for example, on switches manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007]. The company produces switches with a different number of channels from 2 × 1 to 4 × 1, so the N × 2 switch will contain 2 power dividers and switches. The specific types of elements are determined by the number of channels N in the AFAR. With the number of channels N = 4, the switch 15 can be implemented on a Hittite chip HMC596LP4 [Hittite Microwave 2008 Designer’s Guide. 2008. Vol. 1-3].

The directional coupler 6 provides branching of a small part of the signal power by 40-60 dB less than the main signal and can be performed, for example, on the basis of a directional coupler of the BDCA series and attenuator of the PAT series manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The first 9 and second 12 switches are designed to connect HO 6 to the input of the second attenuator 10 or to the output of the power divider 1. The switches can be made on the basis of PSW or HSWA series switches manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The first 2 and second 11 phase shifters and power divider 1 can be made on the basis of JSPHS series phase shifters and LRPS power dividers manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007], or based on strip phase shifters [Design of phased array antennas. Ed. D.I. Voskresensky. M .: Radio engineering. 2003, p. 532-536].

The power amplifier 4 can be performed on amplifiers of the ZHL series manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The control signal generator 20 can be made on the basis of a frequency synthesizer that generates a given frequency of the control signal of the KSN series [Mini-Circuits. IF / RF Components Guide. 2007].

LNA 8 can be made in the form of series-connected protection devices based on the PLS series limiter manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007] or executed according to the scheme given in [Gassanov L.G., Lipatov A.A. and other solid-state microwave devices in communication technology. M.: Radio and communications, 1988, p.143], and low-noise amplifier circuits, for example, from the [Hittite Microwave 2008 Designer’s Guide. 2008. Vol. 1-3].

The circulator 5 can be performed on the basis of the circuit shown in [Gassanov L.G., Lipatov A.A. and other solid-state microwave devices in communication technology. M .: Radio and communications, 1988, pp. 44-45].

The first 3 and second 10 attenuators can be made on the basis of digital attenuators of the DAT series manufactured by Mini-Circuits [Mini-Circuits. IF / RF Components Guide. 2007].

The emitter 7 can be implemented, for example, on the basis of a symmetric vibrator described in [Design of phased antenna arrays. Ed. D.I. Voskresensky. M .: Radio engineering. 2003, p. 532-536].

A device that implements the calibration method AFAR, works as follows.

Before calibration, a phase shift of 0 ° is set in all the channels in the first 2 and second 11 phase shifters, 0 dB attenuation is set in the first 3 and second 10 attenuators. Calibration of the transceiver path begins with the calibration of the receiving part. By default, the reference channel is the first. BU 16 installs the first 9 and second 12 switches in the reference and calibrated channels so that the second output of the power divider 1 is connected to the control output HO 6. In BU 16, the controller 19 includes a control signal generator 20, which generates a control signal in the form of a sinusoidal signal with a fixed frequency within the operating frequency range controlled by the AFAR. The control signal is supplied to the power divider 17, where it branches into N outputs and is fed to the inputs of the transceiver channels of the AFAR.

In the reference and calibrated i-th channel, the control signal through the power divider 1, the second switch 12, NO 6 and the circulator 5 is fed to the input of the LNA 8, passes through the first switch 9, the second attenuator 10, the second phase shifter 11, and fed through the switch 15 to the measuring inputs BU 16. During the passage of the signal through the receiving part, it acquires a phase shift Δφ _PFP i and changes in amplitude by ΔA _PFP i.

In BU 16, the amplitude of the control signal is measured in the reference channel A of the _PFP 1 using the first amplitude detector 22, the first low-pass filter 24 and the ADC built into the controller 19. The amplitude of the control signal in the calibrated channel A of the _PFP i is measured using the second amplitude detector 23, the second low-pass filter 25 and the ADC built into the controller 19. In the controller 19, the amplitude difference ΔA _PFP i = A _PFP i-A _PFP 0 is determined and rounded to the value of the tuning step of the second attenuator 10.

The phase shift Δφ of the _PFP i signals at the output of the calibrated and reference channels is measured in the control unit 16 using the PD 26, the third low-pass filter 27 and the ADC built into the controller 19. The measured value is rounded up to the tuning step of the second phase shifter 11.

The corresponding attenuation values in the second attenuator 10 and the phase shift in the second phase shifter 11 are set.

Next, the transmitting part of the i-th channel is calibrated.

The first 9 and second 12 commutators in the reference and calibrated channels are installed so that the control output НО 6 is connected to the input of the second attenuator 10. БУ 16 includes the control signal for the measurement time in the GCS 20 and УМ 4 in the calibrated channel. The control signal in the power splitter 17 branches into N channels and is fed to the input of the power splitter 1. Passing through the first phase shifter 2, the first attenuator 3, UM 4, circulator 5, HO 6, the second 12 and first 9 switches, the second attenuator 10 and the second phase shifter 11, the signal enters through the switch 15 to the measuring inputs of the control unit 16.

Since the channel receiving part was first calibrated, during the passage of the signal through the transmitting and receiving part, it acquires a phase shift Δφ of the _PRD i and changes in amplitude by ΔА _PRD i, which are determined only by the parameters of the transmitting part of the channel.

In BU 16, the amplitude of the control signal is measured in the reference channel A of the _Tx 1 using the first amplitude detector 22, the first low-pass filter 24 and the ADC built into the controller 19. The amplitude of the control signal in the calibrated channel A of the _Tx i is measured using the second amplitude detector 23, a second LPF 25 and ADC built into the controller 19. Further, the difference amplitude ΔA i = a _{TX TX TX} i-1 and a is rounded up to the value of the first adjustment step attenuator 3.

The phase shift Δφ of the _{PFD of the} i signals at the output of the calibrated and reference channels is measured in the control unit 16 using the PD 26, the third low-pass filter 27 and the ADC built into the controller 19. The measured value is rounded to the tuning step of the first phase shifter 2.

The corresponding attenuation values in the first attenuator 3 and the phase shift in the first phase shifter 2 are set.

The algorithm of the device shown in Fig.5. The control and data from the controller 19 of the control unit 16 are transmitted to the controllers 13 through transceivers 14 of the transceiver channels on the CAN serial line with serial codes. In addition, each controller has its own address in accordance with the channel number.

After calibration of the i-th channel, the i + 1-th channel is calibrated and so on to N. Thus, all channels are calibrated, and the phase shift and amplitude difference between them will be set with the accuracy of the attenuation adjustment discret in the attenuators 3.10 and the adjustment discret phase shift in phase shifters 2 and 11.

Calibration should be carried out with a repetition period Tk, which is determined by the time the parameters of the AFAR devices change. The value of Tk depends on the element base used, circuit design, etc. and is determined on the prototype of the product.

The algorithm of the device that implements the proposed method (figure 5), includes the following sequence of operations:

1) set the parameters of the devices to their initial state: attenuators 3 and 10 at 0 dB, phase shifters 2 and 11 at 0 °, switch 12 to the position that connects the control output of the directional coupler (BUT) 6 to the output of the power divider 1, switch 9 to , providing the connection of the output of the LNA 8 with the input of the attenuator 10, the switch 15 to the position, providing the connection of the outputs of channels 1 and i with the measuring inputs of the control unit 16, where i = 2;

2) turn on the GCS 20 from the BU 16, while the control signal through the power divider 17 goes to the inputs of all channels, in each channel it passes through the power divider 1, switch 12, NO 6 and circulator 5, and is fed to the input of the LNA 8. With input LNA 8 control signal through the switch 9, the attenuator 10, the phase shifter 11 and the switch 15 is fed to the meter 18. Measure the amplitudes A _PFP i and A _PFP 1, find the difference ΔA _PFP i = A _PFP i-A _PFP 1, round it to step attenuator changes 10;

3) measure the phase shift between two signals Δφ _PRM i, turn off the GCS 20;

4) set the above parameters of the attenuator 10 and the phase shifter 11;

5) set the switch 12 to the position that provides the connection of the control output HO 6 with the input of the switch 9, switch the switch 9 to the position that connects the output of the switch 12 to the input of the attenuator 10;

6) include GCS 20 and UM 4;

7) measuring the amplitude A i and A _TX 1 _TX, find the difference ΔA i = A _{TX TX TX} A i-1, it shall be rounded up to step 3 changes attenuator;

8) measure the phase shift between the two signals Δφ _PRD i, turn off the PA 4 and GKS 20;

9) set the above parameters of the attenuator 3 and phase shifter 2;

10) repeat the above operations 2) ... 9) for all calibrated channels (i = 3 ... N, where N is the number of PAR lights).

As a result of using the proposed solution, the following technical and economic effect was obtained: the requirements for the stability of the control signal generator 20 were reduced and the calibration accuracy was improved.

Find the accuracy of measurement of the prototype for the usual relative instability of crystal oscillators δυ = 5 · 10 ^-6 . Provided the measurement of all the channels for Δt = 1c instability of such a signal delay at the beginning and end of the measurements will be δT = δυ · Δt ^2/16 = 3.1 × 10 ^-7 c, at that frequency, for example, fc = 1800 MHz is constitute the phase incursion δφ = δT · 360 ° · f _C = 558 °. Thus, the accuracy of measuring the phase shift of channels measured at the beginning and end of calibration will be very low.

Since the stability of the process control signal is important only for the time measurement data of one channel Δt ₁ = 0,001 c, which is considerably less than the total measurement time, the instability of such a signal delay will be δT = δυ · Δt ^2/16 = 3.1 × 10 ^{- 13} , which at a frequency, for example, fc = 1800 MHz, will be the phase offset δφ = δT · 360 ° · f _C = 0.2 °.

The proposed method has a more extended area and can be used, in contrast to the prototype, in transceiver AFAR.

Claims (1)

A method for calibrating an active phased array antenna, in which, for calibrating the receiving part of the transceiving channel, a control signal is supplied to the input of the receiving part of each channel, the parameters of the received signal are measured, correction factors are generated based on the measurements, which are then used to adjust the parameters of the channel receiving part, for calibrating the transmitting parts of the transceiver channel provide a control signal to the input of the transmitting part of each channel, measure the parameters of the transmitted signal, form based on the measurements, correction coefficients are used, which are then used to adjust the parameters of the transmitting channel channel, characterized in that the receiving part of the transceiving channels is calibrated in pairs in the receiving mode, while the control signal is removed from the output of the receiving part of the transceiving channels, and the calibration of the transmitting part of the transceiving channels is performed in pairs in the transmission mode, while for calibration use part of the signal power, branched from the output of the corresponding receiver During the calibration, the phase shift and the difference in the amplitudes of the signal from the output of the channel being calibrated relative to the reference channel are measured during calibration; the same channel is used as reference channel for calibration of all channels, for example, the first channel; a sinusoidal signal is used for calibration with a frequency in the operating frequency range of the active phased array antenna.

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