JP6437706B2 - IPM type electric rotating machine - Google Patents

Description

  The present invention relates to an IPM type electric rotating machine, and more particularly to an apparatus that realizes highly efficient rotational driving.

Electric rotating machines mounted on various devices are required to have characteristics corresponding to the mounted devices.
For example, in the case of a drive motor mounted on a hybrid vehicle (HEV: Hybrid Electric Vehicle) together with an internal combustion engine as a drive source, or mounted on an electric vehicle (EV: Electric Vehicle) as a single drive source, the motor rotates at a low speed. It is required to have a wide variable speed characteristic at the same time as generating a large torque in the region.

This type of vehicle is required to improve the energy conversion efficiency of each component, including the electric rotating machine, in order to improve the fuel efficiency. In particular, the in-vehicle electric rotating machine is expected to improve the efficiency in the normal range. It is rare. Furthermore, in-vehicle electric rotating machines are required to have a smaller and higher energy density structure from the viewpoints of installation space restrictions and weight reduction.
By the way, in HEV and EV, generally, a low-speed rotation / low load region of an electric rotating machine is a regular region. For this reason, the ratio of contribution to the torque of the in-vehicle electric rotating machine is larger for the magnet torque than for the reluctance torque according to the magnitude of the armature current, and many high-magnetism permanent magnets are used for higher efficiency. Tend to use.
Because of this tendency, for electric rotating machines, neodymium magnets with a high residual magnetic flux density are placed inside the rotor core in order to improve the energy conversion efficiency, especially in the normal range of low-speed rotation and low load range. An IPM (Interior Permanent Magnet) type which is an embedded permanent magnet type synchronous motor is frequently used. In this IPM type electric rotating machine, a permanent magnet is embedded in the rotor so as to have a V-shape that opens toward the outer peripheral surface, thereby forming a magnetic circuit that can actively use reluctance torque in addition to magnet torque. (For example, Patent Documents 1 and 2). In addition, in the IPM type electric rotating machine, it has also been proposed to form a groove for adjusting the magnetic resistance between the outer peripheral surface of the rotor and the stator side (for example, Patent Documents 3 to 5).

JP 2006-254629 A JP 2012-39775 A Japanese Patent Laid-Open No. 2004-328956 JP 2008-206308 A JP 2008-31316 A

By the way, in recent electric rotating machines, permanent magnets containing rare earths such as Nd, Dy, Tb are often used to increase the magnetic force and heat resistance. Due to the instability, there is a growing need for higher efficiency while reducing the amount of rare earth used.
However, in HEV and EV, the normal area of the electric rotating machine is a low-speed rotation / low load area. Therefore, in order to increase the magnet torque contributing to the area, the IPM type as described in Patent Documents 1 to 5 is used. In motors as well, there is a tendency to increase the usage of permanent magnets with high magnetic force. This is a direction that hinders the solution of the problem of reducing the amount of rare earth used.
Further, in the IPM type electric rotating machine, as described in Patent Documents 3 to 5, even if the magnetic resistance adjusting groove formed on the outer peripheral surface of the rotor is applied as it is, torque ripple or the like cannot be effectively suppressed. .
Therefore, an object of the present invention is to provide a low-cost and high-energy density electric rotating machine that realizes highly efficient rotational driving while reducing the amount of permanent magnets used.

A first aspect of the invention relating to an IPM type electric rotating machine that solves the above problem is that a coil is formed in a slot formed between a rotor in which a plurality of pairs of permanent magnets are embedded and a plurality of teeth facing the rotor. A pair of permanent magnets, wherein the pair of permanent magnets are arranged in a V-shape that opens toward an outer peripheral surface of the rotor, and the pair of permanent magnets is formed. each magnetic pole, fluxes barrier is formed, the flux barrier is formed between the pair of permanent magnets, and the stator side of the stator magnetic pole surface facing the stator of the permanent magnet Is formed by a space formed on the inner side in the radial direction of the rotor with respect to the end of the rotor, and the space is more than the anti-stator side magnetic pole surface opposite to the stator side magnetic pole surface. Extending radially inward Cage, the adjustment groove on the d-axis of the outer circumferential surface of the rotor is provided outside of the number of poles the rotor P, and the length of the permanent magnet W pm, to the shaft center said outer circumferential surface of the rotor Assuming that the radius is R1, the length from the rotor axis to the groove bottom of the adjustment groove is R4, a relationship of 1.38 <(P × Wpm) / R1 <1.75, and 0.98 ≦ R4 / It is formed in a size and shape satisfying the relationship of R1 <1.0, and the width in the circumferential direction of the adjustment groove is equal to or greater than the width in the circumferential direction of the facing surface of the teeth facing the rotor. The width is set to be equal to or smaller than the width between adjacent teeth in the circumferential direction .

According to a second aspect of the invention relating to the IPM type electric rotating machine that solves the above problem, in addition to the specific matter of the first aspect, the adjustment groove is formed on the rotor centered on the axis of the rotor. The outer opening angle on the outer peripheral surface is θa, the inner opening angle of the groove bottom around the axis of the rotor is θb, the opening width of the slot on the outer peripheral surface side of the rotor is SO, and the rotation of the teeth Outer angle 1 ≦ θa (electrical angle), where TB is the width of the outer peripheral surface of the child, TB is the width of the tip of the teeth inside the width TB of the teeth, and TG is the air gap width between the rotor and the teeth. ) ≦ outer angle 2, inner angle 1 ≦ θb (electrical angle) ≦ inner angle 2, outer angle 1 = 2 × tan ⁻¹ ((TB / 2) / (R1 + AG)), outer angle 2 = inner angle 2 = 2 × ^{tan -1 ((SO + (TB} / 2)) / (R1 + AG)), an interior angle 1 = 0 And it is characterized in that formed in size and shape to meet the TW ≦ TB, the relationship.

As described above, according to the first aspect of the present invention, the permanent magnet in the range that generates the magnet magnetic flux in the direction of canceling the armature magnetic flux on the d-axis side is replaced with a gap having a small magnetic permeability. The magnet magnetic flux and the armature magnetic flux do not interfere (cancel) on the d-axis side, and the armature magnetic flux can be restricted from passing through the range. Therefore, the magnet magnetic flux that wastes the armature magnetic flux on the d-axis side can be eliminated, the reluctance torque can be effectively used together with the magnet torque, and the permanent magnet itself can be used while obtaining the torque more than before the replacement of the d-axis side permanent magnet. The amount can be reduced.
Furthermore, by replacing the permanent magnet with a gap, the magnet magnetic flux can be reduced, the induced voltage constant on the high speed rotation side can be reduced, and the output on the high speed rotation side can be improved. Further, the weight can be reduced and the inertia can be reduced.
Further, by reducing the magnetic flux of the magnet, the field weakening region can be reduced (the amount of field weakening can be reduced), and the spatial harmonics that cause magnetostriction can be reduced. For this reason, generation | occurrence | production of an eddy current in a permanent magnet can be restrict | limited, heat_generation | fever can be suppressed, demagnetization by the temperature change of a permanent magnet can be suppressed, a heat-resistant grade can be lowered | hung and cost can be reduced.

In addition, the gap is formed so that the space extending to the d-axis side expands toward the axial center side of the rotor, so that the armature enters the rotor from the q-axis side on one side of the magnetic pole. The magnetic flux can be detoured so as to go to the q axis side on the other side by restricting the magnetic flux from going to the outer peripheral surface side of the permanent magnet, and becomes saturated together with the magnetic flux toward the outer peripheral surface side of the permanent magnet. You can avoid that. Therefore, the reluctance torque due to the armature magnetic flux can be used more effectively, and the total torque can be increased.
Further, the adjustment groove can be adjusted so as to increase the magnetic resistance in the vicinity of the d-axis between the rotor and the stator side teeth, and the magnetic flux in the vicinity of the d-axis is reduced by forming the gap. As a result, an increase in interlinked armature magnetic flux can be suppressed. Therefore, it is possible to prevent the drive efficiency from being lowered due to an increase in torque ripple or iron loss.
As a result, it is possible to realize a low-cost electric rotating machine that rotates with high energy density and high quality. Further, the alignment can be reliably performed using the adjustment groove as a mark, and the assembly can be facilitated.

According to the first aspect of the present invention, the adjustment groove satisfies a relationship of 0.98 ≦ length R4 from the rotor axis to the groove bottom / rotor outer radius R1 <1.0. Thus, harmonic ripple can be suppressed and torque ripple can be reduced.
According to the second aspect of the present invention, further, the size and shape of the adjustment groove is 2 × tan ⁻¹ ((tooth-to-face width TB / 2) / (rotor outer radius R1 + air gap width AG)). ≦ Outer opening angle θa (electrical angle) at the center of axis ≦ 2 × tan ⁻¹ ((slot opening width SO + (tooth facing width TB / 2)) / (rotor outer radius R1 + air gap width AG)), 0 ° ≦ Inner opening angle θb (electrical angle) ≦ 2 × tan ⁻¹ ((slot opening width SO + (tooth facing width TB / 2)) / (rotor outer radius R1 + air gap width AG))), tooth tip By forming so as to satisfy the relationship of the part width TW ≦ the tooth-to-face width TB, the harmonic torque can be effectively suppressed at both low load and maximum load, and torque ripple can be reduced.

FIG. 1 is a diagram showing an embodiment of an IPM type electric rotating machine (motor) according to the present invention, and is a plan view showing a schematic overall configuration thereof. FIG. 2 is a magnetic flux diagram of the armature magnetic flux at the time of low load driving in the structure of the embodiment. FIG. 3 is a magnetic flux diagram of magnet magnetic flux at the time of low load driving in the structure of the embodiment. FIG. 4 is a graph showing torque characteristics with respect to the current phase of a V-shaped IPM motor having no large gap on the d-axis side. FIG. 5A is a magnetic flux diagram of magnet magnetic flux of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 5B is a vector diagram of the magnetic flux in the vicinity of the d-axis of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 6A is a magnetic flux diagram of armature magnetic flux at the time of maximum load driving of a V-shaped IPM motor without a large gap on the d-axis side. FIG. 6B is a vector diagram of armature magnetic flux in the vicinity of the d-axis at the time of maximum load driving of the V-shaped IPM motor without a large gap on the d-axis side. FIG. 7 is a model diagram showing the relative relationship between the magnetic flux vector on the outer peripheral side of the magnetic pole (permanent magnet) and the armature magnetic flux vector when the V-shaped IPM motor without a large gap on the d-axis side is driven at the maximum load. FIG. 8 is a graph showing the correspondence (characteristic) between the current phase and the output torque with respect to the input current of the IPM type motor. FIG. 9 is a magnetic flux diagram of the armature magnetic flux when the V-shaped IPM motor without a large gap on the d-axis side is driven at a low load. FIG. 10 is a path diagram showing a path taken by the combined magnetic flux together with a magnetic flux diagram of the combined magnetic flux of the magnetic flux and the armature magnetic flux when the V-shaped IPM motor without a large gap on the d-axis side is driven at a low load. FIG. 11 is a graph showing a change in generated torque and a reduction rate of torque ripple when the embedded permanent magnet of the V-shaped IPM motor with a d-axis side gap is shortened. FIG. 12 is a graph showing changes in the fifth-order spatial harmonics superimposed when the embedded permanent magnet of the V-shaped IPM motor with a d-axis side gap is shortened. FIG. 13 is a graph showing a torque generation ratio in a low load driving region of a V-shaped IPM motor without a large gap on the d-axis side and a V-shaped IPM motor with a d-axis side gap. FIG. 14 is a graph showing a torque generation ratio in a maximum load drive region of a V-shaped IPM motor without a large gap on the d-axis side and a V-shaped IPM motor with a d-axis side gap. FIG. 15 is a magnetic flux diagram showing an armature magnetic flux at the time of maximum load driving of a V-shaped IPM motor with a d-axis side gap. FIG. 16 is a magnetic flux diagram showing a combined magnetic flux of a magnetic flux and an armature magnetic flux when a V-shaped IPM motor with a d-axis side gap is driven at a low load. FIG. 17 is a magnetic flux diagram showing a combined magnetic flux of the magnet magnetic flux and the armature magnetic flux when the V-shaped IPM motor with the d-axis side gap is driven at the maximum load. FIG. 18A is a magnetic flux diagram of magnet magnetic flux of a V-shaped IPM motor having no large gap on the d-axis side and having no center groove formed thereon. FIG. 18B is a vector diagram of a combined magnetic flux of an armature magnetic flux and a magnet magnetic flux in the vicinity of the d-axis at the maximum load of a V-shaped IPM motor having no large gap on the d-axis side and having no center groove formed. FIG. 19A is a magnetic flux diagram of the magnetic flux of a V-shaped IPM motor with no center groove formed with a large gap on the d-axis side. FIG. 19B is a vector diagram of a combined magnetic flux of an armature magnetic flux and a magnet magnetic flux in the vicinity of the d-axis at the maximum load of a V-shaped IPM motor with no center groove formed with a large gap on the d-axis side. FIG. 20 shows a one-tooth linkage comparing the structure without a large groove on the d-axis side shown in FIG. 18A and the structure without a large groove on the d-axis side shown in FIG. 19A. It is a graph which shows a magnetic flux waveform. FIG. 21 is a graph showing the content of spatial harmonics superimposed on one inter-linkage magnetic flux waveform by expanding the magnetic flux waveform shown in FIG. 20 by Fourier series. FIG. 22 is a vector diagram of a combined magnetic flux of an armature magnetic flux and a magnet magnetic flux in the vicinity of the d-axis at the maximum load of a V-shaped IPM motor having a center groove formed with a large gap on the d-axis side. FIG. 23 is a graph showing a torque waveform at the maximum load for comparing the present embodiment with the structure having no center groove shown in FIG. 19A. FIG. 24 is a graph comparing the degree of superposition of harmonic torque superimposed on the torque waveform by expanding the torque waveform shown in FIG. 23 by Fourier series. FIG. 25 is an enlarged structural view of one magnetic pole of the rotor showing parameters used when determining the dimension and shape of the center groove. FIG. 26 is a graph showing changes in torque ripple when the ratio of R4 to the outer radius R1 in the dimensional shape of the center groove shown in FIG. 25 is changed as a parameter. FIG. 27 is a graph showing a phase voltage waveform and a line voltage waveform when the outer opening angle θa in the dimensional shape of the center groove shown in FIG. 25 is changed as a parameter. FIG. 28 is a graph showing a torque waveform at low load comparing the present embodiment with the structure without the center groove shown in FIG. 19A. FIG. 29 is a graph comparing the degree of superposition of harmonic torque superimposed on the torque waveform by expanding the torque waveform shown in FIG. 28 by Fourier series.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. 1 to 29 are diagrams showing an embodiment of an IPM type electric rotating machine according to the present invention. Here, in the description of the present embodiment, the rotation direction is illustrated by taking as an example a case where the rotor is rotated counterclockwise (CCW) with respect to the stator.
In FIG. 1, an electric rotating machine (motor) 10 includes a stator (stator) 11 formed in a substantially cylindrical shape, and a rotary drive shaft 13 that is rotatably housed in the stator 11 and coincides with an axis. And a fixed rotor (rotor) 12. The electric rotating machine 10 has a performance suitable for mounting in a wheel or wheel as a drive source similar to that of an internal combustion engine, for example, in a hybrid vehicle (HEV) or an electric vehicle (EV).

The stator 11 is formed with a plurality of stator teeth 15 extending in the normal direction of the axial center so that the outer peripheral surface 12a of the rotor 12 faces the inner peripheral surface 15a via the gap G. . A three-phase winding (not shown) constituting a coil for generating a magnetic flux for rotationally driving the rotor 12 accommodated inside is wound around the stator teeth 15 by distributed winding.
The rotor 12 is manufactured to have an IPM (Interior Permanent Magnet) structure in which a pair of permanent magnets 16 are embedded as one magnetic pole so as to be V-shaped to open toward the outer peripheral surface 12a. The rotor 12 is formed such that a V-shaped space 17 that is fitted in a corner portion 16a of a plate-like permanent magnet 16 extending in the front and back direction of the drawing and is housed in a stationary state faces the outer peripheral surface 12a. .
The V-shaped space 17 is a space 17a in which the permanent magnet 16 is fitted and accommodated, and spaces 17b and 17c (hereinafter referred to as flux) that are located on both sides in the width direction of the permanent magnet 16 and function as flux barriers that restrict the wraparound of the magnetic flux. Barriers 17b and 17c). In the V-shaped space 17, the permanent magnet 16 is extended in the normal direction between the spaces 17 c so that the permanent magnet 16 can be positioned and held against the centrifugal force during high-speed rotation. A center bridge 20 for connecting and supporting is formed.

In the electric rotating machine 10, the space between the stator teeth 15 on the stator 11 side constitutes a slot 18 for forming a coil by being wound through a winding. On the other hand, in the rotor 12, six stator teeth 15 on the stator 11 side face each of the eight sets of permanent magnets 16. In short, the electric rotating machine 10 is constructed such that the six slots 18 on the stator 11 side correspond to one magnetic pole formed on the pair of permanent magnets 16 side on the rotor 12 side. That is, the electric rotating machine 10 is wound with 8 poles (4 pole pairs), 48 slots, single-phase distributed winding 5 pitches, with the N and S poles of the permanent magnet 16 alternately arranged for each adjacent magnetic pole. It is made to a wired three-phase IPM motor. In other words, the electric rotating machine 10 is manufactured in an IPM type structure in which the number of slots per phase per pole q = (number of slots / number of poles) / number of phases = 2.
Thus, the electric rotating machine 10 can be driven to rotate by energizing the coil in the slot 18 of the stator 11 and passing the magnetic flux through the rotor 12 facing the stator teeth 15. At this time, the electric rotating machine 10 (the stator 11 and the rotor 12) tries to minimize the magnetic path through which the magnetic flux passes, in addition to the magnet torque caused by the attractive force and the repulsive force generated between the permanent magnets 16. It can be rotationally driven by the total torque including the reluctance torque. Therefore, the electric rotating machine 10 can output electrical energy that is energized and input as mechanical energy from the rotary drive shaft 13 that rotates integrally with the rotor 12 with respect to the stator 11.
In addition, the stator 11 and the rotor 12 are laminated in the axial direction so that a thin plate made of electromagnetic steel plate material such as silicon steel has a thickness corresponding to a desired output torque, and is caulked so as to maintain the laminated state. 19 or the like.

Here, as shown in FIG. 2 as a magnetic flux diagram, the electric rotating machine 10 includes a plurality of stator teeth 15 corresponding to a pair of permanent magnets 16 constituting one magnetic pole, and the outer peripheral side of the stator 11 ( A winding coil is distributedly wound in the slot 18 so as to form a magnetic path (armature magnetic flux) that passes through the rotor 12 from the back side of the stator teeth 15. The permanent magnet 16 is accommodated in the fitting space 17a of the formed V-shaped space 17 so as to follow the magnetic path of the armature magnetic flux Ψr, in other words, so as not to prevent the formation of the armature magnetic flux Ψr. Has been.
The magnetic path (magnet magnetic flux Ψm) of the permanent magnet 16 extends vertically from the N and S poles on the front and back surfaces of the pair of permanent magnets 16 constituting one magnetic pole as shown in FIG. 3 as a magnetic flux diagram. In particular, on the stator 11 side, the corresponding stator teeth 15 pass through the back side.

In the IPM structure in which the permanent magnet 16 is embedded in the V-shape in the rotor 12, the direction of the magnetic flux generated by the magnetic pole, that is, the central axis between the V-shaped permanent magnets 16 is defined as the d-axis. The central axis between the permanent magnets 16 between adjacent magnetic poles that are orthogonal to each other electrically and magnetically is defined as the q axis. The rotor 12 is formed so that the inner space 17c located on the d-axis side of the V-shaped space 17 is enlarged to a large gap toward the axis and functions as a flux barrier 17c.
As a result, in the electric rotating machine 10, as shown in FIG. 2, the armature magnetic flux Ψr entering the rotor 12 from the stator teeth 15 is greatly increased so as not to go around the outer periphery of the V-shaped space 17 ( It is formed so as to take a path detouring toward the axis) and returning to the stator teeth 15. In short, in the electric rotating machine 10, the rotor 12 is constructed as a V-shaped IPM motor with a d-axis gap.
In addition, the electric rotating machine 10 is arranged so that the armature magnetic flux Ψr entering from the stator teeth 15 corresponding to the d-axis does not overlap with many fifth-order and seventh-order spatial harmonics that cause an increase in torque ripple. A center groove (adjustment groove) 21 extending in a direction parallel to the inner peripheral surface 15a of the stator teeth 15 (axial direction) is formed on the outer peripheral surface of the stator teeth 15. The optimum dimensional shape of the center groove 21 will be described later.

Ｐｐ：極対数、Ψｍ：電機子（ステータティース１５）鎖交磁石磁束、
ｉｄ：線電流のｄ軸成分、ｉｑ：線電流のｑ軸成分、
Ｌｄ：ｄ軸インダクタンス、Ｌｑ：ｑ軸インダクタンス As described above, in the case of the electric rotating machine 10 having the IPM structure in which the permanent magnet 16 is embedded in the V shape in the rotor 12, the torque T can be expressed by the following equation (1), as shown in FIG. In addition, high torque and high efficiency operation is realized by driving at a current phase that maximizes the sum of the magnet torque Tm and the reluctance torque Tr.

Pp: number of pole pairs, Ψm: armature (stator teeth 15) interlinkage magnet magnetic flux,
id: d-axis component of line current, iq: q-axis component of line current,
Ld: d-axis inductance, Lq: q-axis inductance

By the way, in the case of the rotor 12A of related technology provided with the flux barrier 17d equivalent to the flux barrier 17b outside the V-shaped space 17 instead of the flux barrier 17c of the d-axis side gap, the magnetic flux diagram of FIG. A magnetic path of the illustrated permanent magnet 16 is formed, and the magnet magnetic flux Ψm is a vector Vm in the direction illustrated in the magnetic flux vector diagram of FIG. 5B. Further, the armature magnetic flux Ψr generated by energizing the coil accommodated in the slot 18 is formed in the magnetic path shown in the magnetic flux diagram of FIG. 6A, and becomes the vector Vr in the direction shown in the magnetic flux vector diagram of FIG. 6B. It has become.
In this type of electric rotating machine, at the time of maximum load driving, the current phase angle is advanced to achieve high torque and high efficiency driving. In the related art rotor 12A, as shown in the magnetic flux vector diagrams of FIGS. 5B and 6B, in the small region A1 near the d axis located on the outer peripheral side of the V-shaped space 17 (magnetic pole), the magnetic flux Ψm and the armature The reluctance torque Tr is in a state of being driven while canceling out (cancelling) the magnet torque Tm because the magnetic flux Ψr has a reverse magnetic field relationship. In short, as shown in FIG. 7, the magnetic pole outer peripheral side small area A1 is an interference area where the magnet magnetic flux Ψm and the armature magnetic flux Ψr face each other in a reverse positional relationship at an included angle of 90 degrees or more. The armature magnetic flux Ψr is wasted to suppress (cancel) the magnetic flux Ψm generated in the range B on the d-axis side of the permanent magnet 16 adjacent to the small side area A1.
From this, it can be said that the d-axis side range B of the permanent magnet 16 corresponding to the magnetic pole outer peripheral side small area A1 does not actively contribute to the torque T. By using a magnetic circuit that maintains the same salient pole ratio while reducing the portion B, the magnet amount of the permanent magnet 16 itself can be reduced.
Here, since the torque T is expressed by the above equation (1), when the magnet amount of the permanent magnet 16 is reduced, the reluctance torque Tr is increased to be equivalent to the case where the magnet amount of the permanent magnet 16 is not reduced. can do. The reluctance torque Tr can be increased by increasing the difference between the d-axis inductance Ld and the q-axis inductance Lq, that is, the salient pole ratio.
Therefore, in the rotor 12 of the present embodiment, the salient pole ratio is increased while the amount of the permanent magnet 16 is reduced by replacing the d-axis side range B of the permanent magnet 16 with a gap (restricted region) having a small magnetic permeability. Thus, a torque T equal to or higher than that before replacement can be obtained. In other words, the reluctance torque Tr can be increased by effectively using the armature magnetic flux Ψr that has been wasted to suppress the magnetic flux Ψm generated in the d-axis side range B of the permanent magnet 16. Even if the magnet amount of the magnet 16 is reduced, an equivalent torque T can be obtained.

β：電流位相角度、Ｉａ：相電流値 The torque T can also be expressed as in the following formula (2). In the low load region where the current value Ia is small, the ratio of the magnet torque Tm is high, and the current value Ia is low as shown in FIG. The current phase β at the maximum torque becomes closer to zero. Waveforms i to v in FIG. 8 indicate current phase-torque characteristics at the respective current values Ia (i) to Ia (v), and the magnitude of the current value Ia is i <ii <iii <iv <iv <. The relationship is v. Therefore, the ratio (dependence) of the magnet torque Tm naturally increases during low-load driving, but a magnetic circuit that effectively uses the magnet torque Tm to the maximum is desirable.

β: current phase angle, Ia: phase current value

In the related art rotor 12A, as shown in FIG. 9, in the low load region with a low current value, the current phase β is driven under a condition close to zero, so that the magnetic flux amount of the armature magnetic flux Ψr becomes the q axis. (Between adjacent permanent magnets 16 of different magnetic poles). For this reason, as a path of the magnetic flux Ψs obtained by synthesizing the magnetic flux Ψm with the armature magnetic flux Ψr, it is preferable to use a magnetic circuit that passes through the magnetic paths MP1 and MP2 shown in FIG. As a result, the synthesized magnetic flux Ψs can increase the q-axis inductance Lq by dispersing the q-axis magnetic path (magnetic flux) (avoid saturation), and can actively use the reluctance torque Tr. can do.
The magnetic path MP1 is connected to the rotor teeth 12A via the air gap G from the stator teeth 15 on the stator 11 side and enters between the magnetic poles, and then forms a magnetic pole on the rotation direction traveling side (left side in the figure). A path is taken through the permanent magnet 16 on the side from the inner peripheral side. Further, the magnetic path MP1 takes a path that passes through the outer peripheral area A2 of the magnetic pole and returns to the stator teeth 15 through the air gap G again.
The magnetic path MP2 enters between the magnetic poles in the same manner as the magnetic path MP1, and then passes through the outer peripheral side area A2 of the magnetic pole through the separation-side permanent magnet 16 that forms the magnetic pole on the rotation direction traveling side from the inner peripheral side. Then, a path to return to the stator teeth 15 again through the air gap G is taken.

For example, in the magnetic paths MP1 and MP2, when both end sides (magnetic pole outer end portions) of the pair of permanent magnets 16 are cut and moved to the inside, a large flux barrier exists on both end sides and the vicinity of the center of the magnetic pole In particular, it becomes difficult to take a path on the right side of the magnetic pole outer peripheral side area A2, and the entire area A2 cannot be used effectively.
On the other hand, when the center side (the inner end of the magnetic pole) of the pair of permanent magnets 16 is cut and moved to the outside, a large flux barrier exists on the center side, and the magnetic flux path can be dispersed on both sides of the magnetic pole. In addition, the magnetic flux can pass through the region A2 evenly by actively and effectively including the path on the right side of the magnetic pole outer region A2. In the case of this structure, the magnetic path that connects the N pole and the S pole of the adjacent permanent magnet 16 of the magnetic pole after the permanent magnet 16 of the magnetic pole on the reverse side in the rotation direction has passed through from the outer periphery to the inner periphery. MP3 can also be taken. In this magnetic path MP3, it is possible to pass through the outer periphery side area A2 of the magnetic pole on the traveling side in the rotation direction through the same path as the magnetic path MP1, and the dispersion efficiency of the magnetic flux is high.
Therefore, the rotor 12 has an embedded structure of a pair of permanent magnets 16 that form magnetic poles, while maintaining a V shape so as not to disturb the armature magnetic flux Ψr that generates the reluctance torque Tr, the rotor 12 It is preferable to adopt a shape that approaches the outer end portion. Further, it is preferable to adopt a structure in which a flux barrier 17c that restricts the magnetic flux from taking a short circuit path is formed between the pair of permanent magnets 16 (the inner end of the magnetic pole). Further, a center groove 21 is formed on the outer peripheral surface of the rotor 12 on the d-axis to limit the saturation of the armature magnetic flux Ψr entering from the stator teeth 15 on the stator 11 side, in other words, to distribute the magnetic flux Ψr. It is preferable to adopt a structure that By adopting such a structure, the rotor 12 can disperse the q-axis magnetic path (magnetic flux), increase the q-axis inductance Lq, and actively use the reluctance torque Tr.

The permanent magnet 16 compares and determines the optimum value of the length (width) Wpm in the longitudinal direction in the drawing with reference to the case where the length Wpm is not shortened.
Specifically, with the number of poles P and the outer radius R1 from the axis of the rotor 12 to the outer peripheral surface being fixed values, the length Wpm of the permanent magnet 16 installed at the outer end of the magnetic pole is a variable (inner end side). It is determined by changing the ratio δ calculated by the following equation (3), with the position of the end side as displacement). As a determining factor, the change in the torque T at the maximum load per unit unit and the change in the reduction rate of the torque ripple that is the fluctuation range of the torque T with respect to the ratio δ are displayed as a graph by magnetic field analysis. Then, as shown in FIG. In per unit unit, for example, 1.0 [p. u. ] Means the same.
δ = (P × Wpm) / R1 (3)
In FIG. 11, the ratio δ = 1.84 is the case of the permanent magnet 16 having a shape dimension (magnet reduction amount 0%) that does not shorten the length Wpm, and the ratio shape δ = 1.38 (magnet reduction amount 24. 7%), it can be seen that a torque T equivalent to 1.0 (p.u.) can be obtained. The permanent magnet 16 can obtain an equivalent torque T by setting the ratio δ = 1.38 even during normal low-speed rotational load.
Here, in FIG. 11, a rotor 12A of related technology including flux barriers 17b and 17d having the same size on the inner and outer ends of the V-shaped space 17 is set as a comparison target. On the other hand, in the case of the rotor 12 of the present embodiment, the armature magnetic flux Ψr can be effectively divided and distributed by providing the flux barrier 17c and the center groove 21. Therefore, in the rotor 12, the reluctance torque Tr can be generated effectively, and the torque T is improved and the torque ripple is reduced even at the ratio δ = 1.84 where the permanent magnet 16 has the same length Wpm. Yes. That is, in FIG. 11, the change of the torque T and the torque ripple with respect to the ratio δ is illustrated by shortening the length Wpm of the permanent magnet 16 with the structure of the rotor 12. When the length Wpm of the permanent magnet 16 is shortened with the structure of the rotor 12A of the related art, there is no significant change in the torque T from the ratio δ = 1.84 to the ratio δ = 1.38 (1 .0 [pu]]).

  In addition, in the electric rotating machine, an induced voltage (counterelectromotive voltage) corresponding to the amount of permanent magnet to be embedded is generated with the rotation of the rotor, and spatial harmonics of magnetostriction caused by the field weakening are superimposed. become. In this spatial harmonic, the fifth, seventh, eleventh, and thirteenth components cause torque ripple and cause an increase in iron loss. From this, for example, when the generation of fifth-order spatial harmonics with respect to the ratio δ is graphed in units of per unit, the result is as shown in FIG. 12, and the fifth-order space becomes smaller as the ratio δ = 1.75 or less. It can be seen that the generation of harmonics can be suppressed. In this case, the magnet amount of the permanent magnet 16 can be reduced by 4.7% or more, and the internal efficiency of the permanent magnet 16 can be improved while reducing the iron loss by reducing the spatial harmonics of the magnetostriction and improving the driving efficiency. The generation of eddy currents at this point can be limited to suppress heat generation.

From this, in the rotor 12 of the present embodiment, in order to reduce the amount of the permanent magnet 16 used while obtaining the torque T equivalent to the rotor 12A of the related art, the length Wpm of the permanent magnet 16 is shortened. (The magnet amount is reduced by 24.7%) and the ratio δ is preferably about 1.38, and the torque ripple can be reduced. In short, the permanent magnet 16 has a ratio δ = 1.38 (magnet reduction amount 24.7%) to 1.75 (magnet reduction amount 4.7%) in accordance with desired characteristics such as torque T and torque ripple. The size and shape may be appropriately selected.
Therefore, the electric rotating machine 10 is a V-shaped IPM motor with a d-axis gap formed by shortening the length Wpm of the permanent magnet 16 so as to have the equivalent torque T and forming the dimensional shape with the ratio δ = 1.38. When the magnetic field analysis is performed in the case of a V-shaped IPM motor that does not shorten the permanent magnet 16, the ratio of the magnet torque Tm and the reluctance torque Tr changes as shown in FIGS. It turns out that output is possible. The V-shaped IPM motor with a d-axis gap has a structure with a large gap flux barrier 17c on the d-axis side, and the simple V-shaped IPM motor has a structure with a small flux barrier 17d on the d-axis side. It is.
FIG. 13 illustrates the ratios of torques Tm and Tr in the low load region, and FIG. 14 illustrates the ratios of torques Tm and Tr in the maximum load region. In any case, in the case of a V-shaped IPM motor with a d-axis gap, it can be seen that the reluctance torque Tr is increased instead of decreasing the magnet torque Tm in order to shorten the permanent magnet 16. That is, the electric rotating machine 10 replaces the permanent magnet 16 near the d-axis to form a large gap space flux barrier 17c and a center groove 21, so that the magnetic pole outer peripheral small area A1 shown in FIG. 6B and FIG. The magnet magnetic flux Ψm that cancels the armature magnetic flux Ψr can be reduced. As a result, the electric rotating machine 10 can increase the q-axis inductance Lq so that the difference (saliency ratio) from the d-axis inductance Ld is larger than that of the non-shortened V-shaped IPM motor, and the reluctance torque Tr can be increased. The same torque T can be secured by making effective use.

With this structure, as shown in FIG. 15 as a magnetic flux diagram, the electric rotating machine 10 causes the armature magnetic flux Ψr concentrated in the small area A1 on the outer peripheral side of the pair of permanent magnets 16 forming the magnetic poles to The magnetic path Mr1 that passes through the magnetic pole outer peripheral side small area A1 can also be effectively divided (divided) into a magnetic path Mr2 that bypasses the inner peripheral side of the d-axis side space 17c of the V-shaped space 17. As a result, the electric rotating machine 10 reduces the magnetic interference between the magnet magnetic flux Ψm and the armature magnetic flux Ψr (d-axis / q-axis), and on the traveling side (left side in the drawing) of the magnetic pole outer peripheral small area A1. It is possible to effectively contribute to the generation of the torque T by avoiding local magnetic saturation.

Therefore, as shown in the magnetic flux diagram of FIG. 16, the electric rotating machine 10 passes through a magnetic path MP0 through which the combined magnetic flux Ψs of the magnet magnetic flux Ψm and the armature magnetic flux Ψr mainly passes through the permanent magnet 16 during low load driving. On the other hand, when the maximum load is driven, the combined magnetic flux Ψs can be divided into the magnetic path MP1 and the magnetic path MP2 as shown in the magnetic flux diagram of FIG. As a result, the magnetic interference can be reduced and the local magnetic saturation state can be avoided, and the torque T equal to or higher than that of the permanent magnet 16 can be efficiently generated while the amount of the permanent magnet 16 is reduced . Contact name synthetic flux Ψs during low load driving, the ratio of magnetic flux Ψm is greater than the armature flux Pusaiaru.
Further, the electric rotating machine 10 replaces the permanent magnet 16 with, for example, a ratio δ = 1.44 and replaces it with a low-permeability flux barrier 17c (reduces the magnetic flux Ψm) to reduce the magnet amount by 23%. As the inertia (inertial force) is reduced, the induced voltage constant can be reduced by about 13.4%, and the output on the high-speed rotation side can be increased. Furthermore, in the electric rotating machine 10, the heat generation due to the eddy current generated in the permanent magnet 16, iron loss, and electromagnetic noise can be suppressed by reducing the spatial harmonics that become magnetic distortion.

In the rotor 12A shown in FIG. 18A, since the permanent magnet 16 exists up to the vicinity of the d-axis, a large amount of magnet magnetic flux Ψm is generated in the magnetic pole outer peripheral region A2. On the other hand, in the rotor 12C not provided with the center groove 21 shown in FIG. 19A, a gap flux barrier 17c is formed in the vicinity of the d-axis, so that the magnetic flux Ψm generated from the permanent magnet 16 is reduced. The orthogonality decreases, in other words, the magnetic flux density of the magnet magnetic flux Ψm near the d-axis decreases. For this reason, for the q-axis magnetic path Ψq, the inductance increases as the magnetic resistance near the d-axis decreases. As a result, in the rotor 12C , due to the difference in density of the magnetic flux interlinking with the outer peripheral surface 12a, harmonics are superimposed on the magnetic flux, and the efficiency is reduced due to an increase in torque ripple and iron loss. .
For example, in the vicinity of the d-axis of the rotor 12A, as shown in the magnetic flux vector diagram at the maximum load in FIG. 18B, the magnetic flux density linked from the facing stator teeth 15D corresponding to the magnetic path loop of the armature magnetic flux Ψr. Is not expensive. On the other hand, in the vicinity of the d-axis of the rotor 12C , as shown in the magnetic flux vector diagram at the maximum load in FIG. 19B, the interlinkage magnetic flux density is higher than the magnetic flux in the stator teeth 15D in FIG. Magnetic flux to be increased.
This means that one tooth chain that passes through the gap G between the rotor 12A (without the flux barrier 17d and the center groove 21) and the rotor 12C (without the flux barrier 17c and the center groove 21) and one stator tooth 15. Comparing the alternating magnetic flux waveforms, as shown in the graph of FIG. 20, the rotor 12C is more likely to flow the magnetic flux at the location indicated by “P” in the drawing where the vicinity of the d-axis is affected, and the harmonics are more easily superimposed. It has become. For example, when the magnetic flux waveform shown in FIG. 20 is expanded in the Fourier series, as shown in FIG. 21, the magnetic flux waveform of the rotor 12C has a higher content ratio of the fifth and seventh spatial harmonics than the rotor 12A. It can also be seen from the overlapping.

Therefore, the electric rotating machine 10 forms a center groove 21 that adjusts so as to increase the magnetic resistance in the gap G with the inner peripheral surface 15a of the stator teeth 15 on the d-axis of the outer peripheral surface 12a of the rotor 12. doing. In the rotor 12 in which the center groove 21 is formed, as shown in the magnetic flux vector diagram at the maximum load in FIG. 22, it is possible to suppress an increase in the magnetic flux entering from the stator teeth 15 facing near the d-axis of the rotor 12. is made of.
Further, in the rotor 12 (with center groove 21) and the rotor 12C (without center groove 21), comparing the torque waveform, as shown in the graph of FIG. 23, based on the rotor 12C (1.0 [pu]]), the torque waveform of the rotor 12 with the center groove 21 can reduce the amplitude, and torque ripple can be suppressed. Further, when the torque waveform shown in FIG. 23 is expanded by Fourier series, as shown in FIG. 24, the torque waveform of the rotor 12 with the center groove 21 has higher harmonics of the 6th, 12th, 18th and 24th. Wave torque can be greatly reduced. FIG. 23 shows the torque waveform of the instantaneous torque based on the average torque of the rotor 12C (1.0 [pu]).

By the way, in the case of three phases, the torque ripple of the electric rotating machine 10 has a 6f-order component (in terms of electrical angle) due to the spatial harmonics superimposed on the magnetic flux waveform for each pole of one phase and the time harmonics included in the phase current ( f = 1, 2, 3,...: natural number).
The cause of torque ripple will be described below. The three-phase output (electric power) P (t) and the torque τ (t) have an angular velocity of ωm, an induced electromotive force of each phase Eu (t), Ev (t), If Ew (t) and the current of each phase are Iu (t), Iv (t), and Iw (t), they can be obtained by the following equations (4) and (5).
P (t) = _Eu (t) _Iu (t) + _Ev (t) _Iv (t) + _Ew (t) _Iw (t) (4)
τ (t) = P (t) / ω _m
= [E _u (t) I _u (t) + E _v (t) I _v (t) + E _w (t) I _w (t)] (5)
The three-phase torque is the sum of the torques of the U-phase, V-phase, and W-phase, where m represents the harmonic component of the current, n represents the harmonic component of the voltage, and the U-phase current I _u (t) Is represented by the following equation (6), the U-phase torque τ _u (t) can be expressed by the following equation (7).

また、この誘起電圧は、磁束を時間微分して求めることができることから、各誘起電圧に含まれる高調波の次数と１相１極磁束に含まれる高調波も同じ次数成分が発生することになる。その結果、３相交流モータにおいては、磁束（誘起電圧）に含まれる空間高調波次数ｎと相電流に含まれる時間高調波次数ｍとの組み合わせが６ｆになるときに、その６ｆ次成分のトルクリプルが発生していることになる。
よって、３相モータのトルクリプルは、上述するように、１相１極における磁束波形における空間高調波ｎと相電流の時間高調波ｍにおいては、ｎ±ｍ＝６ｆ（ｆ：自然数）のときに発生することから、例えば、１１次と１３次の空間高調波（ｎ＝１１、１３）が重畳していると相電流の基本波（ｍ＝１）との合わせにより１２次の高調波トルクが発生することが分かる。 Since the phase current I (t) and the phase voltage E (t) are both symmetrical waves, “n” and “m” are only odd numbers. The V-phase torque and the W-phase torque other than the U-phase are “+ 2π / 3 (rad)”, “−2π / 3 (” with respect to the U-phase induced voltage E _u (t) and the U-phase current I _u (t), respectively. rad) "phase difference, the overall torque is canceled (offset) so that only the coefficient term of" 6 "remains,
6f = n ± m (f: natural number), s = nα _n + mβ _m , t = nα _n −mβ _m
And can be expressed as the following equation (8).

In addition, since this induced voltage can be obtained by time differentiation of the magnetic flux, the same order component is generated in the harmonic order contained in each induced voltage and the harmonic contained in the 1-phase 1-pole magnetic flux. . As a result, in the three-phase AC motor, when the combination of the spatial harmonic order n included in the magnetic flux (induced voltage) and the time harmonic order m included in the phase current becomes 6f, the torque ripple of the 6f-order component Will occur.
Therefore, the torque ripple of the three-phase motor is, as described above, when n ± m = 6f (f: natural number) in the space harmonic n in the magnetic flux waveform in one phase and one pole and the time harmonic m in the phase current. For example, if the 11th and 13th spatial harmonics (n = 11, 13) are superposed, the 12th harmonic torque is combined with the fundamental wave of the phase current (m = 1). It can be seen that it occurs.

In the electric rotating machine 10, the optimum dimensional shape of the center groove 21 in the rotor 12 is determined based on torque characteristics such as the torque ripple.
As shown in FIG. 25, the center groove 21 has a ratio R4 / ratio to the outer radius R1 to the outer peripheral surface 12a of the rotor 12 by changing the separation distance R4 from the axial center to the groove bottom 21a in the normal direction. The dimension and shape are determined by the torque ripple shown in FIG. 26 obtained when R1 is used as a parameter.
First, the depth of the center groove 21 is formed in the following dimension shape so that the torque ripple generated at the maximum load can be reduced on the basis of the dimension shape without the center groove 21 (R4 / R1 = 1.0).
0.98 ≦ R4 / R1 <1.0

Further, the center groove 21 of the rotor 12 needs to have a dimensional shape determined from the relative relationship with respect to the stator teeth 15 on the stator 11 side, and as shown in FIG. The outer opening angle θa of the outer peripheral surface 12a and the inner opening angle θb of the groove bottom 21a inside the outer peripheral surface 12a can be defined.
When the rotor 12 is changed with the outer opening angle θa of the center groove 21 as a parameter, the peak F and the top W in the figure are shown in the graph in FIG. 27 in which the phase voltage and the line voltage are associated with each other. It is affected at the points indicated by.
Specifically, for example, the width of G1 to G3 of the U-phase voltage waveform in FIG. 27 depends on the width of the outer opening angle θa of the center groove 21 from the relative positional relationship between the stator 11 and the rotor 12. Change. When the outer opening angle θa is narrowed, the U-phase voltage waveform becomes narrower between G1 and G3 and becomes a sharp waveform with the top W being the highest vertex, and the line voltage waveform has a peak F approaching the top W. Thus, the waveform approximates a triangular wave. On the contrary, the U-phase voltage waveform becomes a waveform in which the top W between G1 and G3 becomes flat when the outer opening angle θa of the center groove 21 is increased, and the line voltage waveform has a peak F from the top W. It becomes a waveform that approximates a trapezoidal wave that spreads apart and becomes easier to superimpose the fifth and seventh spatial harmonics.

Here, as described above, the center groove 21 needs to increase the magnetic resistance in the gap G between the rotor 12 and the stator teeth 15 (decrease the magnetic permeability), while making the outer opening angle θa too wide. In addition, since the fifth and seventh spatial harmonics are easily superimposed, it is necessary to form the minimum necessary size and shape.
As shown in FIG. 25, the structure of the rotor 12 and the stator 11 includes the opening width SO of the slot 18 on the rotor 12 side, the facing width TB of the inner peripheral surface 15a of the stator teeth 15, and the inner periphery of the stator teeth 15. Assuming that the tip end portion width TW inside the surface 15a and the air gap width AG of the gap G between the rotor 12 and the stator teeth 15 are as follows.
First, since it is necessary to increase the magnetic resistance in the gap G, the center groove 21 needs to be equal to or larger than the facing width TB of the stator teeth 15. As a lower limit value of the outer opening angle θa, the shape surrounded by the facing width TB and the axis of the rotor 12 approximates to an isosceles triangle (2 × right triangle).
2 × tan ⁻¹ ((TB / 2) / (R1 + AG)) ≦ θa
It can be.
Further, the slot 18 needs to satisfy an opening width SO> air gap width AG of the slot 18 in consideration of automatic coil insertion and a necessary energy density. From this relationship, the magnetic resistance in the gap G is lower than the opening space of the slot 18, and it is necessary to reduce the amount of magnetic flux interlinking from the tip corner K of the stator teeth 15 (see FIG. 22 ) to the rotor 12 side. For this reason, the center groove 21 needs to be equal to or smaller than the width to the inner peripheral surface 15a of the adjacent stator teeth 15, and as an upper limit value of the outer opening angle θa, similarly,
θa ≦ 2 × tan ⁻¹ ((SO + (TB / 2)) / (R1 + AG))
It can be.

Next, the inner opening angle θb of the groove bottom 21a of the center groove 21 is set to an outer opening angle θa equal to or smaller than the width to the inner peripheral surface 15a of the adjacent stator teeth 15 as an upper limit, similarly to the outer opening angle θa.
θb ≦ 2 × tan ⁻¹ ((SO + (TB / 2)) / (R1 + AG))
It can be.
On the other hand, the lower limit value of the inner opening angle θb of the groove bottom 21 a of the center groove 21 is adjusted so that the lower limit value of the outer opening angle θa is the facing width TB of the stator teeth 15 and the magnetic resistance in the gap G is increased. Therefore, 0 ° ≦ θb without groove bottom 21a
It is good.
The facing width TB and the tip width TW of the stator teeth 15 are not satisfied when the tip of the stator teeth 15 is pointed.
TW ≦ TB
It becomes.

Here, in the rotor 12, also in the low load, when comparing the rotor 12C and torque waveforms of the center groove 21 without, as shown in the graph of FIG. 28, based on the rotor 12C (1 0.0 [pu]]), the torque waveform of the rotor 12 with the center groove 21 has a smaller amplitude, and torque ripple can be suppressed. Further, when the torque waveform shown in FIG. 28 is expanded by Fourier series, as shown in FIG. 29, the torque waveform of the rotor 12 with the center groove 21 can reduce the sixth-order harmonic torque.
In the above, the influence of the center groove 21 on the torque characteristics will be mainly described. However, the center groove 21 is useful because it can be used as a mark at the time of manufacturing such as assembly. For example, when the so-called skew is applied in a state where the positional relationship of the permanent magnet 16 in the axial direction is twisted, the presence or absence of the skew can be confirmed from the linearity of the center groove 21 in the axial direction. .

As described above, in the present embodiment, the d-axis side range B of the permanent magnet 16 is reduced and replaced with the large flux barrier 17c, so that the magnet magnetic flux Ψm in the direction that cancels the armature magnetic flux Ψr is eliminated to interfere (cancel) each other. And the passage of the armature magnetic flux Ψr in the range B can be restricted.
Therefore, a large magnet torque Tm and a reluctance torque Tr can be obtained by effectively using the armature magnetic flux Ψr and the magnet magnetic flux Ψm on the d-axis side while reducing the amount of permanent magnets 16 used. In addition, the output on the high speed rotation side can be increased by reducing the induced voltage constant, and the heat generation due to the eddy current of the permanent magnet 16 can be suppressed to suppress the demagnetization due to the temperature change, thereby lowering the heat resistant grade. Can reduce costs.
Further, the center groove 21 of the rotor 12 has a harmonic torque of 0.98 ≦ R4 / R1 <1.0 with respect to the outer radius R1 of the rotor 12 by setting the length R4 to the groove bottom 21a. And torque ripple can be effectively reduced. Further, the center groove 21 has 2 × tan ⁻¹ ((tooth-to-face width TB / 2) / (rotor outer radius R1 + air gap width AG)) ≦ outer opening angle θa ≦ 2 × tan−1 ((slot opening Width SO + (tooth-to-face width TB / 2)) / (rotor outer radius R1 + air gap width AG)), 0 ° ≦ inner opening angle θb ≦ 2 × tan−1 ((slot opening width SO + (tooth-to-face width TB / 2)) / (Rotor outer radius R1 + air gap width AG)), and the shape of the tooth tip width TW ≦ tooth-to-face width TB to further suppress harmonic torque and further reduce torque ripple. Can do.
As a result, the rotor 12 in the stator 11 can be produced at low cost and can be driven to rotate with high energy density and high quality.

Here, in this embodiment, the electric rotating machine 10 having a configuration of an 8-pole 48-slot motor will be described as an example. However, the present invention is not limited to this. If the structure has a number of slots per phase per pole q = 2, The present invention can be preferably applied as it is. For example, it can also be applied to a motor structure of 6 poles, 36 slots, 4 poles, 24 slots, and 10 poles and 60 slots.
The scope of the present invention is not limited to the illustrated and described exemplary embodiments, but includes all embodiments that provide the same effects as those intended by the present invention. Further, the scope of the invention is not limited to the combinations of features of the invention defined by the claims, but may be defined by any desired combination of particular features among all the disclosed features. .

  Although one embodiment of the present invention has been described so far, it is needless to say that the present invention is not limited to the above-described embodiment, and may be implemented in various forms within the scope of the technical idea.

10 Electric rotating machine (IPM type)
11 Stator 12 Rotor 12a Outer peripheral surface 13 Rotating drive shaft 15 Stator teeth 15a Inner peripheral surface 16 Permanent magnet 16a Corner portion 17 V-shaped space 17b, 17c Flux barrier 18 Slot 20 Center bridge 21 Center groove 21a Groove bottom A1 Magnetic pole outer peripheral side Small area A2 Magnetic pole outer side area B d-axis side area G Gap MP0, MP1 to MP3, Mr1, Mr2 Magnetic path R1 Outer radius R4 Separation distance Ψm Magnet flux Ψr Armature flux Ψs Composite flux θa Outer opening angle θb Inner opening angle

Claims (2)

An electric rotating machine comprising: a rotor in which a plurality of pairs of permanent magnets are embedded; and a stator in which a coil is accommodated in a slot formed between a plurality of teeth facing the rotor,
The pair of permanent magnets are arranged in a V-shape that opens toward the outer peripheral surface of the rotor, each magnetic pole pair of the permanent magnets is formed, and fluxes barrier is formed,
The flux barrier is formed between the pair of permanent magnets, and is radially inward of the rotor with respect to the stator side end of the stator side magnetic pole surface facing the stator of the permanent magnet. It is composed of a space formed on the side,
The space extends to the inner side in the radial direction of the rotor from the anti-stator side magnetic pole surface opposite to the stator side magnetic pole surface,
An adjustment groove is provided on the d-axis of the outer peripheral surface of the rotor;
The number of poles of the rotor is P,
The length of the permanent magnet is Wpm,
An outer radius from the rotor axis to the outer peripheral surface is R1,
The length from the axis of the rotor to the groove bottom of the adjustment groove is R4,
1.38 <(P × Wpm) / R1 <1.75
And 0.98 ≦ R4 / R1 <1.0
Formed in size and shape to satisfy the relation,
The circumferential width of the adjustment groove is set to be equal to or larger than the circumferential width of the facing surface of the teeth facing the rotor and equal to or smaller than the width between the teeth adjacent in the circumferential direction across the one tooth. An IPM type electric rotating machine characterized by that. The adjustment groove is
The outer opening angle at the outer peripheral surface of the rotor around the axis of the rotor is θa,
An inner opening angle of the groove bottom around the axis of the rotor is θb,
The opening width of the slot on the outer peripheral surface side of the rotor is SO,
The facing width of the teeth with respect to the outer peripheral surface of the rotor is TB,
The width of the tip portion inside the facing width TB of the teeth is TW,
The air gap width between the rotor and the teeth is AG,
Outside angle 1 ≦ θa (electrical angle) ≦ outside angle 2,
Inner angle 1 ≦ θb (electrical angle) ≦ inner angle 2,
If
Outside angle 1 = 2 × tan ⁻¹ ((TB / 2) / (R1 + AG)),
Outer angle 2 = Inner angle 2 = 2 × tan ⁻¹ ((SO + (TB / 2)) / (R1 + AG)),
Interior angle 1 = 0 °,
And,
TW ≦ TB,
The IPM type electric rotating machine according to claim 1, wherein the electric shape rotating machine is formed in a size and shape that satisfies the above relationship.

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