JP6225849B2 - Control device for rotating electrical machine - Google Patents

Description

  The present invention relates to a control device applied to a rotating electrical machine electrically connected to a power conversion circuit having a switching element.

  As this type of control device, for example, as can be seen in Patent Document 1 below, there is known a device that reduces the harmonic current flowing through the motor in order to reduce the torque fluctuation of the motor. Specifically, the harmonic current for reducing the harmonic current that causes the torque fluctuation or the like is superimposed on the fundamental wave current that flows through the electric motor in the fixed coordinate system by turning on and off the switching elements constituting the inverter. Here, the on / off operation of the switching element for superimposing the harmonic current is performed by PWM control based on the magnitude comparison between the command voltage calculated by the current feedback control and the carrier.

Japanese Patent No. 3852289

  By the way, as a switching element on / off operation method, there is a method using a pulse pattern in addition to the above-described method using PWM control. Specifically, the control device includes storage means for storing an on / off operation command of the switching element related to the electrical angle of the electric motor, for example, for each amplitude of the output voltage vector of the inverter. In such a configuration, first, the phase of the output voltage vector of the inverter in the primary rotating coordinate system is set as an operation amount for feedback control of the control amount of the electric motor to the target value. Here, the primary rotating coordinate system is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the fundamental current flowing in the motor in the fixed coordinate system. Then, the on / off operation command selected based on the amplitude of the output voltage vector is output to the switching element by shifting the phase with respect to the detected value of the electrical angle, thereby turning on / off the switching element.

  Here, also in the technique using the pulse pattern, it is desired to apply a technique of superimposing the harmonic current in order to reduce the harmonic current that causes the torque fluctuation of the motor.

  The main object of the present invention is to provide a control device for a rotating electrical machine that can suitably reduce the harmonic current flowing in the rotating electrical machine.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

The present invention is applied to a three-phase rotating electrical machine (10) electrically connected to a power conversion circuit (20) having switching elements (SUp to SWn), and a fundamental wave current flowing through the rotating electrical machine in a fixed coordinate system. The power conversion circuit in the primary rotary coordinate system is used as an operation amount for feedback control of the control amount of the rotating electrical machine to its target value. Voltage phase setting means (30d) for setting the phase of the output voltage vector, and voltage amplitude setting means (30h, 30h) for setting the amplitude of the output voltage vector as an operation amount for controlling the control amount to the target value. 30j, 30k, 32) and an on / off operation finger of the switching element associated with the electrical angle of the rotating electrical machine for each amplitude parameter including the amplitude. Storage means (33U to 33W) for storing the voltage, the on / off operation command selected based on the amplitude parameter including the amplitude set by the voltage amplitude setting means, the voltage based on the detected value of the electrical angle. Based on the operation means (33U to 33W) for turning on and off the switching element and the phase current detection value flowing in the rotating electrical machine by shifting the phase set by the phase setting means and outputting to the switching element, Extraction means (31a, 31b) for extracting the harmonic current flowing in the rotating electrical machine, “1 ± 6n” (n is an integer other than 0) is an integer k, the electrical angle of the rotating electrical machine is θe, In a fixed coordinate system, the harmonic current of each phase that fluctuates at an angular velocity k times the electrical angular velocity of the rotating electrical machine is represented by Irk × cos (k × θe + φk), Irk × co. s (k × θe-2 / 3π + φk), in the case of the Irk × cos (k × θe + 2 / 3π + φk), based on the harmonic currents extracted by the extraction means, before Symbol electrical angular speed in the fixed coordinate system A harmonic signal for reducing a harmonic current of each phase that fluctuates at an angular velocity of k times, and a harmonic signal that fluctuates at an angular velocity that is “k−1” times the electrical angular velocity in the primary rotational coordinate system. Generating means (31i, 31j; 31i, 31j, 31s, 31t) used in the operation means for reducing harmonic currents fluctuating at an angular velocity k times the electrical angular velocity in the fixed coordinate system. Superimposing means (31k, 31l) for superimposing the harmonic signal generated by the generating means on at least one of the amplitude and the phase.

  When a harmonic signal that fluctuates at an angular velocity “k−1” times the electrical angular velocity is superimposed on at least one of the amplitude and phase of the output voltage vector used in the operating means, the electrical angular velocity k is applied to the rotating electrical machine in the fixed coordinate system. Double harmonic current (hereinafter referred to as kth harmonic current) flows. For this reason, it is possible to reduce the k-th order harmonic current flowing in the rotating electrical machine by superimposing a harmonic signal that cancels the k-th order harmonic current to be reduced on at least one of the amplitude and the phase. it can.

  Therefore, in the above invention, the harmonic current flowing through the rotating electrical machine is extracted based on the detected phase current value flowing through the rotating electrical machine. The extracted harmonic current becomes information for grasping the harmonic current to be reduced. And the said harmonic signal is produced | generated based on the extracted harmonic current. By superimposing the harmonic signal thus generated on at least one of the amplitude and phase, the k-th order harmonic current flowing in the rotating electrical machine can be suitably reduced.

1 is an overall configuration diagram of a motor control system according to a first embodiment. The block diagram of motor control. The figure for demonstrating (lambda) axis | shaft. The figure which shows the calculation method of the angle which d axis | shaft and (lambda) axis | shaft make. The figure for demonstrating (lambda) axis | shaft. The figure which shows the calculation method of (lambda) axis current. The figure which shows the relationship between a fundamental wave component and a harmonic component. The block diagram of a harmonic processing part. The block diagram of the harmonic signal production | generation part concerning 2nd Embodiment.

(First embodiment)
DESCRIPTION OF EMBODIMENTS Hereinafter, a first embodiment in which a control device for a rotating electrical machine according to the present invention is applied to a vehicle (for example, an electric vehicle or a hybrid vehicle) including a multi-phase rotating machine (three-phase rotating electrical machine) as an in-vehicle main unit is described with reference to the drawings. However, it explains.

  As shown in FIG. 1, the motor control system includes a motor generator 10, an inverter 20 as a “power conversion circuit”, and a control device 30 that controls the motor generator 10. In the present embodiment, the motor generator 10 is an in-vehicle main machine and is connected to drive wheels (not shown). In the present embodiment, an IPMSM that is a salient pole machine is used as the motor generator 10.

  The motor generator 10 is connected to a battery 22 as a DC power source via an inverter 20. The output voltage of the battery 22 is, for example, 100 V or more. A smoothing capacitor 24 that smoothes the input voltage of the inverter 20 is provided between the battery 22 and the inverter 20.

  The inverter 20 includes a series connection body of upper arm switches SUp, SVp, SWp and lower arm switches SUn, SVn, SWn. The U-phase of the motor generator 10 is connected to the connection point of the U-phase upper and lower arm switches SUp and SUn, and the V-phase of the motor generator 10 is connected to the connection point of the V-phase upper and lower arm switches SVp and SVn. The W phase of the motor generator 10 is connected to the connection point between the upper and lower arm switches SWp and SWn of the W phase. Incidentally, in the present embodiment, as each of the switches SUp to SWn, a voltage-controlled semiconductor switching element is used, and more specifically, an IGBT is used. And each freewheel diode DUp-DWn is connected to each switch SUp-SWn in antiparallel.

  The motor control system further includes phase current detection means, voltage detection means, and rotation angle detection means. Specifically, the phase current detection means includes a V-phase current sensor 42V that detects a current flowing in the V-phase of motor generator 10 and a W-phase current sensor 42W that detects a current flowing in the W-phase. The voltage detection means includes a voltage sensor 44 that detects an input voltage of the inverter 20 (a DC voltage output from the battery 22). The rotation angle detection means includes a rotation angle sensor 46 (for example, a resolver) that detects the rotation angle (electrical angle θe) of the motor generator 10.

  The control device 30 is mainly composed of a microcomputer, and operates the inverter 20 to feedback control the control amount (torque in this embodiment) of the motor generator 10 to a target value (hereinafter, target torque Trq *). Specifically, the control device 30 generates and generates operation signals gUp to gWn corresponding to the switches SUp to SWn based on the detection values of the various sensors in order to turn on and off the switches SUp to SWn constituting the inverter 20. The operated operation signals gUp to gWn are output to the switches SUp to SWn. Here, the upper arm operation signals gUp, gVp, and gWp and the corresponding lower arm operation signals gUn, gVn, and gWn are complementary signals (inverted signals). That is, the upper arm switch and the corresponding lower arm switch are alternately turned on. The target torque Trq * is, for example, a control device provided outside the control device 30, and is output from a control device higher than the control device 30.

  Next, torque control of the motor generator 10 executed by the control device 30 will be described using FIG. This control includes phase control and amplitude control.

  First, phase control will be described. The two-phase converter 30a (corresponding to “two-phase converter”) includes a V-phase current IV detected by the V-phase current sensor 42V, a W-phase current IW detected by the W-phase current sensor 42W, and a rotation angle sensor 46. The U-phase current IU, V-phase current IV, and W-phase current IW in the three-phase fixed coordinate system are converted into the d-axis currents Idr and q in the primary rotation coordinate system (dq coordinate system) based on the electrical angle θe detected by It converts into shaft current Iqr. Here, the primary rotating coordinate system is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the fundamental wave component of the phase current in the three-phase fixed coordinate system. The U-phase current IU may be calculated from the V-phase current IV and the W-phase current IW based on Kirchhoff's law.

  Torque estimator 30b calculates estimated torque Te of motor generator 10 based on the d and q axis currents Idr and Iqr output from two-phase converter 30a. Here, the estimated torque Te may be calculated using a map in which the d-axis current Idr and the q-axis current Iqr are associated with the estimated torque Te, or may be calculated using a model formula.

  The torque deviation calculation unit 30c calculates the torque deviation ΔT by subtracting the estimated torque Te calculated by the torque estimator 30b from the target torque Trq *. The control device 30 includes a filter (low-pass filter) that removes high-frequency components from the estimated torque Te calculated by the torque estimator 30b, and subtracts the estimated torque Te from which high-frequency components have been removed from the target torque Trq *. The torque deviation ΔT may be calculated.

  The phase setting unit 30d (corresponding to “voltage phase setting means”) uses a voltage as an operation amount for performing feedback control of the estimated torque Te to the target torque Trq * based on the torque deviation ΔT calculated by the torque deviation calculation unit 30c. The phase φ is calculated. Specifically, the voltage phase φ is calculated by proportional integral control using the torque deviation ΔT as an input. In the present embodiment, the voltage phase φ is defined with the positive direction of the d-axis as a reference, and the counterclockwise direction from this reference (the direction rotating from the positive direction of the d-axis to the positive direction of the q-axis) is defined as the positive direction. Has been. Therefore, when the estimated torque Te is insufficient with respect to the target torque Trq *, the voltage phase φ is increased (advanced), and when the estimated torque Te is excessive with respect to the target torque Trq *, the voltage The phase φ is decreased (retarded).

  The harmonic processing unit 31 superimposes and outputs the phase harmonic signal φr on the voltage phase φ calculated by the phase setting unit 30d. The harmonic processing unit 31 will be described in detail later.

  The electrical angle addition unit 30e adds the electrical angle θe to the output value δ of the harmonic processing unit 31. The first shift unit 30f outputs a value obtained by subtracting “2π / 3” from the output value “θe + δ” of the electrical angle adder 30e. The second shift unit 30g outputs a value obtained by adding “2π / 3” to the output value “θe + δ” of the electrical angle addition unit 30e. Thereby, the output value “θe + δ” of the electrical angle adder 30e, the output value “θe + δ-2π / 3” of the first shift unit 30f, and the output value “θe + δ + 2π / 3” of the second shift unit 30g are mutually equal to the electrical angle. Thus, the signal is shifted by “2π / 3” (120 °). In the present embodiment, hereinafter, the output value of the electrical angle adder 30e is referred to as a U-phase reference angle θU, the output values of the first and second shift units 30f and 30g are referred to as V, and the W-phase reference angles θV and θW as I will call it.

  Next, amplitude control will be described. The command voltage setting unit 30h calculates the standardized voltage amplitude “Vn / ω” using the target torque Trq * as an input. Here, the normalized voltage amplitude “Vn / ω” is a value obtained by dividing the command amplitude (hereinafter, voltage amplitude Vn) of the output voltage vector of the inverter 20 in the primary rotating coordinate system by the electrical angular velocity ω. The voltage amplitude Vn is defined as the square root of the sum of the square value of the d-axis component Vd and the square value of the q-axis component Vq of the output voltage vector. In the present embodiment, the normalized voltage amplitude is calculated using a map in which the target torque Trq * and the normalized voltage amplitude are related.

  The speed calculation unit 30 i calculates the electrical angular speed ω of the motor generator 10 based on the electrical angle θe detected by the rotation angle sensor 46. The speed multiplier 30j calculates the voltage amplitude Vn by multiplying the normalized voltage amplitude “Vn / ω” by the electrical angular speed ω. Voltage amplitude Vn is an operation amount for performing feedforward control of torque of motor generator 10 to target torque Trq *.

  The correction unit 30k corrects the voltage amplitude Vn by adding the amplitude correction amount ΔV calculated by the correction amount calculation unit 32 to the voltage amplitude Vn output from the speed multiplication unit 30j. The correction amount calculation unit 32 will be described in detail later. In the present embodiment, the command voltage setting unit 30h, the speed multiplication unit 30j, the correction unit 30k, and the correction amount calculation unit 32 correspond to “voltage amplitude setting means”.

  The voltage amplitude “Vn + ΔV” corrected by the correction unit 30 k is input to the harmonic processing unit 31. The harmonic processing unit 31 superimposes and outputs the amplitude harmonic signal Vr on the voltage amplitude “Vn + ΔV”.

  The U, V, and W phase operation signal generation units 33U, 33V, and 33W (corresponding to “operation means”) are the voltage amplitude Vk on which the amplitude harmonic signal Vr is superimposed in the harmonic processing unit 31, and U, V, and W Based on the phase reference angles θU, θV, θW and the input voltage VINV detected by the voltage sensor 44, the operation signals gUp to gWn are generated and output to the switches SUp to Swn. In the present embodiment, each operation signal is generated as follows.

  Each of the operation signal generators 33U to 33W includes a memory (storage means). In the memory, for each modulation factor M, an operation signal waveform (pulse pattern) for one rotation period “2π” of the electrical angle is stored as map data. In the present embodiment, the operation signal waveform can be applied with a fundamental wave voltage for causing a fundamental wave current to flow in each phase winding of the motor generator 10 in one electrical angle cycle. In the present embodiment, in each of the U to W phases, the stored operation signal waveforms are the ON operation period of the upper arm switches SUp, SVp, SWp and the ON operation period of the lower arm switches SUn, SVn, SWn. Is a half-wave waveform. This is a setting for making the output voltage of the inverter 20 balanced in one rotation cycle of the electrical angle. Further, in the present embodiment, in each of the U to W phases, the operation signal waveform is symmetrical with respect to the center (180 °) of one rotation period of the electrical angle. Specifically, the logical values of a pair of timings equidistant from the center are reversed. In the present embodiment, the logic “H” corresponds to the ON operation command for the upper arm switches SUp, SVp, SWp, and the logic “L” corresponds to the ON operation command for the lower arm switches SUn, SVn, SWn. The operation signal waveforms stored in the memory are common to the operation signal generation units 33U to 33W for each modulation factor M.

  Each of the operation signal generation units 33U to 33W first calculates the modulation factor M based on the input voltage VINV and the voltage amplitude Vk output from the harmonic processing unit 31. In the present embodiment, the modulation factor M is a value obtained by standardizing the voltage amplitude Vk with the input voltage VINV. More specifically, the modulation factor M is a value obtained by dividing a value obtained by dividing the voltage amplitude Vk by “½” of the input voltage VINV by “√ (1.5)”. Each of the operation signal generation units 33U to 33W selects a corresponding operation signal waveform based on the calculated modulation factor M. When the operation signal waveform is selected in this way, each of the operation signal generation units 33U to 33W generates an operation signal by setting the output timing of the waveform based on the voltage phase δ output from the harmonic processing unit 31. . That is, for each of the U to W phases, the same operation signal waveform is output with an electrical angle shifted by 120 ° from each other. As a result, sinusoidal phase currents whose phases are shifted from each other by 120 ° in electrical angle can flow.

  Next, the correction amount calculation unit 32 and the harmonic processing unit 31 will be described.

  First, the design method of the correction amount calculation unit 32 will be described with reference to FIGS.

  The voltage equation of the permanent magnet synchronous machine is expressed by the following equation (eq1).

上式（ｅｑ１）において、「ｐ」は微分演算子を示し、「Ｒ」は電機子巻線抵抗を示し、「Ｌｄ」，「Ｌｑ」はｄ，ｑ軸インダクタンスを示し、「ψ」は永久磁石の電機子鎖交磁束の実効値を示す。上式（ｅｑ１）において、モータジェネレータ１０の回転速度が一定となる定常状態を想定し、過渡現象を無視するとの条件を課すと、「ｐ＝０」となる。また、上式（ｅｑ１）に、モータジェネレータ１０の回転速度が十分高く、「Ｒ＜＜ω・Ｌｄ」，「Ｒ＜＜ω・Ｌｑ」の関係が成立するとの条件を課す。以上から、上式（ｅｑ１）は下式（ｅｑ２）のように表される。

In the above equation (eq1), “p” indicates a differential operator, “R” indicates an armature winding resistance, “Ld” and “Lq” indicate d and q axis inductances, and “ψ” is permanent. The effective value of the armature flux linkage of the magnet is shown. In the above equation (eq1), assuming a steady state in which the rotation speed of the motor generator 10 is constant and imposing a condition that the transient phenomenon is ignored, “p = 0” is obtained. Further, a condition that the rotational speed of the motor generator 10 is sufficiently high and the relationship of “R << ω · Ld” and “R << ω · Lq” is satisfied is given to the above equation (eq1). From the above, the above equation (eq1) is expressed as the following equation (eq2).

ｄ，ｑ軸電圧Ｖｄ，Ｖｑと、電圧位相φ及び電圧振幅Ｖｎとの関係は、下式（ｅｑ３）で表される。

The relationship between the d and q axis voltages Vd and Vq, the voltage phase φ, and the voltage amplitude Vn is expressed by the following equation (eq3).

ここで、電圧位相φが微小量Δφだけ変化した場合における電圧方程式は、上式（ｅｑ２），（ｅｑ３）を用いると、下式（ｅｑ４）で表される。

Here, the voltage equation when the voltage phase φ is changed by a minute amount Δφ is expressed by the following equation (eq4) using the above equations (eq2) and (eq3).

上式（ｅｑ４）から上式（ｅｑ２）を減算すると、下式（ｅｑ５）が導かれる。

When the above equation (eq2) is subtracted from the above equation (eq4), the following equation (eq5) is derived.

上式（ｅｑ５）において、右辺の「Ｉｄφ−Ｉｄ」がｄ軸電流変化量ΔＩｄφであり、「Ｉｑφ−Ｉｑ」がｑ軸電流変化量ΔＩｑφである。上式（ｅｑ５）を各電流変化量ΔＩｄφ，ΔＩｑφについて解くと、下式（ｅｑ６）が導かれる。

In the above equation (eq5), “Idφ−Id” on the right side is the d-axis current change amount ΔIdφ, and “Iqφ−Iq” is the q-axis current change amount ΔIqφ. When the above equation (eq5) is solved for each current change amount ΔIdφ, ΔIqφ, the following equation (eq6) is derived.

図３に、１次回転座標系における電圧ベクトルＶｎｖｔ及び電流ベクトルＩｎｖｔを示す。ここで、電流ベクトルは、ｄ軸電流の２乗値及びｑ軸電流の２乗値の和の平方根として定義される。図３には、電圧位相φが微小量Δφだけ変化した場合の電流ベクトルＩｎｖｔの変化分を「ΔＩφ」にて示した。また、電圧振幅が微小量ΔＶｎだけ変化した場合の電流ベクトルＩｎｖｔの変化分を「ΔＩｖｎ」にて示した。この電流ベクトルＩｎｖｔの変化分を拡大した図を図４として示す。上式（ｅｑ６）より、電圧位相φが微小変化した場合において、ｄ軸に対する電流ベクトルＩｎｖｔの変化方向αは、下式（ｅｑ７）で表される。

FIG. 3 shows the voltage vector Vnvt and the current vector Invt in the primary rotating coordinate system. Here, the current vector is defined as the square root of the sum of the square value of the d-axis current and the square value of the q-axis current. In FIG. 3, the change amount of the current vector Invt when the voltage phase φ is changed by a minute amount Δφ is indicated by “ΔIφ”. Further, a change amount of the current vector Invt when the voltage amplitude is changed by a minute amount ΔVn is indicated by “ΔIvn”. FIG. 4 shows an enlarged view of the change in the current vector Invt. From the above equation (eq6), when the voltage phase φ is slightly changed, the change direction α of the current vector Invt with respect to the d axis is represented by the following equation (eq7).

上式（ｅｑ７）のアークタンジェント演算により、例えば、変化方向αを「−π〜＋π」の間で算出することができる。特に本実施形態では、上式（ｅｑ７）の右辺において、括弧内の分母が０となってかつ分子が正の値となる場合、変化方向αを「π／２」として算出する。一方、上式（ｅｑ７）の右辺において、括弧内の分母が０となってかつ分子が負の値となる場合、変化方向αを「−π／２」として算出する。ここで、図５には、電流ベクトルＩｎｖｔの変化方向と直交する方向に延びる座標軸をλ軸として示している。電圧振幅Ｖｎが微小量ΔＶｎだけ変化した場合の電流ベクトルの変化分ΔＩｖｎのうち、λ軸成分（すなわち、上記変化分ΔＩｖｎをλ軸に写像した成分）は、電圧位相φの変化の影響を受けない電流である。本実施形態では、この電流をλ軸電流Ｉλとして振幅補正量ΔＶの算出に用いる。λ軸電流Ｉλを用いることにより、振幅制御と位相制御との干渉を抑制することができる。ここで、λ軸を設定するために必要なパラメータであるｄ軸とλ軸とのなす角度λは、下式（ｅｑ８）で表される。

For example, the change direction α can be calculated between “−π to + π” by the arctangent calculation of the above equation (eq7). Particularly in the present embodiment, when the denominator in the parenthesis is 0 and the numerator is a positive value on the right side of the above equation (eq7), the change direction α is calculated as “π / 2”. On the other hand, in the right side of the above equation (eq7), when the denominator in the parenthesis is 0 and the numerator is a negative value, the change direction α is calculated as “−π / 2”. Here, in FIG. 5, a coordinate axis extending in a direction orthogonal to the changing direction of the current vector Invt is shown as a λ axis. Of the change ΔIvn of the current vector when the voltage amplitude Vn changes by a minute amount ΔVn, the λ-axis component (that is, the component obtained by mapping the change ΔIvn to the λ-axis) is affected by the change of the voltage phase φ. There is no current. In the present embodiment, this current is used as the λ-axis current Iλ for calculating the amplitude correction amount ΔV. By using the λ-axis current Iλ, interference between amplitude control and phase control can be suppressed. Here, the angle λ formed by the d-axis and the λ-axis, which is a parameter necessary for setting the λ-axis, is expressed by the following equation (eq8).

以上を踏まえ、先の図２に戻り、補正量算出部３２について説明する。

Based on the above, returning to FIG. 2, the correction amount calculation unit 32 will be described.

  The λ-axis setting unit 32a is based on the d and q-axis inductances Ld and Lq and the voltage phase φ output from the phase setting unit 30d, and the angle formed between the d-axis and the λ-axis based on the above equation (eq8). λ is calculated. The λ-axis setting unit 32a corresponds to a non-interacting axis setting unit that sets a non-interacting coordinate axis (λ axis) in which the change of the current vector Invt is made non-interfering in the primary rotation coordinate system. To do. The λ axis set in the λ axis setting unit 32a changes each time the driving state of the motor generator 10 changes.

  The command current setting unit 32b sets the d and q axis command currents Id * and Iq * based on the target torque Trq *. In the present embodiment, currents for realizing minimum current / maximum torque control are set as d and q-axis command currents Id * and Iq *. The minimum current / maximum torque control is described, for example, on page 23 of “Design and control of embedded magnet synchronous motor: Yoji Takeda et al., 3 others, Ohmsha, April 20, 2006, first edition”. Has been.

  Based on the command currents Id * and Iq * output from the command current setting unit 32b and the angle λ output from the λ axis setting unit 32a, the λ-axis command current calculation unit 32c is based on the following equation (eq9). Then, the λ-axis command current Iλ * is calculated (see FIG. 6).

ここで、図６には、現在の指令電流ベクトルを「Ｉｎ＊」にて示し、現在の電流ベクトルを「Ｉｎｖｔ」にて示した。

Here, in FIG. 6, the current command current vector is indicated by “In *”, and the current current vector is indicated by “Invt”.

  The λ-axis actual current calculation unit 32d is based on the following equation (eq10) based on the d and q-axis currents Idr and Iqr output from the two-phase conversion unit 30a and the angle λ output from the λ-axis setting unit 32a. Then, the λ-axis current Iλr is calculated (see FIG. 6).

λ軸がモータジェネレータ１０の駆動状態の変化に伴って変化するため、λ軸電流Ｉλｒ及びλ軸指令電流Ｉλ＊もモータジェネレータ１０の駆動状態の変化に伴って都度変化することとなる。

Since the λ axis changes as the driving state of the motor generator 10 changes, the λ axis current Iλr and the λ axis command current Iλ * also change as the driving state of the motor generator 10 changes.

  The current deviation calculator 32e calculates the current deviation ΔIλ by subtracting the λ-axis current Iλr output from the λ-axis actual current calculator 32d from the λ-axis command current Iλ *. The control device 30 includes a filter (low-pass filter) that removes high-frequency components from the λ-axis current Iλr output from the λ-axis actual current calculation unit 32d, and the λ-axis current Iλr from which the high-frequency components have been removed is used as the λ-axis command current. The current deviation ΔIλ may be calculated by subtracting from Iλ *.

  Based on the current deviation ΔIλ output from the current deviation calculation unit 32e, the amplitude correction amount calculation unit 32f performs an operation amount for feedback control of the λ-axis current Iλr to the λ-axis command current Iλ * (in other words, the estimated torque Te Is calculated as an amplitude manipulated variable, and an amplitude correction amount ΔV as an amplitude manipulated variable is calculated. Specifically, the amplitude correction amount ΔV is calculated by proportional-integral control using the current deviation ΔIλ as an input.

  According to such a configuration, since the amplitude correction amount ΔV is calculated based on the λ-axis current Iλr, interference between the amplitude control and the phase control is suppressed. Therefore, the currents Idr and Iqr can be controlled to the command currents Id * and Iq * with high accuracy. Thereby, the responsiveness of the feedback control in the amplitude control can be improved to the same level as the responsiveness of the feedback control in the phase control. Therefore, both high torque controllability and high current controllability can be maintained even when a disturbance occurs or a transient state occurs.

  Next, the harmonic processing unit 31 will be described.

  First, the reason why the harmonic processing unit 31 is provided in the control device 30 will be described.

  When the magnetic characteristics (for example, d and q axis inductances and induced voltage constants) of the motor generator 10 are ideal characteristics, sine wave phase currents that are shifted from each other by 120 ° in electrical angle can be supplied to each phase. A constant torque that does not contain harmonic components can be obtained. However, with an actual motor generator 10, it is difficult to obtain ideal magnetic characteristics. For this reason, as shown in the following equation (eq11), due to the magnetic characteristics deviating from the ideal one, each phase current IU to IW includes a harmonic current.

上式（ｅｑ１１）において、右辺第１項は、振幅Ｉｒ１及び位相φ１とする基本波電流を示し、右辺第２項は、振幅Ｉｒｋ及び位相φｋとする高調波電流を示している。ここで、右辺第２項の「ｋ」は、「１±６ｎ」（ｎは０以外の整数）である。なお、以降、電気角速度ωのｋ倍の角速度で変動する高調波電流をｋ次の高調波電流と称すこととする。

In the above equation (eq11), the first term on the right side indicates the fundamental wave current with the amplitude Ir1 and the phase φ1, and the second term on the right side indicates the harmonic current with the amplitude Irk and the phase φk. Here, “k” in the second term on the right side is “1 ± 6n” (n is an integer other than 0). Hereinafter, a harmonic current that fluctuates at an angular velocity that is k times the electrical angular velocity ω will be referred to as a k-th harmonic current.

  When harmonic current flows to the motor generator 10, there is a concern that torque fluctuation and loss (for example, iron loss) increase. Here, what converted the phase current containing a harmonic current into the d and q axis current of the primary rotation coordinate system is expressed by the following equation (eq12). In FIG. 7, the amplitude of the combined vector of the fundamental current vector and the harmonic current vector in the primary rotation coordinate system is indicated by “It”, and the phase is indicated by “φt”.

上式（ｅｑ１２）は、固定座標系において相電流にｋ次の高調波電流が含まれると、１次回転座標系において「ｋ−１」次の高調波電流が表れることを示している。すなわち、上式（ｅｑ１２）は、１次回転座標系における高調波電流であって、固定座標系におけるｋ次の高調波電流を低減可能な「ｋ−１」次の高調波電流が存在することを示している。ここで、「ｋ−１」次の高調波電流は、電気角速度ωの「ｋ−１」倍の角速度で変動する高調波信号を電圧振幅「Ｖｎ＋ΔＶ」及び電圧位相φのそれぞれに重畳することにより流すことができる。

The above equation (eq12) indicates that if the k-order harmonic current is included in the phase current in the fixed coordinate system, the “k−1” -order harmonic current appears in the primary rotating coordinate system. That is, the above equation (eq12) is a harmonic current in the primary rotation coordinate system, and there exists a “k−1” -order harmonic current that can reduce the k-order harmonic current in the fixed coordinate system. Is shown. Here, the “k−1” -order harmonic current is obtained by superimposing a harmonic signal varying at an angular velocity “k−1” times the electrical angular velocity ω on each of the voltage amplitude “Vn + ΔV” and the voltage phase φ. It can flow.

  In particular, in the present embodiment, since the 6th harmonic current in the primary rotating coordinate system is remarkable, the 6th harmonic current is targeted for reduction. Here, from the relationship of “k = 1 ± 6n”, in order to cancel the sixth-order harmonic current, in the fixed coordinate system, k = −5th-order harmonic current or k = 7 What is necessary is just to superimpose the 7th-order harmonic current made on the phase current. When “k> 0”, the waveform of the harmonic current of each phase changes in the order of the U, V, and W phases as the electrical angle θe progresses. On the other hand, in the case of “k <0”, the waveform of the harmonic current of each phase changes in the reverse order to that in the case of “k> 0” with the progress of the electrical angle θe.

  FIG. 8 shows a block diagram of the harmonic processing unit 31.

  The d-axis deviation calculation unit 31a is a d-axis current deviation ΔId (“d” that is a deviation between the d-axis command current Id * output from the command current setting unit 32b and the d-axis current Idr output from the two-phase conversion unit 30a. Equivalent to “Axial harmonic component”). Specifically, the d-axis current deviation ΔId is calculated by subtracting the d-axis current Idr from the d-axis command current Id *. The q-axis deviation calculating unit 31b is a q-axis current deviation ΔIq (“q” that is a deviation between the q-axis command current Iq * output from the command current setting unit 32b and the q-axis current Iqr output from the two-phase conversion unit 30a. Equivalent to “Axial harmonic component”). Specifically, the q-axis current deviation ΔIq is calculated by subtracting the q-axis current Iqr from the q-axis command current Iq *. In the present embodiment, each of the deviation calculation units 31a and 31b corresponds to “extraction means”.

  The high-order current converter 31c (corresponding to “high-order current calculation means”) uses the following equations (eq13) and (eq14) to convert the current deviations ΔId and ΔIq in the primary rotation coordinate system into high-order rotation coordinates. Conversion into d and q-axis higher order currents Idkr and Iqkr in the system (dk-qk). Here, the high-order rotating coordinate system is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the k-th harmonic current in the fixed coordinate system. The high-order rotational coordinate system is defined by a dk axis and a qk axis that are orthogonal to each other. In this embodiment, the phase φk of the harmonic current is based on the positive direction of the dk axis and is counterclockwise from this reference (the direction rotating from the positive direction of the dk axis to the positive direction of the qk axis). Is defined as the positive direction.

上式（ｅｑ１３），（ｅｑ１４）において、右辺の「（ｋ−１）θｅ」は、１次回転座標系の基準軸（ｄ軸）と高次回転座標系の基準軸（ｄｋ軸）との位相差を示す。

In the above equations (eq13) and (eq14), “(k−1) θe” on the right side is the difference between the reference axis (d axis) of the primary rotation coordinate system and the reference axis (dk axis) of the higher order rotation coordinate system. Indicates the phase difference.

  The target value setting unit 31d (corresponding to “variable setting means”) variably sets the d-axis target high-order current Idk * and the q-axis target high-order current Iqk * based on the target torque Trq * and the electrical angular velocity ω. . Each target higher-order current Idk *, Iqk * is set from the viewpoint of suppressing torque fluctuation and iron loss. Here, for example, the target high-order currents Idk * and Iqk * may be set larger as the target torque Trq * is larger. The target high-order currents Idk * and Iqk * may be set using, for example, a map or a mathematical formula in which the target torque Trq *, the electrical angular velocity ω, and the target high-order currents Idk * and Iqk * are related. Good.

  The d-axis high-order deviation calculation unit 31e calculates a d-axis high-order deviation Δdk that is a deviation between the d-axis target high-order current Idk * and the d-axis high-order current Idkr. Specifically, the d-axis higher-order deviation Δdk is calculated by subtracting the d-axis higher-order current Idkr from the d-axis target higher-order current Idk *. The q-axis higher-order deviation calculating unit 31f calculates a q-axis higher-order deviation Δqk that is a deviation between the q-axis target higher-order current Iqk * and the q-axis higher-order current Iqkr. Specifically, the q-axis higher-order deviation Δqk is calculated by subtracting the q-axis higher-order current Iqkr from the q-axis target higher-order current Iqk *.

  The d-axis feedback control unit 31g (corresponding to “d-axis operation amount calculation means”) converts the d-axis higher-order current Idkr to the d-axis based on the d-axis higher-order deviation Δdk output from the d-axis higher-order deviation calculation unit 31e. A d-axis feedback current Idkf (corresponding to “d-axis operation amount”) is calculated as an operation amount for feedback control to the target higher-order current Idk *. Specifically, the d-axis feedback current Idkf is calculated by proportional-integral control using the d-axis higher-order deviation Δdk as an input.

  The q-axis feedback control unit 31h (corresponding to “q-axis manipulated variable calculation means”) converts the q-axis higher-order current Iqkr to the q-axis based on the q-axis higher-order deviation Δqk output from the q-axis higher-order deviation calculation unit 31f. A q-axis feedback current Iqkf (corresponding to “q-axis operation amount”) is calculated as an operation amount for performing feedback control to the target higher-order current Iqk *. Specifically, the q-axis feedback current Iqkf is calculated by proportional-integral control using the q-axis higher-order deviation Δqk as an input.

  The primary current conversion unit 31i converts the d and q-axis feedback currents Idkf and Iqkf in the higher-order rotational coordinate system into d and q-axis harmonic currents Id1f and Iq1f in the primary rotational coordinate system using the following equation (eq15). Convert.

電圧変換部３１ｊは、ｄ，ｑ軸高調波電流Ｉｄ１ｆ，Ｉｑ１ｆに基づき、振幅高調波信号Ｖｒと、位相高調波信号φｒとを算出する。詳しくは、まず、下式（ｅｑ１６）を用いて、ｄ，ｑ軸高調波電流Ｉｄ１ｆ，Ｉｑ１ｆをｄ，ｑ軸高調波電圧Ｖｄ１ｆ，Ｖｑ１ｆに変換する。

The voltage conversion unit 31j calculates the amplitude harmonic signal Vr and the phase harmonic signal φr based on the d and q axis harmonic currents Id1f and Iq1f. Specifically, first, the d and q axis harmonic currents Id1f and Iq1f are converted into d and q axis harmonic voltages Vd1f and Vq1f using the following equation (eq16).

続いて、下式（ｅｑ１７），（ｅｑ１８）を用いて、ｄ，ｑ軸高調波電圧Ｖｄ１ｆ，Ｖｑ１ｆに基づき、振幅高調波信号Ｖｒと、位相高調波信号φｒとを算出する。

Subsequently, the amplitude harmonic signal Vr and the phase harmonic signal φr are calculated based on the d and q axis harmonic voltages Vd1f and Vq1f using the following equations (eq17) and (eq18).

なお、下式（ｅｑ１８）のアークタンジェント演算により、例えば、位相高調波信号φｒを「−π〜＋π」の間で算出することができる。特に本実施形態では、下式（ｅｑ１８）の右辺において、括弧内の分母が０となってかつ分子が正の値となる場合、位相高調波信号φｒを「π／２」として算出する。一方、括弧内の分母が０となってかつ分子が負の値となる場合、位相高調波信号φｒを「−π／２」として算出する。

Note that, for example, the phase harmonic signal φr can be calculated between “−π to + π” by arctangent calculation of the following equation (eq18). In particular, in this embodiment, when the denominator in the parenthesis is 0 and the numerator is a positive value on the right side of the following equation (eq18), the phase harmonic signal φr is calculated as “π / 2”. On the other hand, when the denominator in the parenthesis is 0 and the numerator is a negative value, the phase harmonic signal φr is calculated as “−π / 2”.

  Incidentally, in the present embodiment, the primary current converter 31 i and the voltage converter 31 j correspond to “generating means”. The above equation (eq16) can be derived by subtracting the above equation (eq2) from the voltage equation shown in the following equation (eq19) including the fundamental wave current and the harmonic current.

振幅重畳部３１ｋは、補正部３０ｋから出力された電圧振幅「Ｖｎ＋ΔＶ」に振幅高調波信号Ｖｒを重畳して出力する。位相重畳部３１ｌは、位相設定部３０ｄから出力された電圧位相φに位相高調波信号φｒを重畳して出力する。なお、本実施形態において、振幅重畳部３１ｋ及び位相重畳部３１ｌが「重畳手段」に相当する。

The amplitude superimposing unit 31k superimposes and outputs the amplitude harmonic signal Vr on the voltage amplitude “Vn + ΔV” output from the correcting unit 30k. The phase superimposing unit 31l superimposes and outputs the phase harmonic signal φr on the voltage phase φ output from the phase setting unit 30d. In the present embodiment, the amplitude superimposing unit 31k and the phase superimposing unit 31l correspond to “superimposing means”.

  As described above, in this embodiment, the operation signal waveform is selected and output to the switches SUp to SWn based on the output value Vk of the amplitude superimposing unit 31k and the output value δ of the phase superimposing unit 31l. As a result, the harmonic current flowing through the motor generator 10 can be reduced, and as a result, torque fluctuation and loss can be reduced.

  Furthermore, in the present embodiment, the target high-order currents Idk * and Iqk * are variably set based on the target torque Trq * and the electrical angular velocity ω when generating the harmonic signals Vr and φr. Such a setting contributes to an improvement in the harmonic current reduction effect.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, a plurality (two) of harmonic currents to be reduced are used. Specifically, in order to reduce the ± 6th harmonic current in the primary rotating coordinate system, the −5th harmonic current and the 7th harmonic current are superimposed on each phase current.

  FIG. 9 is a block diagram of the harmonic processing unit 31 according to the present embodiment. In FIG. 9, the same processes as those shown in FIG. 8 are denoted by the same reference numerals for the sake of convenience.

  In the present embodiment, the processing units 31c to 31j shown in FIG. 8 are replaced with the first higher-order current conversion unit 31c, the first target value setting unit 31d, the first d- and q-axis higher-order deviation calculation units 31e and 31f. 1d, q-axis feedback control units 31g, 31h, first primary current conversion unit 31i, and first voltage conversion unit 31j. Then, the output value of the first higher-order current conversion unit 31c is the first d, q-axis higher order currents Idkr1, Iqkr1, and the output value of the first target value setting unit 31d is the first d, q-axis target higher-order current Idk * 1. , Iqk * 1. In addition, the output values of the first d and q-axis higher order deviation calculation units 31e and 31f are the first d and q-axis higher order deviations Δdk1 and Δqk1, and the output values of the first d and q-axis feedback control units 31g and 31h are the first d, The q-axis feedback currents Idkf1 and Iqkf1 are used, and the output value of the first primary current converter 31i is the first d and q-axis harmonic currents Id1f1 and Iq1f1. The output values of the first voltage converter 31j are the first amplitude harmonic signal Vr1 and the first phase harmonic signal φr1. In the present embodiment, the harmonic signals Vr1 and φr1 are signals for superimposing a −5th harmonic current on each phase current. Therefore, the high-order rotational coordinate system (first high-order rotational coordinate system) in the first high-order current conversion unit 31c is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the −5th harmonic current in the fixed coordinate system. It becomes.

  In addition to such a configuration, the configurations 31m to 31t for generating the second amplitude harmonic signal Vr2 and the second phase harmonic signal φr2 for superimposing the seventh harmonic current on each phase current include harmonic processing units. 31. In the present embodiment, these configurations 31m to 31t are the same as the configurations 31c to 31j for generating the first amplitude harmonic signal Vr1 and the first phase harmonic signal φr1. Specifically, the second higher-order current conversion unit 31m converts the current deviations ΔId and ΔIq into second d- and q-axis higher-order currents Idkr2 and Iqkr2 in the second higher-order rotational coordinate system. Here, the second higher-order rotational coordinate system is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the seventh-order harmonic current in the fixed coordinate system.

  The second target value setting unit 31n variably sets the second d and q-axis target higher-order currents Idk * 2 and Iqk * 2 based on the target torque Trq * and the electrical angular velocity ω. Each target high-order current Idk * 2, Iqk * 2 is set from the viewpoint of suppressing torque fluctuation and iron loss.

  The second d-axis higher-order deviation calculating unit 31o calculates a second d-axis higher-order deviation Δdk2 that is a deviation between the second d-axis target higher-order current Idk * 2 and the second d-axis higher-order current Idkr2. The second q-axis higher-order deviation calculation unit 31p calculates a second q-axis higher-order deviation Δqk2 that is a deviation between the second q-axis target higher-order current Iqk * 2 and the second q-axis higher-order current Iqkr2. The second d-axis feedback control unit 31q calculates the second d-axis feedback current Idkf2 by proportional-integral control with the second d-axis higher-order deviation Δdk2 as an input. The second q-axis feedback control unit 31r calculates the second q-axis feedback current Iqkf2 by proportional-integral control with the second q-axis higher-order deviation Δqk2 as an input.

  The second primary current converter 31s uses the above equation (eq15) to convert the second d and q axis feedback currents Idkf2 and Iqkf2 in the second higher order rotation coordinate system into the second d and q axes in the primary rotation coordinate system. It converts into harmonic current Id1f2 and Iq1f2. The second voltage conversion unit 31t uses the same method as the first voltage conversion unit 31j to generate the second amplitude harmonic signal Vr2 and the second phase harmonic signal φr2 based on the second d and q axis harmonic currents Id1f2 and Iq1f2. And calculate.

  The amplitude superimposing unit 31k superimposes and outputs the first and second amplitude harmonic signals Vr1 and Vr2 on the voltage amplitude “Vn + ΔV” output from the correcting unit 30k. The phase superimposing unit 31l superimposes and outputs the first and second phase harmonic signals φr1 and φr2 on the voltage phase φ output from the phase setting unit 30d.

  According to the present embodiment described above, the harmonic current included in each phase current can be further reduced.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  The method for calculating the d and q-axis higher-order currents Idk and Iqk is not limited to the method exemplified in the first embodiment, and may be described below, for example. In the fixed coordinate system, the harmonic current is extracted by subtracting the fundamental current calculated based on the d and q axis command currents Id * and Iq * from the phase current. Then, the extracted harmonic current is directly converted into a current in a high-order rotation coordinate system without going through the primary rotation coordinate system, thereby calculating d and q-axis high-order currents Idk and Iqk.

  The d and q axis target higher-order currents Idk * and Iqk * may be variably set based on any one of the target torque Trq * and the electrical angular velocity ω. Further, the d and q-axis target high-order currents Idk * and Iqk * may be set to fixed values without being variably set.

  In the first embodiment, the d-axis deviation calculation unit 31a extracts the d-axis harmonic component as the deviation between the d-axis command current Id * and the d-axis current Idr. However, the present invention is not limited to this. For example, the d-axis harmonic component may be extracted by applying a filter (bandpass filter or high-pass filter) process to the d-axis current Idr. The same applies to the q-axis harmonic component.

  In the second embodiment, the number of harmonic currents to be reduced may be three or more.

  In each of the above embodiments, as seen in Japanese Patent Application Laid-Open No. 2012-23943, amplitude correction amount ΔV is calculated as an operation amount for feedback control of d-axis current to its command current as feedback control of voltage amplitude May be. In each of the above embodiments, the correction amount calculation unit 32 may be removed. That is, in amplitude control, voltage amplitude feedback control is not essential.

  In each of the above embodiments, a configuration in which the amplitude harmonic signal is not superimposed on the voltage amplitude may be employed. Moreover, in each said embodiment, you may employ | adopt the structure which does not superimpose a phase harmonic signal on a voltage phase. Even in this case, an effect according to the effect obtained in each of the above embodiments can be obtained.

  In the first embodiment, instead of the modulation factor M, a pulse pattern may be stored in the memory in association with the electrical angle for each voltage amplitude “Vn + ΔV”.

  The rotary electric machine is not limited to IPMSM, and may be SPMSM or a wound field type synchronous machine, for example. Further, the control amount of the rotating electrical machine is not limited to torque, and may be, for example, a rotational speed.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 20 ... Inverter, 30 ... Control apparatus, SUp-SWn ... Switch.

Claims (6)

Applied to a three-phase rotating electrical machine (10) electrically connected to a power conversion circuit (20) having switching elements (SUp to SWn);
In a fixed coordinate system, a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the fundamental current flowing in the rotating electrical machine is a primary rotational coordinate system,
Voltage phase setting means (30d) for setting the phase of the output voltage vector of the power conversion circuit in the primary rotating coordinate system as an operation amount for feedback control of the control amount of the rotating electrical machine to the target value;
Voltage amplitude setting means (30h, 30j, 30k, 32) for setting the amplitude of the output voltage vector as an operation amount for controlling the control amount to the target value;
Storage means (33U to 33W) for storing an on / off operation command of the switching element related to the electrical angle of the rotating electrical machine for each amplitude parameter including the amplitude;
The on / off operation command selected based on the amplitude parameter including the amplitude set by the voltage amplitude setting means is shifted by the phase set by the voltage phase setting means with reference to the detected value of the electrical angle. Operation means (33U to 33W) for turning on and off the switching element by outputting to the switching element;
Extraction means (31a, 31b) for extracting a harmonic current flowing in the rotating electrical machine based on a detected phase current value flowing in the rotating electrical machine;
“1 ± 6n” (n is an integer other than 0) is an integer k, the electrical angle of the rotating electrical machine is θe, and each phase fluctuates at an angular velocity k times the electrical angular velocity of the rotating electrical machine in the fixed coordinate system. Are set to Irk × cos (k × θe + φk), Irk × cos (k × θe−2 / 3π + φk), and Irk × cos (k × θe + 2 / 3π + φk). based on the wave current, a harmonic signal for reducing the phase of the harmonic current that varies k times the angular velocity of the previous SL electrical angular speed in the fixed coordinate system, the electricity in the primary rotating coordinate system Generating means (31i, 31j; 31i, 31j, 31s, 31t) for generating a harmonic signal that fluctuates at an angular velocity "k-1" times the angular velocity;
In the fixed coordinate system, at least one of the amplitude and the phase used in the operation unit is generated by the generation unit to reduce the harmonic current of each phase that fluctuates at an angular velocity k times the electrical angular velocity. And a superimposing means (31k, 31l) for superimposing the harmonic signal. When the phase of the output voltage vector in the coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the harmonic current that fluctuates at an angular velocity k times the electrical angular velocity in the fixed coordinate system or the primary rotating coordinate system changes. A coordinate system having a λ axis extending in a direction orthogonal to the direction in which the current vector flowing through the rotating electrical machine in the primary rotating coordinate system changes as a reference axis is referred to as a high-order rotating coordinate system.
Higher current calculation means for calculating a high-order current obtained by converting the harmonic current extracted by said extraction means into a current in the high-order rotational coordinate system (31c; 31c, 31m) and,
Operation amount calculation means (31g, 31h) for calculating an operation amount in the higher-order rotation coordinate system as an operation amount for feedback control of the higher-order current calculated by the higher-order current calculation means to a target higher-order current. 31g, 31h, 31q, 31r) ,
2. The control device for a rotating electrical machine according to claim 1, wherein the generation unit generates the harmonic signal based on the operation amount calculated by the operation amount calculation unit . The higher-order rotating coordinate system is a coordinate system that rotates at the same angular velocity as the fluctuation angular velocity of the harmonic current that fluctuates at an angular velocity that is k times the electrical angular velocity in the fixed coordinate system,
The high-order current calculating means calculates, as the high-order current, a d-axis high-order current and a q-axis high-order current that are orthogonal biaxial components of the high-order rotating coordinate system,
The operation amount calculation means includes:
D-axis operation amount calculation means (31g; 31g, 31q) for calculating a d-axis operation amount in the higher-order rotation coordinate system as an operation amount for feedback control of the d-axis higher-order current to the d-axis target higher-order current. When,
Q-axis operation amount calculation means (31h; 31h, 31r) for calculating a q-axis operation amount in the higher-order rotation coordinate system as an operation amount for feedback-controlling the q-axis higher-order current to the q-axis target higher-order current It is in and,
The generating means, based on the d-axis operation amount and the q-axis operation amount, as the harmonic signal, an amplitude harmonic signal that is a harmonic component of the amplitude in the primary rotation coordinate system, and the primary rotation Generating a phase harmonic signal that is a harmonic component of the phase in the coordinate system;
The control device for a rotating electrical machine according to claim 2, wherein the superimposing unit superimposes the amplitude harmonic signal on the amplitude and superimposes the phase harmonic signal on the phase.   The variable setting means which variably sets the d-axis target high-order current and the q-axis target high-order current based on at least one of the electrical angular velocity of the rotating electrical machine and the target value of the control amount. Control device for rotating electrical machines. Two-phase conversion means (30a) for converting the phase current detection value into a d-axis current and a q-axis current that are orthogonal two-axis components of the primary rotation coordinate system;
The voltage amplitude setting means sets the amplitude based on a command value of the d-axis current that is a DC component and a command value of the q-axis current that is a DC component,
The extraction means extracts a difference between the d-axis current and the command value of the d-axis current as a d-axis harmonic component, and calculates a difference between the q-axis current and the command value of the q-axis current as a q-axis. Extract as harmonic components,
5. The rotating electrical machine according to claim 3, wherein the high-order current calculation unit calculates the d-axis high-order current and the q-axis high-order current based on the d-axis harmonic component and the q-axis harmonic component. Control device.   The harmonic currents to be reduced by the harmonic signal are a plurality of harmonic currents that fluctuate at different angular velocities among the harmonic currents that fluctuate at an angular velocity k times the electrical angular velocity. The control apparatus of the rotary electric machine of any one of these.

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