JP5027755B2 - Display device and driving method thereof - Google Patents

Description

  The present invention relates to a display device that displays an image by current-driving light emitting elements arranged for each pixel, and a driving method thereof. Specifically, the present invention relates to a so-called active matrix display device that controls the amount of current supplied to a light emitting element such as an organic EL by an insulated gate field effect transistor provided in each pixel circuit, and a driving method thereof.

  In an image display device such as a liquid crystal display, an image is displayed by arranging a large number of liquid crystal pixels in a matrix and controlling the transmission intensity or reflection intensity of incident light for each pixel in accordance with image information to be displayed. This also applies to an organic EL display using an organic EL element as a pixel, but unlike a liquid crystal pixel, the organic EL element is a self-luminous element. Therefore, the organic EL display has advantages such as higher image visibility than the liquid crystal display, no backlight, and high response speed. Further, the luminance level (gradation) of each light emitting element can be controlled by the value of the current flowing therethrough, and is greatly different from a voltage control type such as a liquid crystal display in that it is a so-called current control type.

In the organic EL display, similarly to the liquid crystal display, there are a simple matrix method and an active matrix method as driving methods. Although the former has a simple structure, there is a problem that it is difficult to realize a large-sized and high-definition display. Therefore, the active matrix method is actively developed at present. In this method, a current flowing through a light emitting element in each pixel circuit is controlled by an active element (generally a thin film transistor or TFT) provided in the pixel circuit, and is described in the following patent documents.
JP 2003-255856 A JP 2003-271095 A JP 2004-133240 A JP 2004-029791 A JP 2004-093682 A JP-A-2005-173434

  A conventional pixel circuit is arranged at a portion where a row scanning line for supplying a control signal and a column signal line for supplying a video signal intersect, and includes at least a sampling transistor, a pixel capacitor, a drive transistor, and a light emitting element. . The sampling transistor conducts in response to the control signal supplied from the scanning line and samples the video signal supplied from the signal line. The pixel capacitor holds an input voltage corresponding to the signal potential of the sampled video signal. The drive transistor supplies an output current as a drive current during a predetermined light emission period in accordance with the input voltage held in the pixel capacitor. In general, the output current depends on the carrier mobility and threshold voltage of the channel region of the drive transistor. The light emitting element emits light with luminance according to the video signal by the output current supplied from the drive transistor.

  The drive transistor receives an input voltage held in the pixel capacitor at the gate, causes an output current to flow between the source and the drain, and energizes the light emitting element. In general, the light emission luminance of a light emitting element is proportional to the amount of current applied. Further, the output current supply amount of the drive transistor is controlled by the gate voltage, that is, the input voltage written in the pixel capacitor. The conventional pixel circuit controls the amount of current supplied to the light emitting element by changing the input voltage applied to the gate of the drive transistor in accordance with the input video signal.

Here, the operating characteristic of the drive transistor is expressed by the following Equation 1.
Ids = (1/2) μ (W / L) Cox (Vgs−Vth) ² Formula 1
In the transistor characteristic formula 1, Ids represents a drain current flowing between the source and the drain, and is an output current supplied to the light emitting element in the pixel circuit. Vgs represents a gate voltage applied to the gate with reference to the source, and is the above-described input voltage in the pixel circuit. Vth is the threshold voltage of the transistor. Μ represents the mobility of the semiconductor thin film constituting the channel of the transistor. In addition, W represents the channel width, L represents the channel length, and Cox represents the gate capacitance. As is apparent from the transistor characteristic equation 1, when the thin film transistor operates in the saturation region, if the gate voltage Vgs increases beyond the threshold voltage Vth, the thin film transistor is turned on and the drain current Ids flows. In principle, as shown in the above transistor characteristic equation 1, if the gate voltage Vgs is constant, the same amount of drain current Ids is always supplied to the light emitting element. Therefore, if video signals of the same level are supplied to all the pixels constituting the screen, all the pixels should emit light with the same luminance, and the uniformity of the screen should be obtained.

  However, in reality, thin film transistors (TFTs) composed of semiconductor thin films such as polysilicon have variations in individual device characteristics. In particular, the threshold voltage Vth is not constant and varies from pixel to pixel. As apparent from the transistor characteristic equation 1 described above, if the threshold voltage Vth of each drive transistor varies, even if the gate voltage Vgs is constant, the drain current Ids varies and the luminance varies from pixel to pixel. , Damage the screen uniformity. Conventionally, a pixel circuit incorporating a function for canceling variations in threshold voltages of drive transistors has been developed, and is disclosed in, for example, Patent Document 3 described above.

  However, the variation factor of the output current with respect to the light emitting element is not only the threshold voltage Vth of the drive transistor. As is apparent from the transistor characteristic equation 1 described above, the output current Ids varies even when the mobility μ of the drive transistor varies. As a result, the uniformity of the screen is impaired. Correcting the variation in mobility is also a problem to be solved.

In view of the above-described problems of the conventional technology, it is a general object of the present invention to provide a display device having a function of correcting the mobility of a drive transistor for each pixel and a driving method thereof. In particular, it is an object to provide a display device capable of adaptively correcting mobility with respect to the luminance level of a pixel and a driving method thereof. The following measures were taken in order to achieve this purpose. That is, the display device according to the present invention includes a pixel array section and a drive section that drives the pixel array section, and the pixel array section is arranged at a row scanning line, a column signal line, and a portion where these intersect. A matrix-like pixel, and a power supply line and a ground line for supplying power to each pixel, and the driving unit includes a scanner that sequentially supplies pixels to each scanning line by sequentially supplying a control signal to each scanning line, The pixel includes at least a light emitting element, a sampling transistor, a drive transistor, and a pixel capacitor. The sampling transistor has a gate connected to the scanning line and one of the source / drain connected to the signal line. And the other is connected to the gate of the drive transistor, and the drive transistor and the light emitting element are connected in series between the power supply line and the ground line. Formed, the pixel capacitor is connected between the gate and the light emitting element of the drive transistor, having a slope in the waveform at the time of turning off the sampling transistor at the scanner supplies a control signal.
In an embodiment, by giving a slope to the waveform of the control signal when turning off the sampling transistor, the time for turning off the sampling transistor of a pixel having a higher luminance level held in the pixel capacitor is shortened. In one aspect, when the inclination of the waveform of the control signal is given, the scanner first makes the inclination steep in at least two stages and then makes the inclination gentle. Preferably, the driving unit includes a power pulse generation circuit that generates a power pulse that is a source of a waveform of the control signal and supplies the power pulse to the scanner, and the scanner sequentially extracts the waveform from the power pulse and outputs a control signal. Is supplied to each scanning line as a waveform. In an embodiment, the sampling transistor is turned on according to the control signal supplied from the scanning line, then turned off according to the waveform thereof, and supplied from the signal line during the period from turning on to turning off. The video signal is sampled and held in the pixel capacitor as the luminance level, and the drive transistor causes a driving current to flow through the current path to the light emitting element in accordance with the video signal held in the pixel capacitor. Light at level.

  The display device according to the present invention includes a pixel array section and a drive section for driving the pixel array section, and the pixel array section is arranged at a row scanning line and a column signal line at a portion where they intersect. A matrix-like pixel, and a power supply line and a ground line for supplying power to each pixel, and the driving unit includes a scanner that sequentially supplies a control signal to each scanning line to scan the pixels line by line. The pixel includes at least a light emitting element, a sampling transistor, a drive transistor, and a pixel capacitor. The sampling transistor has a gate connected to the scanning line and one of the source / drain connected to the signal line. The other is connected to the gate of the drive transistor, and the drive transistor and the light emitting element are connected in series between the power line and the ground line to form a current path. The pixel capacitor is connected between the gate of the drive transistor and the light emitting element, applies the control signal to the scanning line, turns on the sampling transistor, and holds the video signal in the pixel capacitor. Thereafter, when the sampling transistor is turned off, the sampling transistor is turned off earlier as the luminance level of the video signal supplied from the signal line is higher.

  According to the present invention, the mobility of the drive transistor is corrected using a part of the period during which the signal potential is sampled into the pixel capacitance (sampling period). Specifically, in the latter half of the sampling period, the switching transistor is turned on to make the current path conductive, and a drive current is passed through the drive transistor. This drive current has a magnitude corresponding to the sampled signal potential. At this stage, the light emitting element is in a reverse bias state, and the drive current does not flow through the light emitting element but is charged to its parasitic capacitance and pixel capacitance. Thereafter, the sampling pulse falls, and the gate of the drive transistor is disconnected from the signal line. During the correction period from when the switching transistor is turned on to when the sampling transistor is turned off, the drive current is negatively fed back from the drive transistor to the pixel capacitor, and that amount is subtracted from the signal potential sampled in the pixel capacitor. Since this negative feedback amount acts in a direction to suppress variation in mobility of the drive transistor, mobility correction can be performed for each pixel. That is, when the mobility of the drive transistor is large, the amount of negative feedback with respect to the pixel capacitance is increased, the signal potential held in the pixel capacitance is greatly reduced, and consequently the output current of the drive transistor is suppressed. On the other hand, when the mobility of the drive transistor is small, the negative feedback amount is also small, and the signal potential held in the pixel capacitor is not significantly affected. Therefore, the output current of the drive transistor does not drop so much. Here, the negative feedback amount has a level corresponding to the signal potential applied directly from the signal line to the gate of the drive transistor. That is, the negative feedback amount increases as the signal potential increases and the luminance increases. As described above, the mobility correction is performed according to the luminance level.

  However, the optimum correction period is not necessarily the same between the case where the luminance is high and the case where the luminance is low. In general, when the luminance is high (white level), the optimum correction period is relatively short. Conversely, when the luminance is intermediate (gray level), the optimum correction period tends to be longer. In the present invention, the correction period is automatically optimized according to the luminance level. That is, the present invention automatically adjusts the second timing at which the sampling transistor is turned off according to the signal potential with respect to the first timing at which the switching transistor is turned on. Specifically, adaptive control is performed so that the correction period is shortened when the signal potential of the video signal supplied from the signal line is high, while the correction period is lengthened when the signal potential of the video signal supplied to the signal line is low. is doing. Thereby, it is possible to optimally variably control the correction period according to the signal potential. With such a configuration, the uniformity of the screen can be further improved.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 1 is a schematic block diagram showing the overall configuration of a display device according to the present invention. As shown in the figure, this image display apparatus basically includes a pixel array unit 1 and a drive unit including a scanner unit and a signal unit. The pixel array unit 1 includes a scanning line WS, a scanning line AZ1, a scanning line AZ2, and a scanning line DS arranged in a row, a signal line SL arranged in a column, and the scanning lines WS, AZ1, AZ2, DS. And a matrix pixel circuit 2 connected to the signal line SL, and a plurality of power supply lines for supplying the first potential Vss1, the second potential Vss2, and the third potential Vcc necessary for the operation of each pixel circuit 2. The signal unit includes a horizontal selector 3 and supplies a video signal to the signal line SL. The scanner unit includes a write scanner 4, a drive scanner 5, a first correction scanner 71, and a second correction scanner 72, and supplies control signals to the scanning line WS, the scanning line DS, the scanning line AZ1, and the scanning line AZ2, respectively. The pixel circuit is sequentially scanned for each row.

  Here, the write scanner 4 is composed of a shift register, operates in response to a clock signal WSCK supplied from the outside, and sequentially rolls a start signal WSST supplied from the outside and outputs it to each scanning line WS. is doing. At that time, the falling waveform of the control signal WS is generated using the power supply pulse WSP supplied from the outside. The drive scanner 5 is also composed of a shift register, operates in response to an externally supplied clock signal DSCK, and sequentially outputs a control signal DS to each scanning line DS by sequentially transferring a start signal DSST supplied from the outside. is doing.

  FIG. 2 is a circuit diagram showing a configuration example of a pixel circuit incorporated in the image display device shown in FIG. As illustrated, the pixel circuit 2 includes a sampling transistor Tr1, a drive transistor Trd, a first switching transistor Tr2, a second switching transistor Tr3, a third switching transistor Tr4, a pixel capacitor Cs, and a light emitting element EL. Including. The sampling transistor Tr1 conducts in response to a control signal supplied from the scanning line WS during a predetermined sampling period, and samples the signal potential of the video signal supplied from the signal line SL into the pixel capacitor Cs. The pixel capacitor Cs applies an input voltage Vgs to the gate G of the drive transistor Trd in accordance with the signal potential of the sampled video signal. The drive transistor Trd supplies an output current Ids corresponding to the input voltage Vgs to the light emitting element EL. The light emitting element EL emits light with a luminance corresponding to the signal potential of the video signal by the output current Ids supplied from the drive transistor Trd during a predetermined light emission period.

  The first switching transistor Tr2 is turned on according to the control signal supplied from the scanning line AZ1 prior to the sampling period, and sets the gate G of the drive transistor Trd to the first potential Vss1. The second switching transistor Tr3 is turned on in response to a control signal supplied from the scanning line AZ2 prior to the sampling period, and sets the source S of the drive transistor Trd to the second potential Vss2. The third switching transistor Tr4 is turned on in response to a control signal supplied from the scanning line DS prior to the sampling period to connect the drive transistor Trd to the third potential Vcc, and thus corresponds to the threshold voltage Vth of the drive transistor Trd. The voltage is held in the pixel capacitor Cs to correct the influence of the threshold voltage Vth. Further, the third switching transistor Tr4 is turned on again in response to the control signal supplied from the scanning line DS during the light emission period, connects the drive transistor Trd to the third potential Vcc, and causes the output current Ids to flow through the light emitting element EL.

  As is apparent from the above description, the pixel circuit 2 is composed of five transistors Tr1 to Tr4 and Trd, one pixel capacitor Cs, and one light emitting element EL. The transistors Tr1 to Tr3 and Trd are N channel type polysilicon TFTs. Only the transistor Tr4 is a P-channel type polysilicon TFT. However, the present invention is not limited to this, and N-channel and P-channel TFTs can be mixed as appropriate. The light emitting element EL is, for example, a diode type organic EL device having an anode and a cathode. However, the present invention is not limited to this, and the light emitting element generally includes all devices that emit light by current drive.

  FIG. 3 is a schematic diagram in which only the pixel circuit 2 is extracted from the image display device shown in FIG. In order to facilitate understanding, the signal potential Vsig of the video signal sampled by the sampling transistor Tr1, the input voltage Vgs and output current Ids of the drive transistor Trd, and the capacitance component Coled of the light emitting element EL are added. . The operation of the pixel circuit 2 according to the present invention will be described below with reference to FIG.

  FIG. 4 is a timing chart of the pixel circuit shown in FIG. With reference to FIG. 4, the operation of the pixel circuit according to the present invention shown in FIG. 3 will be described in detail. FIG. 4 shows the waveforms of control signals applied to the scanning lines WS, AZ1, AZ2 and DS along the time axis T. In order to simplify the notation, the control signals are also represented by the same reference numerals as the corresponding scanning lines. Since the transistors Tr1, Tr2 and Tr3 are N-channel type, they are turned on when the scanning lines WS, AZ1 and AZ2 are at a high level and turned off when the scanning lines are at a low level. On the other hand, since the transistor Tr4 is a P-channel type, it is turned off when the scanning line DS is at a high level and turned on when it is at a low level. This timing chart also shows the change in the potential of the gate G and the change in the potential of the source S of the drive transistor Trd, along with the waveforms of the control signals WS, AZ1, AZ2, and DS.

  In the timing chart of FIG. 4, timings T1 to T8 are defined as one field (1f). Each row of the pixel array is sequentially scanned once during one field. The timing chart shows the waveforms of the control signals WS, AZ1, AZ2, DS applied to the pixels for one row.

  At timing T0 before the field starts, all control line numbers WS, AZ1, AZ2, DS are at a low level. Therefore, the N-channel transistors Tr1, Tr2, Tr3 are in the off state, while only the P-channel transistor Tr4 is in the on state. Therefore, since the drive transistor Trd is connected to the power supply Vcc via the transistor Tr4 in the on state, the output current Ids is supplied to the light emitting element EL according to the predetermined input voltage Vgs. Therefore, the light emitting element EL emits light at the timing T0. At this time, the input voltage Vgs applied to the drive transistor Trd is expressed by the difference between the gate potential (G) and the source potential (S).

  At the timing T1 when the field starts, the control signal DS is switched from the low level to the high level. As a result, the transistor Tr4 is turned off and the drive transistor Trd is disconnected from the power supply Vcc, so that the light emission stops and the non-light emission period starts. Therefore, at the timing T1, all the transistors Tr1 to Tr4 are turned off.

  After timing T1, the control signal AZ2 rises at timing T21, and the switching transistor Tr3 is turned on. As a result, the source (S) of the drive transistor Trd is initialized to the predetermined potential Vss2. Subsequently, at timing T22, the control signal AZ1 rises and the switching transistor Tr2 is turned on. As a result, the gate potential (G) of the drive transistor Trd is initialized to a predetermined potential Vss1. As a result, the gate G of the drive transistor Trd is connected to the reference potential Vss1, and the source S is connected to the reference potential Vss2. Here, Vss1−Vss2> Vth is satisfied, and by setting Vss1−Vss2 = Vgs> Vth, preparation for Vth correction performed at timing T3 is performed. In other words, the period T21-T3 corresponds to a reset period of the drive transistor Trd. Further, when the threshold voltage of the light emitting element EL is VthEL, VthEL> Vss2 is set. Thereby, a minus bias is applied to the light emitting element EL, and a so-called reverse bias state is obtained. This reverse bias state is necessary for normally performing the Vth correction operation and the mobility correction operation to be performed later.

  At timing T3, the control signal AZ2 is set to low level, and then the control signal DS is set to low level. As a result, the transistor Tr3 is turned off while the transistor Tr4 is turned on. As a result, the drain current Ids flows into the pixel capacitor Cs, and the Vth correction operation is started. At this time, the gate G of the drive transistor Trd is held at Vss1, and the current Ids flows until the drive transistor Trd is cut off. When cut off, the source potential (S) of the drive transistor Trd becomes Vss1-Vth. At timing T4 after the drain current is cut off, the control signal DS is returned to the high level again, and the switching transistor Tr4 is turned off. Further, the control signal AZ1 is also returned to the low level, and the switching transistor Tr2 is also turned off. As a result, Vth is held and fixed in the pixel capacitor Cs. Thus, the timing T3-T4 is a period for detecting the threshold voltage Vth of the drive transistor Trd. Here, this detection period T3-T4 is called a Vth correction period.

  After performing the Vth correction in this way, the control signal WS is switched to the high level at the timing T5, the sampling transistor Tr1 is turned on, and the signal potential Vsig of the video signal is written in the pixel capacitor Cs. The pixel capacitance Cs is sufficiently smaller than the equivalent capacitance Coled of the light emitting element EL. As a result, almost most of the signal potential Vsig of the video signal is written into the pixel capacitor Cs. To be precise, for Vss1. The difference Vsig−Vss1 of Vsig is written to the pixel capacitor Cs. Therefore, the voltage Vgs between the gate G and the source S of the drive transistor Trd becomes a level (Vsig−Vss1 + Vth) obtained by adding Vth previously detected and held and Vsig−Vss1 sampled this time. In the following description, assuming Vss1 = 0V for simplification of explanation, the gate / source voltage Vgs becomes Vsig + Vth as shown in the timing chart of FIG. The sampling of the signal potential Vsig of the video signal is performed until timing T7 when the control signal WS returns to the low level. That is, the timing T5-T7 corresponds to the sampling period.

  At timing T6 before the end of the sampling period T7, the control signal DS becomes low level and the switching transistor Tr4 is turned on. As a result, the drive transistor Trd is connected to the power supply Vcc, so that the pixel circuit proceeds from the non-light emitting period to the light emitting period. In this manner, the mobility correction of the drive transistor Trd is performed in the period T6-T7 in which the sampling transistor Tr1 is still on and the switching transistor Tr4 is on. That is, in the present invention, the mobility correction is performed in the period T6-T7 in which the rear part of the sampling period and the head part of the light emission period overlap. Note that, at the beginning of the light emission period in which the mobility correction is performed, the light emitting element EL is actually in a reverse bias state, and thus does not emit light. In the mobility correction period T6-T7, the drain current Ids flows through the drive transistor Trd while the gate G of the drive transistor Trd is fixed to the level of the signal potential Vsig of the video signal. Here, by setting Vss1−Vth <VthEL, the light emitting element EL is placed in a reverse bias state, so that it exhibits simple capacitance characteristics instead of diode characteristics. Therefore, the current Ids flowing through the drive transistor Trd is written into a capacitor C = Cs + Coled obtained by combining both the pixel capacitor Cs and the equivalent capacitor Coled of the light emitting element EL. As a result, the source potential (S) of the drive transistor Trd increases. In the timing chart of FIG. 4, this increase is represented by ΔV. Since this increase ΔV is eventually subtracted from the gate / source voltage Vgs held in the pixel capacitor Cs, negative feedback is applied. In this way, the mobility μ can be corrected by negatively feeding back the output current Ids of the drive transistor Trd to the input voltage Vgs of the drive transistor Trd. The negative feedback amount ΔV can be optimized by adjusting the time width t of the mobility correction period T6-T7. For this purpose, the fall of the control signal WS is inclined.

At timing T7, the control signal WS becomes low level and the sampling transistor Tr1 is turned off. As a result, the gate G of the drive transistor Trd is disconnected from the signal line SL. Since the application of the signal potential Vsig of the video signal is released, the gate potential (G) of the drive transistor Trd can be increased and increases with the source potential (S). Meanwhile, the gate / source voltage Vgs held in the pixel capacitor Cs maintains a value of (Vsig−ΔV + Vth). As the source potential (S) rises, the reverse bias state of the light emitting element EL is canceled, so that the light emitting element EL actually starts to emit light by the inflow of the output current Ids. The relationship between the drain current Ids and the gate voltage Vgs at this time is given by the following equation 2 by substituting Vsig−ΔV + Vth into Vgs of the previous transistor characteristic equation 1.
Ids = kμ (Vgs−Vth) ² = kμ (Vsig−ΔV) ² Equation 2
In the above formula 2, k = (1/2) (W / L) Cox. It can be seen from the characteristic formula 2 that the term Vth is canceled and the output current Ids supplied to the light emitting element EL does not depend on the threshold voltage Vth of the drive transistor Trd. Basically, the drain current Ids is determined by the signal potential Vsig of the video signal. In other words, the light emitting element EL emits light with a luminance corresponding to the signal potential Vsig of the video signal. At that time, Vsig is corrected by the feedback amount ΔV. This correction amount ΔV acts so as to cancel the effect of the mobility μ located in the coefficient part of the characteristic formula 2 just. Therefore, the drain current Ids substantially depends only on the signal potential Vsig of the video signal.

  Finally, when the timing T8 is reached, the control signal DS becomes high level, the switching transistor Tr4 is turned off, the light emission ends, and the field ends. Thereafter, the operation proceeds to the next field, and the Vth correction operation, the signal potential sampling operation, the mobility correction operation, and the light emission operation are repeated again.

  FIG. 5 is a circuit diagram showing a state of the pixel circuit 2 in the mobility correction period T6-T7. As shown in the figure, in the mobility correction period T6-T7, the sampling transistor Tr1 and the switching transistor Tr4 are on, while the remaining switching transistors Tr2 and Tr3 are off. In this state, the source potential (S) of the drive transistor Tr4 is Vss1-Vth. This source potential (S) is also the anode potential of the light emitting element EL. By setting Vss1−Vth <VthEL as described above, the light emitting element EL is placed in a reverse bias state, and exhibits simple capacitance characteristics instead of diode characteristics. Therefore, the current Ids flowing through the drive transistor Trd flows into the combined capacitance C = Cs + Coled of the pixel capacitance Cs and the equivalent capacitance Coled of the light emitting element EL. In other words, a part of the drain current Ids is negatively fed back to the pixel capacitor Cs, and the mobility is corrected.

  FIG. 6 is a graph of the above-described transistor characteristic formula 2, in which Ids is plotted on the vertical axis and Vsig is plotted on the horizontal axis. The characteristic formula 2 is also shown below the graph. In the graph of FIG. 6, a characteristic curve is drawn in a state where the pixel 1 and the pixel 2 are compared. The mobility μ of the drive transistor of the pixel 1 is relatively large. Conversely, the mobility μ of the drive transistor included in the pixel 2 is relatively small. Thus, when the drive transistor is composed of a polysilicon thin film transistor or the like, it is inevitable that the mobility μ varies between pixels. For example, when the signal potential Vsig of the video signal of the same level is written in both the pixels 1 and 2, the output current Ids 1 ′ flowing through the pixel 1 having the high mobility μ is equal to the mobility μ unless the mobility is corrected. A large difference is generated as compared with the output current Ids2 'flowing through the small pixel 2. In this way, a large difference occurs between the output currents Ids due to the variation in the mobility μ, so that unevenness occurs and the uniformity of the screen is impaired.

  Therefore, in the present invention, the variation in mobility is canceled by negatively feeding back the output current to the input voltage side. As apparent from the previous transistor characteristic equation 1, the drain current Ids increases when the mobility is large. Therefore, the negative feedback amount ΔV increases as the mobility increases. As shown in the graph of FIG. 6, the negative feedback amount ΔV1 of the pixel 1 having a high mobility μ is larger than the negative feedback amount ΔV2 of the pixel 2 having a low mobility. Therefore, the larger the mobility μ is, the more negative feedback is applied, and the variation can be suppressed. As shown in the figure, when ΔV1 is corrected in the pixel 1 having a high mobility μ, the output current greatly decreases from Ids1 ′ to Ids1. On the other hand, since the correction amount ΔV2 of the pixel 2 having the low mobility μ is small, the output current Ids2 ′ does not decrease so much to Ids2. As a result, Ids1 and Ids2 are substantially equal, and the variation in mobility is cancelled. Since the cancellation of the variation in mobility is performed in the entire range of Vsig from the black level to the white level, the uniformity of the screen becomes extremely high. In summary, when there are pixels 1 and 2 having different mobility, the correction amount ΔV1 of the pixel 1 having high mobility is smaller than the correction amount ΔV2 of the pixel 2 having low mobility. That is, as the mobility increases, ΔV increases and the decrease value of Ids increases. As a result, pixel current values having different mobilities are made uniform, and variations in mobility can be corrected.

For reference, numerical analysis of the mobility correction described above is performed. As shown in FIG. 5, the analysis is performed by taking the source potential of the drive transistor Trd as a variable V in a state where the transistors Tr1 and Tr4 are turned on. Assuming that the source potential (S) of the drive transistor Trd is V, the drain current Ids flowing through the drive transistor Trd is as shown in Equation 3 below.

Further, Ids = dQ / dt = CdV / dt is established as shown in the following Expression 4 by the relationship between the drain current Ids and the capacitance C (= Cs + Coled).

Both sides are integrated by substituting Equation 3 into Equation 4. Here, the initial state of the source voltage V is -Vth, and the mobility variation correction time (T6-T7) is t. When this differential equation is solved, the pixel current with respect to the mobility correction time t is given as shown in Equation 5 below.

  By the way, the optimum mobility correction time t tends to vary depending on the luminance level of the pixel (the signal potential Vsig of the video signal). This point will be described with reference to FIG. In the graph of FIG. 7, the horizontal axis represents mobility correction time t (T7-T6), and the vertical axis represents luminance (signal potential). In the case of high luminance (white gradation), when the mobility correction time is set to t1 between the drive transistor with high mobility and the drive transistor with low mobility, the luminance levels are exactly equal. That is, when the input signal potential is white gradation, the mobility correction time t1 is the optimum correction time. On the other hand, when the signal potential is intermediate luminance (gray gradation), there is a difference in luminance between the high mobility transistor and the low mobility transistor at the mobility correction time t1, and complete correction cannot be performed. If a correction time t2 longer than t1 is ensured, the luminance is the same level between transistors with high mobility and low mobility. Therefore, when the signal potential is a gray gradation, the optimum correction time t2 is longer than the optimum correction time t1 when the signal potential is white.

  If the mobility correction time t is fixed regardless of the luminance level, the mobility correction cannot be performed completely at all gradations, resulting in unevenness. For example, if the mobility correction time t is matched with the white gradation optimum correction period t1, streaks remain on the screen when the input video signal is in gray gradation. Conversely, when the gray gradation optimum correction period t2 is fixed, stripes appear on the screen when the video signal has a white gradation. That is, if the mobility correction time t is fixed, it is not possible to simultaneously correct the mobility variation over all gradations from white to gray gradation.

  Therefore, the present invention makes it possible to optimally automatically adjust the mobility correction period according to the level of the input video signal. This point will be described in detail with reference to FIG. FIG. 8 shows the falling waveform of the control signal DS applied to the gate of the switching transistor Tr4. In the present embodiment, since the switching transistor Tr4 is a P-channel type, the transistor Tr4 is turned on when the control signal DS falls (T6). This timing T6 is the start time of the mobility correction period as described above. A falling waveform of the control signal WS is also shown together with the control signal DS. This control signal WS is applied to the gate of the sampling transistor Tr1. As described above, since the sampling transistor Tr1 is an N-channel type in this embodiment, the sampling transistor Tr1 is turned off at the time T7 when the control signal WS falls, and the mobility correction period ends.

  As a feature of the present invention, when the waveform of the control signal WS is turned off, the waveform is first sharply dropped to an appropriate potential, and then the pulse is dropped from that to the final potential. Accordingly, two or more mobility correction periods can be provided with a gradation determined by a desired potential as a boundary. For convenience of explanation, the first voltage dropped sharply will be called the 1st voltage, and the final potential dropped after smoothing will be called the 2nd voltage. Here, as a model, the operation is considered with the waveform of the control signal WS set to 1st voltage = 8V, 2nd voltage = 4V. The threshold voltage of the sampling transistor Tr1 is set to Vth (Tr1) = 2V.

  When the white gradation Vsig1 = 8V is written, the sampling transistor Tr1 cuts off at time T7 when the control signal WS drops to Vsig1 + Vth (Tr1) = 10V. That is, when Vsig = 8V is applied from the signal line to the source of the sampling transistor Tr1, the sampling transistor Tr1 is cut off when the gate potential of the sampling transistor Tr1 is higher than the source potential by the threshold voltage 2V. In this way, in the case of white gradation, the mobility correction period t1 = T7−T6 is determined from the control signal DS ON timing T6 to the point T7 until the control signal WS falls steeply to the 1st voltage.

  On the other hand, when the gray gradation Vsig2 = 4V is written, the cutoff voltage of the sampling transistor Tr1 is Vsig2 + Vth (Tr1) = 6V. The timing when the control signal WS falls to the cutoff voltage of 6V is timing T7 '. In the case of gray gradation, the correction time t2 is determined at a point T7 ′ that is smoothed from the 1st voltage of the WS waveform off to the 2nd voltage from the on timing T6 of the control signal DS. That is, the correction period t2 for the gray gradation is longer than the correction time t1 for the white gradation.

  When the gradation is lower, for example, Vsig = 3V, similarly, the cutoff voltage of the sampling transistor Tr1 is 5V, and since the waveform is rounded, the cutoff timing T7 ′ is further shifted backward, and the mobility correction time becomes longer. . In this way, the driving method can make the mobility correction time t longer as the gradation becomes lower.

  In this way, the time T7 from when the control signal DS is turned on to when the control signal WS is turned off first steeply drops to the 1st voltage is set in accordance with the optimum white gradation correction time t1, and thus the white gradation correction time is set. Optimized. The 1st voltage may be set in consideration of the threshold voltage Vth (Tr1) so that the sampling transistor Tr1 is cut off at a sharp point in white gradation. Further, the low gradation can be dealt with by finding the optimum correction time t2 for each gradation, setting the 2nd voltage accordingly, and determining how the falling waveform of the control signal WS falls. In this way, it is possible to automatically adjust the optimum correction time t suitable for each level from high gradation to low gradation, thereby eliminating the unevenness in all gradations by canceling the variation in mobility. Become.

  Hereinafter, an embodiment of the method for generating the falling waveform of the control signal WS shown in FIG. 8 will be described in detail. FIG. 9 is a block diagram showing the overall configuration of the present embodiment. The display device according to the present embodiment is composed of a panel 0 made of a glass plate or the like. A pixel array section 1 is integrated and formed at the center of the panel 0. Around the panel 0, there are formed a light scanner 4, a drive scanner 5, a correction scanner 7 and the like which are a part of the drive unit. Although the horizontal selector is not shown, it can be mounted on the panel 0 like the scanners. Alternatively, an external horizontal selector may be used separately from the panel 0.

  FIG. 10 is a schematic circuit diagram showing one stage of the write scanner 4 shown in FIG. This one stage corresponds to one line of scanning lines formed in the pixel array unit 1. However, the example of FIG. 10 is a reference example, not an embodiment, and is a case where a rectangular control pulse WS is output as in the prior art. As shown in the figure, one stage of the write scanner 4 includes a shift register S / R, two intermediate buffers, a level shifter L / V, and one output buffer connected in series. The power supply voltage WSVdd (18V) of the write scanner 4 is supplied to the final output buffer. In this write scanner, the input waveform IN transferred from the previous stage is delayed by one stage in the shift register, and then supplied to the level shifter L / V through the intermediate buffer, and a voltage suitable for driving the final output buffer. Convert to level. This output buffer generates an output waveform OUT obtained by inverting the input waveform IN and supplies the output waveform OUT to the corresponding scanning line WS. This output waveform is a rectangular wave, the high level is WSVdd, and the reference level is WSVss. Since the output waveform OUT has a vertical fall, the mobility correction period is fixed.

  FIG. 11 shows one stage of the write scanner of this embodiment. In order to facilitate understanding, portions corresponding to those of the light scanner of the reference example shown in FIG. 10 are denoted by corresponding reference numerals. The difference is that the power supply voltage WSVdd supplied to the final output buffer in this embodiment has a pulse waveform that changes from 18 V to 5 V, for example. This power pulse WSP is supplied to the write scanner 4 of the panel 0 from an external discrete circuit. At that time, the phase of the power supply pulse WSP is adjusted in advance so as to be synchronized with the operation of the write scanner 4.

  As shown in the figure, when a rectangular pulse IN is input to the corresponding stage from the previous stage, it is applied to the gate of the output buffer through the shift register S / R, the two intermediate buffers, and the level shift L / V. As a result, the output buffer is opened, and the output waveform OUT is supplied to the corresponding scanning line. At this time, since the power supply pulse WSP is applied to the power supply voltage line WSVdd after the output buffer is turned on, the output waveform falls from 18V to 5V with a predetermined curve. Thereafter, the output buffer is closed, and the output waveform becomes the WSVss level.

  With respect to another control signal DS that regulates the mobility correction period in combination with the control signal WS, the waveform can be created with either of the configurations shown in FIGS.

  12 is a schematic circuit diagram showing a configuration example of the final output buffer of the write scanner shown in FIG. As shown in the figure, this output buffer section is composed of a pair of P-channel transistor TrP and N-channel transistor TrN, and is connected in series between a power supply line WSVdd and a ground line WSVss. An input waveform IN is applied to the gates of the transistors TrP and TrN. A power supply pulse WSP whose phase is adjusted in advance with respect to this input waveform is applied to WSVdd. After the transistor TrP is turned on by the application of the input waveform IN, the falling waveform of the power pulse WSP is taken in by the transistor TrP and supplied as the output waveform OUT to the scanning line WS on the pixel 2 side. In some cases, the rising waveform of the power supply pulse WSP may pass through the transistor TrP because of the operation timing. At this time, a mask signal may be applied to the output stage of the final buffer to cut the trailing edge of the power pulse WSP.

  FIG. 13 is a schematic block diagram illustrating the overall configuration of the display device according to the present embodiment. The panel 0 has the configuration shown in FIG. 9 and incorporates various scanners as a part of the drive unit in addition to the pixel array unit. On the other hand, an external drive board 8 and a discrete circuit 9 which are the remaining part of the drive unit are connected to the panel 0. The drive substrate 8 is made of PLD and supplies clock signals WSCK, DSCK, start pulses WSST, DSST and the like necessary for the operation of the scanner mounted on the panel 0. The discrete circuit 9 is inserted between the driving substrate 8 and the panel 0 and generates necessary power supply pulses. Specifically, the input waveform IN is supplied from the drive substrate 8 side, and the waveform is processed to generate an output waveform OUT, which is supplied to the panel 0 side. The discrete circuit 9 is composed of discrete elements such as transistors, resistors, and capacitors, and supplies at least the power pulse WSP to the power line of the write scanner. In some cases, another power pulse DSP may be supplied to the power line of the drive scanner 5. In this manner, the power supply pulses WSP and DSP are generated by the discrete circuit 9 and are respectively input to the power lines of the light scanner and the drive scanner on the panel 0 side. By generating a power pulse waveform with an external discrete circuit 9 separated from the panel 0, it becomes possible to create an optimal waveform and timing for each individual panel 0, and to improve the yield in the non-uniformity inspection of the panel 0 Contribute.

  FIG. 14 is a circuit diagram showing a simplest configuration example of the discrete circuit 9. As shown in the figure, the discrete circuit 9 comprises one transistor, one capacitor, three fixed resistors, and two variable resistors, and processes the input waveform IN supplied from the drive substrate 8 in an analog manner. Thus, the output waveform OUT is supplied to the panel 0 side. In this embodiment, a rectangular input waveform is processed to generate an output waveform whose falling changes in two stages in a polygonal line shape. As shown in the figure, the falling edge of the output waveform sharply slopes in the first stage and switches to a gentle slope in the second stage.

  FIG. 15 is a circuit diagram showing a more complicated configuration example of the discrete circuit 9. The discrete circuit 9 generates a power pulse WSP having a falling waveform that changes in a curved manner instead of the power pulse WSP having a linear falling waveform shown in FIG. 14 and supplies the power pulse WSP to the panel 0 side. The shape of the curve of the falling waveform can be freely set by the timing adjustment volume.

  FIG. 16 shows the waveform of the power supply pulse WSP generated by the discrete circuit 9 shown in FIG. In correspondence with this, the waveform of another power supply pulse DSP is also shown. Note that the falling waveform of the power supply pulse DSP is vertical and is not particularly inclined. Even in this case, the falling timing of the power pulse DSP (that is, the ON timing T6 of the driving switching transistor Tr4) can be freely adjusted on the discrete circuit side.

  As shown in the figure, the power supply pulse WSP suddenly falls from 17.3 V to the 1st voltage, and then gradually falls to the 2nd voltage. The 1st voltage can be adjusted for each panel between 9 and 11V. Typically, the 1st voltage is set to 10V. The 2nd voltage can also be adjusted in the range of 2 to 6 V for each panel. Typically, the 2nd voltage is set to 5V. In addition, the falling waveform from the 1st voltage to the 2nd voltage can be created by an RC curve or the like.

  By the way, if the power supply pulses WSP and DSP are generated by the discrete circuit, the waveforms of the control signals WS and DS can be adjusted outside the panel, and the operation can be performed at an optimum timing for each panel. Contributes to improved panel yield. However, in order to generate a power supply pulse by an external discrete circuit, a high-output driver and power supply are required, which causes disadvantages such as an increase in power consumption and an increase in component costs.

  Therefore, it is conceivable that the control signal DS is generated by logical processing inside the panel. This embodiment will be described below. In this embodiment, in order to eliminate the disadvantages of high power consumption and increased cost due to generation of the power pulse DSP by the discrete circuit, the control signal DS is created by the logic circuit in the panel, and the mobility correction period is set. Yes. At this time, the mobility correction period can be adjusted by setting an enable signal for the control signal DS. Thus, by generating the control pulse DS by setting the enable signal in the logic circuit in the panel, it is possible to reduce power consumption and cost.

  FIG. 17 is a circuit diagram showing one output stage of the drive scanner 5 having the logical processing function described above. As shown in the figure, the output stage of the drive scanner 5 performs logical processing on the control signals WS, DS1 and DS2 and the enable signals DSEN1 and DSEN2 to produce an output waveform. This output waveform is output as the control signal DS to the scanning line DS of the corresponding row. Here, the control signal WS indicates a WS pulse (WS · S / R · in) input to the shift register S / R of the corresponding stage of the write scanner 4. The control signal DS1 represents a DS pulse (DS · S / R · in) that is input to the shift register S / R of the corresponding stage of the drive scanner 5. The control signal DS2 represents a DS pulse (DS · S / R · out) output from the shift register S / R at the corresponding stage of the drive scanner 5.

  FIG. 18 is a waveform diagram showing each control signal and enable signal supplied to the logic circuit shown in FIG. 17 and related clock signals. In this waveform diagram, up to five waveforms WSCK, WS · S / R · in, WS · S / R · out, WSEN, and WSn mainly represent waveforms of control signals related to the light scanner 4 side. Yes. As apparent from the waveform diagram, the write scanner 4 basically operates in accordance with the clock signal WSCK, and sequentially transfers the start pulse by the shift register S / R to generate the control signal WSn for each case. In the present invention, as described above, one control signal WSn is not directly applied to the corresponding scanning line WSn, and the falling portion of the power pulse WSP is extracted by this signal WSn and supplied to the corresponding scanning line. Like to do.

  Signals DSCK, DS · S / R · in, DS · S / R · out, DSEN1_ODD, DEN1_EVEN, DSEN2, and DSn (OUT) shown in the lower part of FIG. 18 are signal waveforms mainly related to the drive scanner 5.

  The logic circuit shown in FIG. 17 obtains an output waveform OUT by performing the logic processing represented by the logic expression shown in the upper part of FIG. This output waveform OUT is shown at the bottom of the timing chart of FIG. As shown in the figure, the control signal DSn includes a part for defining a correction period for Vth cancellation and a mobility μ correction period. The Vth cancellation period is adjusted by the enable signal DSEN1, while the mobility μ correction period can be adjusted by the enable signal DSEN2.

  As described above, the display device according to the present invention basically includes the pixel array unit 1 and the drive unit that drives the pixel array unit 1. The pixel array unit 1 includes row-like first scanning lines WS and second scanning lines DS, column-like signal lines SL, matrix-like pixels 2 arranged at intersections thereof, and power supply to the respective pixels 2. Power supply line Vcc and ground line Vss. The drive unit sequentially supplies the first control signal WS to the first scanning line WS to scan the pixels 2 line-sequentially in units of rows, and to each second scanning line DS in accordance with the line-sequential scanning. A second scanner 5 that sequentially supplies the second control signal DS and a signal selector 3 that supplies video signals to the column-shaped signal lines SL in accordance with the line sequential scanning are provided.

  Each pixel 2 includes a light emitting element EL, a sampling transistor Tr1, a drive transistor Trd, a switching transistor Tr4, and a pixel capacitor Cs. The sampling transistor Tr1 has a gate connected to the first scanning line WS, a source connected to the signal line SL, and a drain connected to the gate G of the drive transistor Trd. The drive transistor Trd and the light emitting element EL are connected in series between the power supply line Vcc and the ground line to form a current path. When the switching transistor Tr4 is inserted into this current path, its gate is connected to the second scanning line DS. The pixel capacitor Cs is connected between the source S and the gate G of the drive transistor Trd.

  In such a configuration, the sampling transistor Tr1 is turned on in response to the first control signal WS supplied from the first scanning line WS, samples the signal potential Vsig of the video signal supplied from the signal line SL, and holds it in the pixel capacitor Cs. . The switching transistor Tr4 is turned on in response to the second control signal DS supplied from the second scanning line DS to bring the aforementioned current path into a conductive state. The drive transistor Trd causes the drive current Ids to flow to the light emitting element EL through the current path set in a conductive state in accordance with the signal potential Vsig held in the pixel capacitor Cs.

  As a feature of the present invention, the driving unit applies the first control signal WS to the first scanning line WS, turns on the sampling transistor Tr1, and starts sampling of the signal potential Vsig. Correction from the first timing T6 when the switching transistor Tr4 is turned on when applied to the scanning line DS to the second timing T7 when the first control signal WS applied to the first scanning line WS is canceled and the sampling transistor Tr1 is turned off In the period t, the correction for the mobility μ of the drive transistor Trd is added to the signal potential Vsig held in the pixel capacitor Cs, and the mobility correction is performed. At this time, the drive unit shortens the correction period t when the signal potential Vsig of the video signal supplied to the signal line SL is high, while the correction period t decreases when the signal potential Vsig of the video signal supplied to the signal line SL is low. The second timing T7 is automatically adjusted to be longer.

  Specifically, when the first scanner 4 in the driving unit turns off the sampling transistor Tr1 at the second timing T7, the first scanner 4 inclines the falling waveform of the first control signal WS so that the correction period when the signal potential Vsig is high. While t becomes shorter, the second timing T7 is automatically adjusted so that the correction period t becomes longer when the signal potential Vsig of the video signal supplied to the signal line SL is low. Preferably, when the first scanner 4 inclines the falling waveform of the first control signal WS, the signal potential Vsig is high by dividing the inclination first in at least two stages and then gradually reducing the inclination. The correction period t is optimized both at the time and when the signal potential Vsig is low.

  Each pixel 2 has a drive transistor threshold voltage Vth correction function in addition to the mobility correction function described above. That is, the pixel includes additional switching transistors Tr2 and Tr3 that reset or initialize the gate potential (G) and the source potential (S) of the drive transistor Trd prior to sampling of the video signal. Prior to the sampling of the video signal, the second scanner 5 temporarily turns on the switching transistor Tr4 via the second control line DS, and passes the drive current Ids through the reset drive transistor Trd to the threshold voltage Vth. A corresponding voltage is held in the pixel capacitor Cs.

  In addition to various scanners built in the panel, the drive unit includes an external power pulse generation circuit (discrete circuit). The power supply pulse generation circuit 9 generates a first power supply pulse WSP that is a source of the falling waveform of the first control signal WS and supplies the first power supply pulse WSP to the first scanner 4 in the panel. The first scanner 4 sequentially extracts the falling waveform from the first power supply pulse WSP and supplies it to each first scanning line WS as the falling waveform of the first control signal WS.

  In one aspect, the power supply pulse generation circuit 9 also generates a second power supply pulse DSP that is the basis of the waveform of the second control signal DS and supplies the second power supply pulse DSP to the second scanner 5. The second scanner 5 sequentially extracts a part of the waveform from the second power supply pulse DSP and supplies it to each second scanning line DS as the waveform of the second control signal DS at the first timing T6.

  In another aspect, the first scanner 4 generates the waveform of the first control signal WS at the second timing T7 that regulates the end of the correction period t based on the first power pulse WSP supplied from the power pulse generation circuit 9. The second scanner 5 generates a waveform of the second control signal DS at the first timing T6 that regulates the start of the correction period t by internal logic processing.

It is a typical block diagram which shows the principal part of the display apparatus which concerns on this invention. It is a circuit diagram which shows the pixel circuit structure of the display apparatus which concerns on this invention. It is a schematic diagram with which operation | movement description of the display apparatus which concerns on this invention is provided. 3 is a timing chart for explaining the operation of the display device according to the present invention. It is a typical circuit diagram with which it uses for operation | movement description of the display apparatus which concerns on this invention. It is a graph with which it uses for operation | movement description of the display apparatus which concerns on this invention. It is a graph with which it uses for operation | movement description of the display apparatus which concerns on this invention. It is a wave form diagram with which it uses for operation | movement description of the display apparatus which concerns on this invention. It is a schematic diagram which shows the whole structure of the Example of the display apparatus which concerns on this invention. It is a schematic diagram which shows the light scanner which concerns on a reference example. It is a circuit diagram which shows the write scanner which concerns on an Example. It is a schematic diagram which shows the output stage of the light scanner which concerns on an Example. It is a block diagram which shows the whole structure of the display apparatus which concerns on an Example. FIG. 14 is a circuit diagram showing a configuration example of a discrete circuit included in the embodiment shown in FIG. 13. It is a circuit diagram which shows the other structural example of a discrete circuit. It is a wave form diagram which shows the output waveform of a discrete circuit. It is a circuit diagram which shows the structural example of the drive scanner contained in the display apparatus which concerns on this invention. FIG. 18 is a timing chart for explaining operations of the drive scanner shown in FIG. 17. FIG.

Explanation of symbols

  0 panel, 1 pixel array section, 2 pixels, 3 horizontal selector, 4 light scanner, 5 drive scanner, 8 drive board, 9 discrete circuit

Claims (7)

It consists of a pixel array part and a drive part that drives it,
The pixel array unit includes a row-shaped scanning line, a column-shaped signal line, a pixel arranged at a portion where these intersect, a power supply line and a ground line for supplying power to each pixel,
The driving unit includes a scanner that sequentially supplies a control signal to each scanning line to scan the pixels line by line.
The pixel includes at least a light emitting element, a sampling transistor, a drive transistor, and a pixel capacitor.
The gate of the sampling transistor is connected to the scanning line, one of the source / drain is connected to the signal line, the other is connected to the gate of the drive transistor,
The drive transistor and the light emitting element are connected in series between the power supply line and the ground line to form a current path, and power supply from the power supply line to the drive transistor is for switching between a non-light emitting period and a light emitting period of the pixel. Controlled by a power supply control means having a transistor for performing a switching operation ,
One end of the pixel capacitor is connected to the gate of the drive transistor, and the other end of the pixel capacitor is connected between the drive transistor and the light emitting element.
Prior to the light emission period, a video signal is supplied from the signal line to the gate of the drive transistor through the sampling transistor, and under the control of the power supply control means, current is supplied to the pixel capacitor through the drive transistor, and the source of the drive transistor In the control signal supplied by the scanner while the potential rises, the waveform at the time of turning off the sampling transistor is inclined so that the sampling transistor of the pixel whose luminance level of the video signal held in the pixel capacitor is high A display device that shortens the turn-off time. 2. The display device according to claim 1 , wherein the power supply control means comprises a switching transistor connected between the power supply line and the drive transistor. Scanner, when ramping the waveform of the control signal, in at least two separate stages, the steeper the slope at the beginning, a display device according to claim 1 or claim 2, gentle inclination later. The drive unit includes a power pulse generation circuit that generates a power pulse that is a source of the waveform of the control signal and supplies the power pulse to the scanner,
The scanner, in turn, retrieves the waveform of the control signal from the power supply pulse generation circuit, a display device according to claim 1 or claim 2 for supplying to each scanning line. The sampling transistor is turned on according to the control signal supplied from the scanning line, then turned off according to the waveform of the control signal, and the video signal supplied from the signal line is sampled during the period from turning on to turning off. And the luminance level is held in the pixel capacity,
The drive transistor, in response to the video signals held in the pixel capacitance, flow to the light emitting element a drive current through the current path, the display device according to claim 1 or claim 2 emit light at a luminance level. It consists of a pixel array part and a drive part that drives it,
The pixel array unit includes a row-shaped scanning line, a column-shaped signal line, a pixel arranged at a portion where these intersect, a power supply line and a ground line for supplying power to each pixel,
The driving unit includes a scanner that sequentially supplies a control signal to each scanning line to scan the pixels line by line.
The pixel includes at least a light emitting element, a sampling transistor, a drive transistor, and a pixel capacitor.
The gate of the sampling transistor is connected to the scanning line, one of the source / drain is connected to the signal line, the other is connected to the gate of the drive transistor,
The drive transistor and the light emitting element are connected in series between the power supply line and the ground line to form a current path, and power supply from the power supply line to the drive transistor is for switching between a non-light emitting period and a light emitting period of the pixel. Controlled by a power supply control means having a transistor for performing a switching operation ,
One end of the pixel capacitor is connected to the gate of the drive transistor, and the other end of the pixel capacitor is connected between the drive transistor and the light emitting element.
Prior to the light emission period, the video signal is supplied from the signal line to the gate of the drive transistor via the sampling transistor, and under the control of the power supply control means, the current is supplied to the pixel capacitor via the drive transistor. While the source potential rises, the control signal supplied by the scanner has a slope in the waveform when the sampling transistor is turned off, so that sampling of pixels with a high luminance level of the video signal held in the pixel capacitor is performed. A method for driving a display device, in which a transistor is turned off more quickly. 7. The display device driving method according to claim 6 , wherein the power supply control means comprises a switching transistor connected between the power supply line and the drive transistor.

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