JP2009124878A - Power transmitter, and power transmitting device and power receiving device for power transmitter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a power transmitter that includes a capacitor, having optimal power-factor improving performance at a high frequency, and that has excellent power transmission performance, a power transmitting device for a power transmitter, and a power receiving device for a power transmitter. <P>SOLUTION: The power transmitter 100 is composed of a power transmitting part 30 at least including an AC power supply 30b for converting DC power into AC power and a power transmitting coil 1, and a power receiving part 400 at least including a load RL and a power receiving coil 2. If dielectric absorption of a power-factor improving capacitor C1 is set as K, a capacitor satisfying a condition that K is ≤1% is equipped in series between the power transmitting coil 1 and the AC power supply 30b so as to transmit power at a frequency of ≥10 kHz. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention includes a separable power transmission unit and a power reception unit, a power transmission device that transmits power by a mutual induction effect generated between a power transmission coil of the power transmission unit and a power reception coil of the power reception unit, and a power transmission device of the power transmission device And a power receiving apparatus.

  The power transmission device in which the power transmission coil and the power reception coil can be separated is in a separated state in which the distance between the coils is separated when power transmission is not performed. For example, at the time of power transmission, as shown in FIG. 114, the power transmission coil 1 and the power reception coil 2 are configured to face each other. When an alternating current is passed from the power transmission control circuit 3 to the power transmission coil 1 via the capacitor C1, an electromotive force is induced in the power receiving coil 2 by the mutual induction action, and the alternating current due to the electromotive force flows to the load RL through the power reception control circuit 4. Power transmission is performed.

  The power transmission coil 1 or the power reception coil 2 is configured by, for example, winding the conductor 1x shown in the plan view of FIG. 115A in a spiral shape, and along the line 6B-6B of FIG. 115A (FIG. 115B). As shown in the cross-sectional view shown in FIG. Since the two coils formed by winding the conductor 1x in a spiral shape are opposed to each other as shown in FIG. 115B, it is synonymous with inductively coupling the two coils. It is assumed that both coils are in an inductively coupled state.

  In FIG. 115B, the same power transmission coil 1 and power reception coil 2 are used. This is because in the conventional example cited below, the opposing state indicating inductive coupling uses the same coil for the power transmission coil 1 and the power reception coil 2. Naturally, a coil in which the power transmission coil 1 and the power reception coil 2 are different can be used. Hereinafter, when it is simply indicated as “coil” including the conventional example, it refers to the power transmission coil 1 or the power reception coil 2 or both coils. Moreover, in this application, since the vocabulary used differs with the literature referred, vocabulary is demonstrated.

  A portion including the power transmission control circuit 3 and the power transmission coil 1 in FIG. 114 is denoted as a power transmission side, a primary side, an input side, and the like, and the power transmission coil 1 is represented by a power transmission coil, a power transmission coil, a primary coil, a primary side coil, and the like. Is written. In addition, a part including the power reception control circuit 4 and the power reception coil 2 in FIG. 114 is expressed as a power reception side, a secondary side, an output side, and the like, and the power reception coil 2 is a power reception coil, a power reception coil, a secondary coil, and a secondary side. Indicated as a coil. Next, a resistance component existing in series in an equivalent circuit of a coil or a capacitor is generally expressed as ESR, and is called Equivalent Series Resistance in Japanese. In this application, reference is made to “Effective Series Resistance” because the frequency characteristics of ESR are referred to.

  A power transmission device using a coil having a configuration as shown in FIG. 115 is described in Japanese Patent Laid-Open No. 8-148360 (Patent Document 1). This Patent Document 1 describes a donut-shaped planar spiral coil as Comparative Example 1. In other words, this coil was made by winding 5 turns of 100 copper wires with an insulation coating having a diameter of 100 μm, and making them with an outer diameter of 30 mm, an inner diameter of 15 mm, and a thickness of 1.5 mm. Not equipped with magnetic material. The side that is connected to the power supply with these facing each other is the primary side (input side), and the side that generates output by the mutual induction action is the secondary side (output side).

  Moreover, in the Example of patent document 1, the measurement data in which the power transmission frequency is 100 kHz is described, and it is described that the power transmission frequency is not limited to 100 kHz. That is, paragraph number 0040 of Patent Document 1 describes that the power transmission frequency can be arbitrarily selected.

  Another example of the coil having such a configuration is described in Japanese Patent Laid-Open No. 4-122007 (Patent Document 2). In this Patent Document 2, as a comparative example 1, there is a flat spiral coil, in which an enameled copper wire having a diameter of 1 mm is wound for 25 turns, an outer diameter is 80 mm, an inner diameter is 24 mm, and a magnetic core is not provided. Are listed. The one that is connected to the power supply with these facing each other is the primary side (input side), and the one that generates output by the mutual induction action is the secondary side (output side).

  In the air-core coil, the coil characteristics (inductance and effective series resistance) change due to the proximity of the metal. In order to prevent this influence, many inventions have been filed in which a coiled antenna is equipped with a combination of a magnetic material and a metal plate. However, the coiled antenna transmits energy by electromagnetic waves and does not use a mutual induction action. In a power transmission device using a mutual induction action, Japanese Patent Application Laid-Open No. 2006-42519 (Patent Document 3) is formed of a magnetic material on the opposite side of a plane spiral coil for the purpose of eliminating unnecessary radiation. And a coil having a structure in which a metal plate is attached to the opposite side of the opposing surface of the coil of magnetic material.

  Furthermore, Japanese Patent Laid-Open No. 2002-353050 (Patent Document 4) describes that a diamagnetic metal is used as a metal material that eliminates the influence of a metal material approaching the coil. Also in patent document 4, a coil is described as "magnetic field type antenna", and the coil used for a mutual induction effect and the antenna which transmits / receives electromagnetic waves are confused.

  116 is a simplified two-terminal equivalent circuit diagram when power transmission coil 1 is viewed from power transmission control circuit 3 shown in FIG. In FIG. 116, Le is the residual (leakage) inductance of the two-terminal circuit. The value of the residual inductance Le (H) theoretically includes the self-inductance of the power transmission coil 1 alone, the self-inductance of the power reception coil 2 alone, the coupling coefficient between the power transmission coil 1 and the power reception coil 2, and the load resistance connected to the power reception coil 2. The value is determined by the effective series resistance value of the power receiving coil 2 described later.

  Similarly, in FIG. 116, Re is a pure resistance component of a two-terminal circuit. The value of the pure resistance component Re (Ω) theoretically includes the self-inductance of the power transmission coil 1 alone, the self-inductance of the power reception coil 2 alone, the coupling coefficient between the power transmission coil 1 and the power reception coil 2, and the load connected to the power reception coil 2. The resistance value is determined by the effective series resistance value of a power transmission coil 1 and a power reception coil 2 described later.

  The self-inductance L1 (H) of the single power transmission coil 1 and the self-inductance L2 (H) of the single power reception coil are constant values. As will be described later, the residual inductance Le () varies with the variation in the coupling coefficient between the power transmission coil 1 and the power reception coil 2, the variation in the load resistance value connected to the power reception coil 2, and the change in the effective series resistance of the power reception coil 2 due to the power transmission frequency. H) changes. Further, depending on the coil configuration, the residual inductance Le (H) changes with changes in the inductance of the power transmission coil 1 and the inductance of the power reception coil 2 when the power reception coil 2 faces the power transmission coil 1. Therefore, the value of the residual inductance Le is larger or smaller than the value of L1.

  Similarly, with a change in the coupling coefficient between the power transmission coil 1 and the power reception coil 2, a change in the load resistance value connected to the power reception coil 2, a change in the effective series resistance of the power transmission coil 1, and a change in the effective series resistance of the power reception coil 2 The pure resistance component Re (Ω) changes. Due to the inductance L1 (H) of the power transmission coil 1 alone, the inductance L2 (H) of the power reception coil 2 alone, and the coupling coefficient between the power transmission coil 1 and the power reception coil 2, the value of the pure resistance component Re is larger than the value of the load RL. Become smaller.

  The residual inductance Le causes reactive power and reduces the power factor. Therefore, as shown in FIG. 116, the power factor may be improved by canceling the positive residual reactance component caused by the residual inductance Le using the capacitor C1. FIG. 116 is a two-terminal equivalent circuit when a capacitor is connected in series only to the power transmission coil 1. The two-terminal equivalent circuit when a capacitor is connected in series only to the power receiving coil 2 or the two-terminal equivalent circuit when a capacitor is connected in series to both the power transmitting coil 1 and the power receiving coil 2 is described in the embodiment. explain.

  Many patents for single capacitors with good characteristics have been filed. However, in the power transmission device, there is hardly any document that mentions the configuration and characteristics of an appropriate capacitor that is used to cancel the positive residual inductance Le and improve the power factor. Patent Document 4 described above only describes that a film capacitor having a low dielectric loss tangent is used as a capacitor used in the power transmission device.

Next, terms relating to the capacitor will be described. Capacitors that are capacitive elements are often written as capacitors in Japanese. In the present application, including the case of quoting documents, the notation is all unified as “capacitor”. In general, the capacitance that is the capacitance value of the capacitor is often expressed as capacitance. However, since the notation of capacitance is easily confused with a capacitor, the notation of “capacitance” is all unified. Furthermore, the relative dielectric constant, which is the ratio to the dielectric constant in vacuum, is often simply expressed as “dielectric constant”. The dielectric constant εo, which is a physical constant, is a dielectric constant in a vacuum, and εo = 8.85 × 10 ⁻¹² (F / m). In the present application, the configuration of the capacitor is obtained using the dielectric constant εo in vacuum. Therefore, the relative dielectric constant εs, which is the ratio to the dielectric constant εo in vacuum, is unified as “relative dielectric constant” or εs.

  In the power transmission device in which the power transmission unit and the power reception unit can be separated, a capacitor can be connected to the power transmission coil 1 or the power reception coil 2 in series or in parallel. Therefore, a resonance circuit is formed by the power transmission unit alone or the power reception unit alone. In this technical field, most of the literature focuses on the resonance frequency of the power transmission unit and the power reception unit alone. Patent Document 4 also describes that the resonance frequency of the power reception unit is set lower than the resonance frequency of the power transmission unit. However, Patent Document 4 does not describe at all that a series circuit in which a coil and a capacitor are connected in series has the above-described effect of power factor improvement.

  Furthermore, Japanese Patent Laid-Open No. 4-286438: US Pat. No. 5,126,589 (Patent Document 5) also describes that the resonance frequency of a single power transmission unit is set higher than the resonance frequency of a single power reception unit. This is the same as the relationship between the resonance frequencies of the power reception unit and the power transmission unit in Patent Document 4.

  Japanese Patent Application Laid-Open No. 8-340285 (Patent Document 6) describes that the resonance frequency of a single power reception unit is set higher than the power transmission frequency of the power transmission unit. That is, the description is completely opposite to Patent Document 4 and Patent Document 5 described above.

Further, in a power transmission device based on a mutual induction action, a method for preventing heat generation by a metal body close to a power transmission unit coil, and a method for controlling transmission power by providing a signal transmission method between a power transmission unit and a power reception unit are disclosed in Japanese Patent Application Laid-Open No. 10-101. No. -271713: described in US Pat. No. 5,896,278 (Patent Document 7).
JP-A-8-148360 (paragraph number 0027) JP-A-4-122007 (Tables 1 and 2) JP 2006-42519 A (paragraph numbers 0019, 0020, claim 1, claim 2, FIG. 2, FIG. 3) JP 2002-353050 A (paragraph numbers 0007, 0015, 0017, 0018, 0019, claim 3, claim 5) JP-A-4-286438 (paragraph number 0007, claim 1) JP-A-8-340285 (paragraph number 0094, claim 2) JP-A-10-271713 (paragraph numbers 0005, 0009, etc.)

  A power transmission device in which a power transmission unit and a power reception unit can be separated can transmit power to an apparatus without using an electric wire or a mechanical contact. There are various application uses and advantages when it is possible to send electric power necessary for the operation of electrical equipment and electronic equipment without using electric wires or mechanical contacts. However, in the conventional technology, the configuration and characteristics of the power transmission coil that transmits power using the mutual induction effect, and the function and effect are not clarified. Therefore, a power transmission device in which the power transmission coil 1 and the power reception coil can be separated and a conventional example related to the coil of the power transmission device will be considered.

  First, Patent Document 1 describes that the power transmission frequency can be arbitrarily selected. However, the power transmission means is a transformer. Although the primary and secondary coils are inseparable, it is clear that a transformer designed for a 50 Hz to 60 Hz commercial power supply cannot be used at any frequency, for example, 5 Hz or 10 kHz. That is, the transformer which is a power transmission means has a lower limit and an upper limit of usable frequencies. However, there is no conventional technique that considers a frequency range that can be used as a power transmission coil.

  Further, in a transformer in which the primary coil and the secondary coil cannot be separated, the coupling coefficient between both the coils is in a tightly coupled state. On the other hand, a transformer capable of separating the primary coil and the secondary coil is in a loosely coupled state in which the coupling coefficient between the two coils is at most about 0.9. Therefore, the coil described as an Example in patent document 1 and patent document 2 equips the coil wound by planar spiral shape with a magnetic material, and is trying to ensure the coupling coefficient between both coils. That is, the coils described in Patent Document 1 and Patent Document 2 are both comparative examples, and when an air-core planar spiral coil is used, performance cannot be improved unless a magnetic material is provided. Can be seen.

  However, the advantage of the planar spiral coil lies in its shape. In particular, if the power receiving coil provided on the device side is not thin, a problem in mounting occurs. In particular, in a small portable device incorporating a secondary battery, the coil volume is required to be as small as possible due to space constraints. In order to improve the power transmission performance, for example, as described in Patent Document 1, a plate material made of a magnetic material must be provided on the opposite side of the opposing surface of the coil. However, in this case, there is a problem that the volume of the coil increases and it is difficult to incorporate the coil into the device.

  Both Patent Document 1 and Patent Document 2 compare the comparative example and the example, and describe that power cannot be transmitted efficiently with an air-core planar spiral coil. However, the reason is not specified.

  Therefore, in Patent Document 1, the disclosure data regarding the air-core coil cited as Comparative Example 1 will be examined. First, the inventor of the present application created the same coil as that disclosed in Patent Document 1, and measured the characteristics of the coil. The coil described as a comparative example in Patent Document 1 is merely winding five turns of a thick conducting wire having a wire diameter of 1.5 mm in which 100 insulation-coated copper wires having a diameter of 100 μm are bundled. For this reason, the self-inductance is as small as about 0.8 μH, and the mutual inductance is also reduced due to the coil shape. Therefore, a power factor falls and apparent power and reactive power become large. Further, since the wire diameter is large and the number of turns is small, there is a problem that the effective series resistance of the coil becomes too small at about 17 mΩ at a frequency of 100 kHz described in paragraph No. 0051 of Patent Document 1.

  FIG. 117 is an equivalent circuit diagram when the coil of Comparative Example 1 described in Patent Document 1 is used for the power transmission coil 1 and the power reception coil 2. Two transformers are used to form a transformer composed of a power transmission coil 1 and a power reception coil 2 as shown in FIG. In this case, at a frequency of 100 kHz, the impedance Z of the primary coil viewed from the AC power supply V side when the load resistance RL is 10Ω is a very small value, Z = approximately 0.6Ω. In FIG. 117 of the present application, the internal resistance of the AC power supply V indicated by R3 is usually 0.5Ω to several tens of Ω. Therefore, when the primary side coil is connected to the AC power source V, the AC power source V becomes close to a short-circuited state. For this reason, the resistance R3 of the AC power source V consumes a considerable amount of power, and the power cannot be efficiently transmitted, and the transmittable power value is reduced.

  Originally, the coil described in Patent Document 1 is intended to secure a self-inductance by equipping a magnetic material on the opposite side of the coil facing surface, confine magnetic flux when the coils face each other, and increase the coupling coefficient. Created with. For this reason, it is not optimized as an air-core coil.

  Next, the reason why the coil of Comparative Example 1 disclosed in Patent Document 2 is inferior in performance with an air core will be described. In the coil configured as in Comparative Example 1 disclosed in Patent Document 2, when the frequency increases, the effective series resistance of the coil increases due to the skin effect and eddy current loss. It is known that this characteristic has a more significant effect as the wire diameter of the single conductor is thicker. The inventor of the present application tried to make a trial of a coil that is almost equivalent to the coil described as Comparative Example 1 in Patent Document 2. As a result, it has been found that at 50 kHz, the effective series resistance of the coil becomes 0.266Ω, which is about three times or more the direct current resistance of the coil of about 0.08Ω.

  114 is indicated by an AC power supply V in FIG. 117, and R3 is an internal resistance of the AC power supply V. R 1 is an effective series resistance of the power transmission coil 1. R2 is an effective series resistance of the power receiving coil 2. RL is a load resistor connected to the power reception control circuit 4.

  If the coil described as the comparative example 1 of patent document 2 is used for both the primary side and the secondary side coil, as shown in FIG. 117, the effective series resistance R1 is connected in series to the AC power source V. And by connecting the effective series resistance R2 in series with the load resistance RL, power loss occurs at least at two locations R1 and R2. The only way to avoid this is to lower the frequency and reduce the effects of the skin effect and eddy current loss. However, when the frequency is lowered, the reactance of the coil decreases. As a result, the impedance Z of the power transmission coil 1 decreases, and excessive apparent power is input to the power transmission coil 1. Then, an excessive current due to the apparent power flows through the power transmission coil 1, and power loss due to the effective series resistance R1 and the partial resistance R3 of the AC power supply occurs. Therefore, in the Example of patent document 2, the magnetic material is equipped in order to ensure the inductance and reactance of a coil, and to reduce apparent power. In order to use a coil with an air core, a coil that can be operated at a high frequency must be realized so as to ensure reactance. That is, a coil having a high frequency and a low effective series resistance R1 may be realized.

  Contrary to patent document 1, although the coil of the comparative example 1 of patent document 2 has high inductance, the effective series resistance of a coil is also high. Therefore, it is not suitable for use with an air core as described above. That is, both the coil described in the comparative example of Patent Document 1 and the coil described in Comparative Example 1 of Patent Document 2 have a low Q in the high frequency region, as will be described later.

  On the other hand, although the coil configuration is different, many inventions for improving the frequency characteristics of the effective series resistance of the coil have been filed. These are intended to suppress the increase in effective series resistance due to the increase in frequency and increase the Q of the coil in a high frequency region where the reactance of the coil becomes large. However, none of Patent Documents 1 to 4 shows any description about inductance, which is an important characteristic of the coil. Even if the configuration of the coil is defined and the increase in effective series resistance is suppressed, if the inductance decreases, the Q of the coil may decrease in some cases. It cannot be said that a coil with good performance could be realized with the configuration rule in which Q decreases.

  In Patent Document 1 and Patent Document 2, the method opposite to the above-described method for reducing the effective series resistance is adopted, and the effective series resistance due to the increase in the frequency is obtained by equipping the coil with a magnetic material having a high magnetic permeability. It is presumed that a method of increasing the Q of the coil by increasing the inductance rather than the increasing rate is used.

  In other words, in order to realize a coil with good power transmission performance, it is possible to secure self-inductance and mutual inductance (coupling coefficient) and to prevent the coil heat generation caused by power loss due to effective series resistance. Coil must be selected. Then, it is necessary to define the coil operating conditions by defining the coil characteristics, and it is not sufficient to simply improve the frequency characteristics of the effective series resistance of the coil.

  As described above, it is a conventional theory that an air-core power transmission coil in which a conducting wire is wound on a flat plate in a single layer spiral shape has poor power transmission performance. Therefore, the power transmission performance is improved by using a magnetic material or the like. There is no prior art that considers the relationship between the effective series resistance and the frequency of the above-described power transmission coil, which is one factor affecting the power transmission performance, together with the configuration of the power transmission coil. In other words, the conventional technology cannot realize a power transmission coil wound in a spiral suitable for use in a power transmission device. Moreover, the operating condition of the coil for electric power transmission wound by the spiral shape is not prescribed | regulated. Therefore, a power transmission device with good power transmission performance cannot be realized. The first problem in this field is that an optimal power transmission coil used in the power transmission device described above cannot be realized.

  On the other hand, paragraph number 0021 of Patent Document 3 describes that the magnetic sheet eliminates unnecessary radiation due to the “magnetic field” generated by the coil. And it is described that a metal sheet excludes the unnecessary radiation by the "electric field" which a coil generates.

  As described above, when the mutual induction action is used, energy is transmitted not by electromagnetic waves but by “magnetic flux”. That is, the power transmission coil 1 and the power reception coil that are inductively coupled constitute a transformer. The transformer is equipped with a core, so that the coupling coefficient can be approximately 1, and in this case, the power factor is approximately 1. On the other hand, in the power transmission device in which the power transmission coil 1 and the power reception coil can be separated, the coupling coefficient between the two coils is in a loosely coupled state with a maximum of 0.9. Therefore, the power factor is reduced to about 0.2. When the coupling coefficient is smaller than 1, a leakage magnetic flux is generated. However, as will be described later, the presence of the leakage magnetic flux does not cause a loss of energy. Therefore, Patent Document 3 also confuses power transmission by mutual induction and power transmission by electromagnetic waves.

  The sheet of magnetic material, like the transformer core, has the effect of confining the magnetic flux and increasing the coupling coefficient. This is also specified in (Action) at the lower right of page 2 of Patent Document 2. As far as the present inventors have verified, it has been confirmed that when a magnetic material plate is mounted on a coil, the inductance of the coil increases and the effective series resistance also increases. A magnetic material plate formed by molding at least magnetic material powder changes the coil characteristics (inductance and effective series resistance) when a metal material comes close to the opposite surface of the coil. That is, another problem in this field is that when the metal plate comes close to the air-core coil, the coil characteristics change whether the coil operates as a transformer or as an antenna.

  However, as in Patent Document 1 and Patent Document 2 described above, many attempts have been made to improve the power transmission performance by using a magnetic material, but a power transmission device with good performance has not been realized. As will be described later, this is because, as in Patent Document 1 to Patent Document 4, a high-performance power transmission device cannot be realized simply by defining a specific configuration of the coil.

  Furthermore, paragraph number 0019 of Patent Document 4 describes that a metal plate is disposed between the primary coil and the secondary coil, but this is presumed to be an error on the opposite side of the coil facing surface. Paragraph No. 0020 of Patent Document 4 describes a problem with an air-core coil in which, when a metal body is close to a coil (magnetic field antenna), power transmission performance is affected. As the material of the metal plate used to solve the problem, Paragraph No. 0019 of Patent Literature 4 includes diamagnetic metals such as copper (Cu), zinc (Zn), lead (Pb), and bismuth (Bi). Are listed. However, bismuth of atomic number 57, which is a diamagnetic metal, is described, although bismuth, which is a rare earth element of atomic number 83 and all the isotopes except one are all radioactive elements and is not normally used. There is no description of (La) or gold (Au), which is a plate magnetic metal having an atomic number of 79 and is a commonly known noble metal.

  As a result of a further trial by the inventor of the present application, it has been found that it is not the magnetic properties of the metal, but the thickness of the metal plate, except for the ferromagnetic metal, that affects the coil characteristics. Although details will be described later, a high-performance power transmission device that has solved the above-described problems has not been realized yet. However, these problems cannot be solved unless a coil having a good power transmission performance with an air core, which is the first problem, cannot be realized. In that sense, it cannot be said that the solving means described in Patent Literature 1 and Patent Literature 2 realize a practical power transmission device.

  In paragraph No. 0017 of Patent Document 4, the dielectric loss tangent tan δ is small as a capacitor for canceling the residual reactance Le of the power transmission coil 1 and improving the power factor (described as being for resonance in Patent Document 4). It is described that a capacitor and a capacitor having a dielectric material such as polystyrene or polypropylene are used. However, a capacitor using polystyrene or polypropylene as a dielectric does not always have a smaller dielectric loss tangent tan δ at any frequency than a capacitor using other dielectrics.

  Furthermore, when the inventors of the present application made additional trials, some polystyrene capacitors could not be used due to heat generation. In addition, the polypropylene capacitor has different power transmission performance depending on the configuration, and a phenomenon in which the power transmission performance decreases depending on the capacitance and frequency is observed.

  FIG. 118 is a diagram illustrating an equivalent circuit of a capacitor. In FIG. 118, Xc represents the reactance of the capacitor, Rc represents the effective series resistance of the capacitor, and Lc represents the parasitic inductance of the capacitor. The dielectric loss tangent tan δ is defined as tan δ = 1 / Q (no unit) using the Q of the capacitor. The Q of the capacitor is represented by Q = Xc / Rc, where Xc is the reactance of the capacitor and Rc is the effective series resistance of the capacitor at the frequency at which the Q of the capacitor is measured. According to JIS regulations, the dielectric loss tangent tan δ of a capacitor is measured at 1 kHz or 1 MHz by capacitance. When the capacitance is C (F), the reactance Xc (Ω) of the capacitor is Xc = 1 / ωC (Ω), and Xc is an angular frequency ω (ω = 2πf, f is a frequency (Hz)). And the capacitance C is the inverse of the product.

When the symbol in FIG. 118 is used, the dielectric loss tangent tan δ of the capacitor is tan δ = ωC · Rc = Rc · Xc. The capacitance C of the capacitor is usually 10 ⁻⁶ F or less. Therefore, at a low frequency such as 1 kHz, the value of ωC is a very small value of 10 ⁻⁶ or less. On the other hand, the effective series resistance Rc of the capacitor differs depending on the type of capacitor and the capacitance, but is usually 10Ω or less at 1 kHz. Therefore, the value of the dielectric loss tangent tan δ of a general capacitor is usually tan δ <0.001, which is a JIS prescribed value described later.

  In general, a capacitor having a low effective series resistance Rc (Ω) in a high frequency region is a ceramic capacitor. Therefore, the ceramic capacitor has a high Q and a low dielectric loss tangent tan δ. The inventors of the present invention actually measured the Q of a multilayer ceramic capacitor having a capacitance of 0.1 μF at 1 MHz. It was 3 degrees.

  Further, when the Q of a polypropylene capacitor having a capacitance of 0.1 μF, which is different from a polypropylene capacitor having a good power transmission performance described later, was measured at 1 MHz, Q≈5 and tan δ = 1 / 5≈0. 2. The tangent angle δ was δ≈11.3 degrees. That is, as long as the dielectric loss tangent tan δ in the high frequency region is compared, the ceramic capacitor has better characteristics than the polypropylene capacitor. However, as far as the inventor of the present application experimentally verified, the multilayer ceramic capacitor has a lower power transmission performance than the polypropylene capacitor.

  Patent Document 4 simply describes that “dielectric loss tangent is small”. If it conforms to the JIS standard, as described above, an optimum capacitor for the power transmission device can be selected simply by measuring the dielectric loss tangent at 1 kHz and comparing the performance. However, as will be described later, the optimum capacitor for the power transmission device cannot be selected only by the dielectric or tangent of the capacitor. That is, it is necessary to specify a specific value of dielectric loss tangent considering frequency characteristics, a specific capacitance, a relationship between both, and a characteristic factor other than the dielectric loss tangent.

  Therefore, the dielectric loss tangent tan δ of the capacitor used for improving the power factor of the power transmission device needs to be determined in consideration of the dielectric (capacitor type), capacitance, frequency characteristics, and the like. . The frequency characteristics of the dielectric loss tangent tan δ will be described later.

  Further, paragraph number 0018 and claim 3 of Patent Document 4 describe a method of connecting two capacitors to both ends of a coil. These are also described in FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6396, FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6459, FIG. 1 of Japanese Patent Application Laid-Open No. 2005-6460, and the like. However, although this circuit configuration has the effect of increasing the withstand voltage of the capacitor, there is no prior document that mentions the correlation between this circuit configuration and power transmission performance.

  That is, in the conventional technology, the configuration and characteristics of the capacitor used for power factor improvement of the power transmission device, the circuit configuration and the operation effect using the power factor improvement capacitor of the power transmission device are not clarified at all, Therefore, a power transmission device with good power transmission performance cannot be realized. In other words, the performance evaluation method and the usage method (connection method) of the capacitor used in the high frequency power circuit have not been established. This is the second problem in this field.

  Patent Document 4 describes that the resonance frequency of the power receiving unit alone is set lower than the resonance frequency of the power transmission unit alone. Further, Patent Document 5 also describes that the resonance frequency of the power transmission unit alone is set higher than the resonance frequency of the power reception unit alone. Patent Document 5 describes that even when an inductance change of the power transmission coil 1 occurs, the resonance coupling with the power receiving coil can be maintained. However, Patent Document 5 does not clearly describe the cause of the inductance change of the power transmission coil 1. On the other hand, Patent Document 6 describes that the resonance frequency of a single power receiving unit is set to be higher than the power transmission frequency, and describes a method that is completely opposite to the above-described previous case.

  In each of the preceding examples described above, a resonance circuit is constituted by two separable two-terminal circuits, and the relationship between the two resonance frequencies is defined. Assuming that the resonance frequency of the power receiving unit alone is fk (Hz) and the resonance frequency of the power transmission unit alone is fh (Hz), as described above, fk> fh depending on the preceding case, and fh in other preceding cases. The opposite provision is made such that> fk. These differences are presumed to be due to the fact that the principle of the transformer and the operating principle of the power transmission device are not accurately grasped in the prior art.

  In particular, the residual inductance Le (H) shown in FIG. 116 is the configuration of the power transmission coil 1 and the power reception coil 2 (air core, cored, shape, etc.), the inductance of the power transmission coil 1 and the power reception coil 2, the power reception coil and the power transmission coil. It varies depending on various factors such as the coupling coefficient between 1, the effective series resistance of the power receiving coil, and the load resistance value on the power receiving side.

  In FIG. 116, Le = L1 (H) in the power transmission coil 1 alone. However, as will be described later, when the power receiving coil faces the power transmitting coil 1, both Le> L1 and Le <L1 are satisfied. When a resonance circuit is provided on the power transmission side, the resonance frequency fh (Hz) of the power transmission side alone and the resonance frequency fe (Hz) on the power transmission side when the power reception coil faces the power transmission coil 1 are different. If Le> L1, fe <fh, and if Le <L1, fe> fh. These will be described later with examples.

  In addition, FIG. 2 of Patent Document 4 does not have a circuit configuration that allows a current to flow in both directions, such as flowing out current and drawing in current. Therefore, since this circuit cannot drive a circuit in which a coil and a capacitor are connected in series, the circuit of FIG. 2 of Patent Document 4 does not operate.

  In this technical field, technical development has been carried out for the purpose of improving transmission efficiency and transmission power. However, as described above, in an actual power transmission device, power loss occurs due to the effective series resistance of the coil and the capacitor and the output impedance Zs of the AC power supply 30b. In the conventional technique, when the transmission power is increased, the coil, the capacitor, and the AC power supply 30b generate heat due to the power loss, so that the power transmission performance cannot be improved.

  As described above, in a power transmission device in which a power transmission unit and a power reception unit are separable, by providing a capacitor, independent resonance circuits are formed by the power transmission unit alone and the power reception unit alone. However, the basic circuit theory that is an equivalent circuit on the power transmission side when the power transmission coil and the power reception coil are inductively coupled is not disclosed at all in the prior art in this technical field. It is necessary to describe an equivalent circuit of an accurate transformer to which a load resistor is connected to clarify the difference from the circuit theory, and to examine an equivalent circuit of a simplified transformer having two terminals as shown in FIGS. There is.

  In addition, FIG. 2 of Patent Document 4 does not have a circuit configuration that allows a current to flow in both directions, such as flowing out current and drawing in current. Therefore, since this circuit cannot drive a circuit in which a coil and a capacitor are connected in series, the circuit of FIG. 2 of Patent Document 4 does not operate.

  As described above, a circuit configuration in which capacitors are connected so as to cancel the positive reactance caused by the residual inductance Le (H), a configuration of a circuit that drives a two-terminal circuit, a resonance frequency of a single power transmission circuit, and a two-terminal circuit are driven Of the conventional example such as the relationship of the frequency to be transmitted, the relationship between the residual inductance Le (H) and the inductance L1 (H) of the power transmission coil 1 alone, the effective series resistance of the coil and the capacitor, and the output impedance Zs of the AC power supply 30b. There remain many problems in this technical field that cannot be solved by specifying the resonance frequency of the power transmission unit and power reception unit alone. The solutions to these problems are described in detail below.

  Further, as described in paragraph No. 0005 of Patent Document 5, when an alternating current flows through the power transmission coil 1, if a metal body approaches the power transmission coil 1, the vortex described in Paragraph No. 0019 of Patent Document 4 will be described. The metal body generates heat due to the current. This is the same principle as an electromagnetic cooker.

  In Patent Document 5, when the power transmission unit is in a standby state, power is intermittently supplied to the power transmission coil 1. When a regular power receiving unit is attached to the power transmitting unit, the power transmitting unit continuously supplies power to the power transmitting coil 1 by a signal from the power receiving unit. However, in patent document 5, the core is equipped in the power transmission coil 1 and the receiving coil. As will be described later, the coupling coefficient decreases in a power transmission coil equipped with a core. Therefore, a power factor will also fall. There is also a patent document that employs the same technique as Patent Document 5 in a power transmission device using a planar air-core coil. There is also a patent document including means for detecting a metal body by a single power transmission unit.

  As described above, another problem in this field is that the metal body generates heat due to overcurrent loss or hysteresis loss when the metal body is close to the power transmission coil 1. Furthermore, the power transmission unit must determine when a normal power receiving coil to which a metal body and a load are connected faces each other, which is another problem in this field.

  However, the fundamental problem is that, as in Patent Documents 1 to 4, a coil having good power transmission performance cannot be realized simply by defining a specific configuration of the coil. In patent document 1 and patent document 2, the effect is claimed by showing an example in which a specific configuration of the coil is defined. However, an equivalent effect is not always obtained until a configuration factor other than the specific configuration of the coil changes. In other words, not only the specific configuration of the coil but also the characteristics of coils with various configurations are defined, and a coil and power transmission device with good power transmission performance are realized unless a coil with good power transmission performance is selected. I can't do it.

  For example, in the non-contact transformer described in Patent Document 2, it is obvious that the embodiment as in Patent Document 5 cannot be applied. That is, a power transmission coil with good power transmission performance, a power transmission device with good power transmission performance, and a power transmission coil whose characteristics do not vary due to the proximity of a metal body can be realized as in safety measures as in Patent Document 7. There is. In other words, first, it is necessary to solve the first problem.

  Next, clarify the configuration and characteristics of the capacitor used for power factor improvement of the power transmission device, the circuit configuration and operation effect using the power factor improvement capacitor of the power transmission device, and provide a power transmission device with good power transmission performance. It is necessary to solve the second problem of realizing.

  Accordingly, an object of the present invention is to provide a power transmission device with good power transmission performance by defining dielectric absorption of a capacitor.

  The present invention is configured such that an AC power source that is power conversion means for converting DC power to AC power, a power transmission unit including at least a power transmission coil, a load, and a power reception unit including at least a power reception coil can be separated. In the power transmission device that transmits power from the power transmission unit to the power reception unit with the power transmission coil and the power reception coil facing each other, at least one of the AC power source and the power transmission coil, and between the load and the power reception coil. Including a series circuit in which at least one nonpolar capacitor is connected in series to cancel the residual reactance component of the power transmission coil and improve the power factor, and the effective series resistance of one of the opposing coils is reduced. Rw (Ω), the effective series resistance of one coil when both ends of the other coil facing one coil are short-circuited is Rs (Ω), and one coil is Rs> Rw, Is set to f1 (Hz), one coil and the other coil are selected so that f1 (Hz) is 100 kHz or more, and the frequency for driving one coil is The frequency is set to less than f1 (Hz).

  In the present invention, a residual reactance component of the power transmission coil is provided by providing a series circuit in which at least one nonpolar capacitor is connected in series between at least one of the AC power source and the power transmission coil and between the load and the power reception coil. Can be canceled to improve the power factor.

  In a preferred embodiment, the dielectric loss tangent tan δ at a predetermined frequency of the capacitor is 1% or less.

  In this example, by considering the frequency characteristic of the dielectric tangent tan δ, not 1 kHz as defined by JIS, and defining the numerical value of the dielectric tangent tan δ at the frequency used for power transmission, a power transmission device with good power transmission performance is obtained. realizable.

  Preferably, the dielectric loss tangent tan δ has a value at 200 kHz of 1% or less.

  In general, the upper limit of the frequency used for power transmission is defined as about 250 kHz due to radio interference. Therefore, a power transmission device with good power transmission performance can be realized by selecting a capacitor having a dielectric tangent tan δ value of 1% or less at 200 kHz and using it as a power factor improving capacitor of the power transmission device.

  Preferably, the temperature change of the capacitance of the capacitor is within ± 5% between 0 ° C. and 80 ° C.

  As the capacitance of the capacitor changes with temperature, the negative reactance that cancels the positive residual reactance changes. Therefore, the reactance of the LC series circuit does not become zero, and a positive or negative reactance remains. If the power transmission device has a configuration in which the frequency of the AC power supply is variable, it can be adjusted to a frequency at which the reactance of the LC series circuit becomes zero. However, the power transmission device having such a configuration must be equipped with a capacitor temperature detection means, and the control circuit becomes complicated. Therefore, the change in capacitance of the capacitor due to temperature needs to be within ± 5% between 0 ° C. and 80 ° C. Naturally, it is preferable that the temperature change of the capacitance of the capacitor is small.

  Preferably, when an AC constant current is passed through the capacitor, the fluctuation of the voltage across the capacitor is less than ± 0.1% after 5 seconds after the current starts flowing, and the measurement time of 5 minutes after the current starts flowing. The capacitance fluctuation rate is 1% or less, and at least one of the following conditions is satisfied: the capacitance fluctuation rate is 0.2% or less in a measurement time of 1 minute after the current starts to flow.

  As in the previous example, when an AC constant current is passed through the capacitor, the voltage across the capacitor is not stable, which means that the capacitance changes over time. When an AC constant current is passed through the capacitor, if the voltage across the capacitor is not stable, the capacitance changes with time. Therefore, in consideration of heat generation due to the effective series resistance Rc of the capacitor, which will be described later, it is necessary that the time during which the voltage across the capacitor stabilizes is 5 seconds or less when an AC constant current is passed through the capacitor.

  Preferably, the output voltage of the AC power supply is a square wave with a duty of 50% whose level changes between zero potential and a predetermined potential, and the frequency at which the reactance Xc of the capacitor and the reactance Xi of the power transmission coil are equal is fr. , The output frequency of the AC power supply is set to fr. At the output frequency fr, the positive peak value of the voltage across the capacitor with respect to the zero potential is Vp, and the negative peak of the voltage across the capacitor with respect to the zero potential is When the value is Vn, the ratio of Vp and Vn is Vp / Vn, and the predetermined coefficient is B, the capacitor satisfies at least Vp / Vn ≦ B.

  In this example, a capacitor with poor power transmission performance that cannot be excluded by the definition of the dielectric loss tangent tan δ can be swept out of the specified value. Therefore, it is possible to select a capacitor with good power transmission performance more accurately. As a result, a power transmission device with good power transmission performance can be realized.

  Preferably, in a series resonance circuit in which a capacitor is connected in series to an AC power source and a reference coil, the output frequency of the AC power source is set to a resonance frequency determined by the inductance of the reference coil and the capacitance of the capacitor, When Q of the series resonance circuit is Qr, when the resonance current is flowing through the series resonance circuit, the output voltage of the AC power supply is Vt, the voltage across the capacitor is Vc, and the step-up ratio for boosting Vt to Vc is H, H = If Vc / Vt, then H> 0.9 × Qr is satisfied.

  Q and Qr of the series resonant circuit have a relationship of 1 / Qr = 1 / Qi + 1 / Qc, where Qi of the reference coil is Qi and Q of the capacitor is Qc. At the series resonance frequency, the reactance of the reference coil and the capacitor becomes zero, and a voltage Qr times the AC power supply voltage is generated at both ends of the reference coil and the capacitor. This is called the boosting effect of the series resonance circuit.

  The boosting effect is determined only by the effective series resistance Ri and reactance Xi of the reference coil, and the effective series resistance Rc and reactance Xc of the capacitor. Therefore, the step-up ratio H can be measured at an arbitrary frequency and the theoretical value Qr can be measured and calculated as long as it is within a frequency range in which the capacitor described later can be used. The numerical values of the boost ratio H and the theoretical value Qr differ depending on the frequency, but if the measured boost ratio H and the theoretical value Qr satisfy the relationship H> 0.9 × Qr, the power transmission performance of the capacitor Is good. By providing a capacitor having such characteristics, a power transmission device with good power transmission performance can be realized.

  Preferably, of the two terminals of the capacitor, the dielectric absorption when a positive voltage is applied to one terminal is Kp, and the dielectric absorption when a negative voltage is applied to one terminal is Kn, Kp> Kn, When the ratio Kr of Kp and Kn is Kr = Kp / Kn, and Kp <Kn, and Kr is Kr = Kn / Kp, the capacitor is 1 <Kr <1 .5 is satisfied.

  In this example, it is possible to select a capacitor having a better power transmission performance than the definition of the dielectric absorption K in the previous invention. As a result, a power transmission device with good power transmission performance can be realized.

  More preferably, the integrating circuit is constituted by a capacitor, and the integrating circuit includes a current source that allows a constant current of the same value to flow in both forward and reverse directions, an initialization unit that zeros the voltage across the capacitor, and an integration time. Pulse number measuring means for measuring at least 1000 counts of reference pulses for measurement, and when the initial voltage of the capacitor is set to zero by the initializing means, the pulse measurement value of the pulse number measuring means is initially set to zero Then, a constant current Ip in the positive direction is passed through the capacitor for a predetermined time T, and after a predetermined time T, a predetermined constant current In in the negative direction is passed through the capacitor, and the voltage across the capacitor is zero. If Tn is the time to become, | Ip | = | In |, where N is the pulse count of T and Nn is the pulse count of Tn, Nn and N Absolute value of selects 0.004 × N counts following capacitor.

  In this example, for example, a double integration type A / D converter is used, and the reference voltage and the input voltage of the A / D converter are made the same. When the output of the A / D converter, for example, the display shows a deviation from the theoretical value 1, the performance of the power factor capacitor used for the power transmission measure can be determined. This A / D converter has a resolution of 1000 counts or more. If the count number is N and the deviation (digit) is 0.004 × N count or less, the power factor improvement performance of the capacitor is good. If the count number N is, for example, 10,000, a more accurate determination can be made. The theoretical value 1 may be 2 n bits, for example, 2047 if it is 11 bits.

  Further, a constant current In in the negative direction is passed through the capacitor for a predetermined time T, and after a predetermined time T has elapsed, a predetermined constant current Ip in the positive direction is passed through the capacitor until the voltage across the capacitor becomes zero. If the time is Tp, and | Ip | = | In |, where the pulse count number of T is N and the pulse count number of Tp is Np, the absolute value of the difference between Np and N is Select a capacitor of 0.004 × N counts.

  In this example, when the reference voltage of the A / D converter is the same as the input voltage, the polarity of the input voltage is inverted. As described above in terms of dielectric absorption, even a nonpolar capacitor has a different display when the same positive voltage and the same negative voltage are applied to the A / D converter. By the above method, the power factor improvement performance of the capacitor can be accurately determined.

  Preferably, the constant current In in the negative direction is supplied to the capacitor for a predetermined time T, and after the predetermined time T has elapsed, the predetermined constant current Ip in the positive direction is supplied to the capacitor, so that the voltage across the capacitor becomes zero. If the time to Tp is | Ip | = | In |, the pulse count number of the predetermined time T is N, the pulse count number of the time Tp until the voltage across the capacitor becomes zero is Nn, , A capacitor whose absolute value of the difference between Np and Nn is 0.003 × N counts or less is selected.

  In this example, when the reference voltage of the A / D converter is the same as the input voltage, the polarity of the input voltage is inverted. As described above in terms of dielectric absorption, even a nonpolar capacitor has a different display when the same positive voltage and the same negative voltage are applied to the A / D converter. By the above method, the power factor improvement performance of the capacitor can be accurately determined.

  Preferably, when the effective series resistance of the power transmission coil is Ri and the effective series resistance of the capacitor is Rc, the power transmission unit transmits the power at a minimum frequency that satisfies Ri> Rc.

  By defining the minimum frequency, it is possible to define the operating conditions of the capacitor according to each of the inventions described above. As a result, a power transmission device with good power transmission performance can be realized.

  Preferably, one of the coils is used as at least one of the power transmission coil and the power reception coil, and the two coils cannot be separated.

  Note that it is possible to measure the characteristics of the power transmission coil and the power reception coil before fixing, and to measure the characteristics of both coils facing each other. Both coils designed as an integral structure from the beginning cannot be confirmed unless they are actually assembled. However, in the embodiment of the present invention, after measuring the characteristics and actually confirming the power transmission performance, the coils are fixed. can do. In addition, it is possible to realize a transformer that is light, thin, and has an air core and good characteristics, in which the winding ratio of the receiving coil to the receiving coil can be set to an arbitrary ratio.

  Further, preferably, it includes power conversion means for converting DC power to AC power, and when the output frequency of the power conversion means is fa (Hz), fa is set to a frequency less than f1.

  In the power transmission device of the present invention, the power transmission unit and the power reception unit can be separated, and the power transmission unit can transmit power to not only a specific power reception unit but also a power reception unit suitable for the power transmission unit.

  Further, preferably, when both ends of the other coil facing one coil are opened, the maximum frequency satisfying Rn (Ω) and Rs> Rn ≧ Rw is set to f2 (Hz ), Fa is set to a frequency less than f2.

  In this example, by satisfying Rs> Rn ≧ Rw at the frequency at which power is transmitted, a coil having a smaller effective series resistance Rw (Ω) can be selected, and an optimum frequency range for power transmission can be defined. In addition, by using a coil that satisfies the condition of Rs> Rn ≧ Rw at the frequency at which power is transmitted, both the single coil and the transformer facing the coil have ideal theoretical characteristics. As a result, it becomes possible to improve the power transmission performance as compared with the prior art.

Further, preferably, the thermal resistance of one coil is θi (° C./W), the allowable operating temperature of one coil is Tw (° C.), the ambient temperature of the place where one coil is installed is Ta (° C.), and the power When the alternating current flowing through one coil is Ia (A), the relationship of Rw ≦ (Tw−Ta) / (Ia ² × θi) is established in fa. Power is transmitted from the power transmission unit to the power reception unit.

  In this way, by defining the thermal conditions based on the effective series resistance Rw (Ω) and the alternating current Ia (A), a turn that determines the upper limit of the alternating current Ia of at least one coil or the effective series resistance Rw of one coil. The upper limit of the number and the frequency region where the effective series resistance Rw (Ω) is small can be defined.

  Another aspect of the present invention is a power transmission device including a power transmission unit of the power transmission device of the present invention in which a capacitor is connected to at least a power transmission coil of the power transmission unit, and the power transmission unit transmits power to the power reception unit.

  In the power transmission device according to the present invention, the power transmission unit and the power reception unit can be separated. The power transmission unit can send power not only to a specific power reception unit but also to a power reception unit suitable for the power transmission unit. The power transmission unit of such a power transmission device is equipped with the capacitor described above.

  Furthermore, power transmission performance can be secured in a wide frequency range by installing a capacitor in the power receiving coil of the power receiving unit. In addition, power transmission performance is also improved by providing a capacitor in the power receiving coil of the power receiving unit.

  Another aspect of the present invention is a power reception device including a power reception unit of the power transmission device of the present invention in which a capacitor is connected to at least a power transmission coil of the power transmission unit, and the power reception unit receives power from the power transmission unit.

  In the power transmission device according to the present invention, the power transmission unit and the power reception unit can be separated. Similar to the previous invention, the power reception unit can receive power from not only a specific power transmission unit but also a power transmission unit suitable for the power reception unit. The power transmission unit of such a power transmission device is equipped with the capacitor described above.

  Furthermore, power transmission performance can be secured in a wide frequency range by installing a capacitor in the power receiving coil of the power receiving unit. In addition, power transmission performance is also improved by providing a capacitor in the power receiving coil of the power receiving unit.

  According to the present invention, by providing a series circuit in which at least one nonpolar capacitor is connected in series between at least one of the AC power source and the power transmission coil and between the load and the power reception coil, Since the power factor can be improved by canceling the residual reactance component, a power transmission device with good power transmission performance can be realized.

(Description of power transmission equipment)
FIG. 1 is a block diagram of a power transmission device 100 according to an embodiment of the present invention. In FIG. 1, a power transmission device 100 includes a power transmission unit 30 that operates as a power transmission device and a power reception unit 40 that operates as a power reception device. The power transmission unit 30 includes a DC power source 12, a power transmission control circuit 30a, a power transmission coil 1, and a capacitor C1 connected in series to the power transmission coil 1. Power reception device 40 includes power reception coil 2, power reception control circuit 40a, and load RL. The power transmission coil 1 and the power reception coil 2 are disposed to face each other.

  The power transmission unit 30 and the power reception unit 40 are configured to be separable. When the power transmission unit 30 and the power reception unit 40 are coupled, the power transmission coil 1 and the power reception coil 2 are arranged to face each other, so that the power transmission coil 1 and the power reception coil 2 function as a transformer.

  The power transmission control circuit 30a of the power transmission unit 30 includes at least an AC power source 30b that is a power conversion unit such as an inverter circuit that converts the DC voltage Vd of the DC power source 12 into AC power. The series circuit in which the power transmission coil 1 and the capacitor C <b> 1 are connected by the AC power is driven by a sine wave, a square wave, or the like below a predetermined frequency to be described later, and power is transmitted to the power receiving unit 40. The power reception unit 40 receives the power transmitted from the power transmission coil 1 by the power reception coil 2. The power reception control circuit 40a supplies the received power to the load RL. The power reception control circuit 40a includes a rectifier circuit that converts AC power into DC power. When the load RL operates with AC power such as an incandescent bulb or LED, the power reception control circuit 40a can be omitted and the load RL can be directly connected to the power receiving coil 2.

  In addition, alternating current means what can send an electric current through the coil connected to the output terminal to the forward direction and a reverse direction. Hereinafter, the power conversion means for converting the DC power source Vd into AC power is referred to as AC power source 30b. The output frequency of the AC power supply 30b is expressed as fa (Hz). Further, a frequency at which the power transmission coil 1 is driven by the AC power supply 30b is expressed as fd (Hz). The frequency at which the power receiving coil 2 receives power is denoted as fj (Hz). In this case, naturally, fa = fd = fj (Hz). fa, fd, and fj are frequencies used for power transmission. fa and fd are parameters of the power transmission unit, and the effect is only the difference between the drive unit and the driven unit. However, the operational effects of fd and fj are different and will be described below.

(Description of operation of power transmission device)
The opposing power transmission coil 1 and power reception coil 2 shown in FIG. 1 are air-core coils, and the effective series resistance of one of the coils alone is Rw (Ω). Let Rs (Ω) be the effective series resistance of one coil when the other coil facing one coil is short-circuited. As will be described later, when the frequency is low, the relationship between Rw and Rs is Rs> Rw. On the other hand, when the frequency increases, the relationship between Rw and Rs is Rs <Rw. The frequency satisfying Rs <Rw varies depending on the coil. That is, there is an upper limit value for the frequency satisfying the relationship of Rs> Rw, and the upper limit value varies depending on the coil. As described above, this upper limit value is a criterion for selecting a coil with good power transmission performance, can define a frequency range in which the coil constituting the power transmission device can be used, and can realize a power transmission device with good power transmission performance. is there.

  Therefore, in the power transmission device 100 according to the embodiment of the present invention, when one coil has a maximum frequency satisfying Rs> Rw as f1 (Hz), the AC power supply included in the power transmission unit 30 The output frequency fa (Hz) is set to a frequency region less than f1 (Hz), and power is transmitted to the power receiving unit 40. When fa (Hz) is set as described above, one coil or the other coil that is the power transmission coil 1 is driven at the frequency fd = fa (Hz). That is, the power transmission coil 1 satisfies the condition of fd <f1. As a matter of course, the power transmission coil 1 satisfies the relationship of Rs> Rw at fd (Hz). Further, the frequency fj (Hz) at which the power receiving coil 2 receives power is conditional on being less than f1 (Hz). That is, the power receiving coil satisfies the condition of fj <f1. As a matter of course, the power receiving coil 2 satisfies the condition of Rs> Rw at fj (Hz).

  As described above, the notation that “the driving frequency fd (Hz) is set to less than f1 (Hz)” of the power transmission coil indicates that the driving frequency fd (Hz) of the power transmission coil satisfies the condition “fd <f1”. It is synonymous with “Yes”. The expression “satisfying the condition of fd <f1” is synonymous with the expression “the power transmission coil satisfies the relationship of Rs> Rw at fd (Hz)”. In addition, the notation that the power receiving coil receives the power “frequency fj (Hz) is less than f1 (Hz)” means that the frequency fj (Hz) at which the power receiving coil receives power is “fj”. It is synonymous with the notation <satisfying the condition of f1. The expression “satisfying the condition of fj <f1” is synonymous with the expression “the power receiving coil satisfies the relationship of Rs> Rw at fj (Hz)”. Since the frequency of the received power cannot be set in the power receiving unit, the frequency fj (Hz) at which the power receiving coil receives power is defined, and f1 of the power receiving coil determined by the power receiving coil and the power transmitting coil exceeds fj Is a condition. Hereinafter, the conditions that the power transmission apparatus should satisfy are defined by any of the above-described notations.

  Furthermore, the effective series resistance of one coil when the other coil facing one coil is opened is Rn (Ω). A maximum frequency satisfying Rs> Rn ≧ Rw is defined as f2 (Hz). As will be described later, f2 (Hz) is lower than f1 (Hz). The power transmission device 100 sets the output frequency fa (Hz) of the AC power supply 30b included in the power transmission control circuit 30a to a frequency region less than f2 (Hz), and transmits power to the power receiving unit 40. When fa (Hz) is set as described above, one coil or the other coil that is the power transmission coil 1 is driven at the frequency fd = fa (Hz).

  That is, the power transmission coil 1 satisfies the condition of fd <f2. As a matter of course, the power transmission coil 1 satisfies the relationship Rs> Rn ≧ Rw at fd (Hz). At this time, the frequency fj (Hz) at which one coil or the other coil as the power receiving coil 2 receives power is set to be less than f2 (Hz). That is, the power receiving coil 2 satisfies the condition of fj <f1. As a matter of course, the power receiving coil 2 satisfies the relationship of Rs> Rn ≧ Rw at fj (Hz). Hereinafter, similarly to the relationship between f1 and fd or the relationship between f1 and fj described above, the relationship between f2 and fd or the relationship between f2 and fj is defined by any notation.

(Description of specific example of coil)
Hereinafter, the specific structure of the coil used for the power transmission device in the embodiment of the present invention will be described. The coil of each embodiment described below is used as the power transmission coil 1 or the power reception coil 2 of the power transmission device 100.

  2A and 2B are diagrams illustrating an example of an air-core coil. FIG. 2A is a plan view, and FIG. 2B is an enlarged view taken along line 1B-1B in FIG. .

  As shown in FIG. 2A, the coil 1a according to an embodiment of the present invention is configured by winding a conductive wire 11 in a flat and air-core single layer spiral shape so that adjacent conductive wires 11 are in close contact with each other. The As shown in FIG. 2 (B), the conductor 11 has a circular cross section, and the maximum diameter d1 (mm) is not particularly limited. Preferably, for example, a single conductor 12 having a wire diameter of 0.2 mm or more is covered with an insulating coating. 13 is applied. The insulation coating 13 may be either a strong coating that is thin like a formal wire or a thick coating such as a vinyl wire.

  Furthermore, the self-inductance of the coil 1a is at least 2 μH or more. Furthermore, let Rw (Ω) be the effective series resistance of the coil 1a alone shown in FIG. Let Rs (Ω) be the effective series resistance of the coil 1a when the other coil facing the coil 1a is short-circuited. At this time, the highest frequency satisfying Rs> Rw is defined as f1 (Hz). The coil 1a which is the power transmission coil 1 or the other coil is driven by the AC power supply 30b at fd (Hz) which is a frequency less than f1 (Hz). The output frequency fa (Hz) of the AC power supply 30b is set to a frequency less than f1 (Hz). In addition, the frequency fj (Hz) at which the coil 1a that is the power receiving coil or the other coil receives power is less than f1 (Hz). Preferably, when the coil 1a is used for both one coil and the other coil, Rs> Rw is satisfied at 100 kHz.

  The reason why the coil outer diameter D (mm) is selected to be 25 times or more of the maximum diameter d1 (mm) of the single conductor 12 is to ensure a necessary coupling coefficient. The reason why the number of turns of the conducting wire 11 is selected to be 8 or more is to obtain a self-inductance of 2 μH or more. Although not only in this embodiment but also in other embodiments, it is desirable to provide the coil with a predetermined inner diameter on which no conducting wire is wound. The inner diameter may be any dimension as long as the outer diameter D is satisfied.

  Furthermore, the effective series resistance of the coil 1d when the other opposing coil is opened is Rn (Ω). At this time, the highest frequency satisfying Rs> Rn ≧ Rw is defined as f2 (Hz). The coil 1a that is the power transmission coil 1 or the other coil is driven by the AC power supply 30b at a frequency fd (Hz) less than f2 (Hz). The output frequency fa (Hz) of the AC power supply 30b is set to a frequency less than f2 (Hz). The frequency fj (Hz) at which the coil 1a, which is the power receiving coil, or the other coil receives power is less than f2 (Hz).

Furthermore, when the thermal resistance of the coil 1a is θi (° C / W), the allowable operating temperature of the coil 1a is Tw (° C), the ambient temperature of the place where the coil 1a is installed is Ta (° C), and power is transmitted When the alternating current flowing in the coil 1a is Ia (A), the relationship Rw ≦ (Tw−Ta) / (Ia ² × θi) is satisfied when the coil 1a is transmitting power. .

  The coil 1a configured as described above can be used as the power transmission coil 1 or the power reception coil 2 of the power transmission device shown in FIG. 1 in which the power transmission unit 30 and the power reception unit 40 can be separated.

  In addition, in embodiment of FIG. 2 (A), conducting wire is wound circularly. However, the shape is not limited to a circle, but an oval shape shown in FIG. 3A, an oval shape shown in FIG. 3B, a square shape shown in FIG. 3C, a rectangle shape shown in FIG. 3D, and FIG. It can be wound in an arbitrary shape such as a polygon such as a hexagon shown in FIG. The same applies to other embodiments described later. However, when the shape of the coil is other than a circle, the coil outer diameter D defines the minimum outer dimension D ′ of the coil as shown in FIGS. 3 (A) to 3 (E).

Next, the relationship described above, Rs> Rw, Rs> Rn ≧ Rw, Rw ≦ (Tw−Ta) / (Ia ² × θi) will be described. In addition, since this description has the same effect also in embodiment of the other coil mentioned later, description is abbreviate | omitted in embodiment described below.

(Explanation of transformer made up of coils)
4 shows an equivalent circuit of the transformer, FIG. 5 shows an equivalent circuit of the single coil, and FIG. 6 shows an equivalent circuit of the single transformer configured as shown in FIG. 93 described in the conventional example. FIG. FIG. 7 is a diagram illustrating an equivalent circuit of the transformer when the secondary coil is short-circuited, and FIG. 8 illustrates an equivalent circuit of the transformer when the load resistor RL is connected to the secondary coil. FIG.

When the power transmission coil 1 and the power reception coil 2 are arranged to face each other, they act as a transformer. Here, in order to refer to circuit theory, the power transmission coil 1 is referred to as a primary coil, and the power reception coil 2 is referred to as a secondary coil. In order to obtain the theoretical relationship between Rw, Rn, and Rs, the impedance Z1 on the primary side of the transformer is obtained in advance. In FIG. 4, L1 (H) is the inductance of the primary side coil, L2 (H) is the inductance of the secondary side coil, M (H) is the mutual inductance between the primary side coil and the secondary side coil, and V1 (V ) Is the voltage across the primary coil, V2 (V) is the voltage across the secondary coil (load resistance RL), I1 (A) is the current flowing through the primary coil, and I2 (A) is the secondary coil. , RL represents a load resistance (pure resistance), and Z1 (Ω) represents a primary side input impedance (complex impedance). In FIG. 4, the following circuit equation is established, and the pure resistance component (effective series resistance) and reactance component (inductance) of Z1 can be obtained by solving the following simultaneous equations. The circuit equation of FIG. 4 is described below. Note that j ² = −1, ω is an angular frequency, and ω = 2πf (f is a frequency, Hz).

V1 = jωL1 · I1 + jωM · I2 (1)
V2 = jωM · I1 + jωL2 · I2 (2)
V2 = −RL · I2 (3)
Since it is desired to obtain Z1 = V1 / I1, V2 and I2 can be eliminated from the above three simultaneous equations. Substituting equation (3) of the above simultaneous equations into equation (2) and eliminating V2,
0 = jωM · I1 + (jωL2 + RL) I2
When the above equation is solved with respect to I2 and substituted into the above equation (1), and I2 is eliminated,
V1 = (jωL1 + ω ² M ² / (jωL2 + RL)) I1
Since Z1 = V1 / I1, from the above equation, Z1 is
Z1 = jωL1 + ω ² M ² / (jωL2 + RL)
It becomes. Since an actual transformer has an effective series resistance R1 in the primary coil and an effective series resistance R2 in the secondary coil, considering the circuit of FIG. 6 and assuming RL = R2,
Z1 = R1 + jωL1 + ω ² M ² / (jωL2 + R2)
It becomes. In the above equation, ω ² M ² / (jωL2 + R2),
Multiplying (−jωL2 + R2) / (− jωL2 + R2) = 1,
Z1 = R1 + jωL1 + ω ² M ² (−jωL2 + R2) / (ω ² L2 ² + R2 ² )
Then, when the real and imaginary terms are arranged,
Z1 = R1 + R2 · ω ² M ² / (ω ² L2 ² + R2 ² ) + jω (L1−L2 · ω ² M ² / (ω ² L2 ² + R2 ² ))
Assuming that A ² = ω ² M2 ² / (ω ² L2 ² + R2 ² ), Z1 is
Z1 = (R1 + A ² R2) + jω (L1−A ² L2) (4)
It becomes. Since ω ² > 0, M ² ≧ 0, L2 ² > 0, and R2 ² > 0, obviously, A ² ≧ 0. That is, in FIG. 6, the input impedance Z1 of the primary coil is
Z1 = R1 + jωL1 (5)
As can be seen from the comparison between the equations (5) and (4), as shown in FIG. 7, when the secondary coil of the transformer is short-circuited, the effective series resistance R1 of the primary coil is It can be seen that the inductance L1 increases and the inductance L1 decreases. These are known circuit theories.

  The above formulas (4) and (5) are basic formulas cited to explain the relationship of Rs> Rw and Rs> Rn ≧ Rw.

  Next, a specific example of the coil 1a shown in FIG. There are some overlaps, but the definition of the symbol is clear. Rw is the effective series resistance of the coil 1a alone (R1 in FIG. 5), and Rn is the effective series resistance of the coil 1a when another coil is opposed to the coil 1a and the opposed coil is opened (in FIG. 6). R1) and Rs are the effective series resistance of the coil 1a when the other coil is opposed to the coil 1a and the opposed coil is short-circuited (R1 in FIG. 7), and kr is approximately obtained from Rw and Rs. The coupling coefficient between the two coils.

  Further, when the inductance of the coil 1a alone is Lw, another coil is opposed to the coil 1a, and the inductance of the coil 1a when the opposed coil is short-circuited is Ls, it is approximately obtained from Lw and Ls. The coupling coefficient is expressed as ki. An approximate method for obtaining kr and ki will be described later.

  When L1 indicates the coil itself, L1 is a symbol, and when L1 is a numerical value of inductance, a unit is added as L1 (H). The same applies to resistors such as R1 and Rw. However, when Rs> Rw, etc. are described with equal signs or inequality signs, when Rw etc. are described before or after when Rw etc. is described in the formula or when there is a description indicating that it is used for calculation , “Rw is 2Ω”, etc. When the unit and the unit are described immediately after Rw, etc., when it is clear that it is a numerical value in the explanation of the characteristic diagram, etc., the unit addition is omitted. ing.

  In the following description, the primary side and the secondary side of the transformer with opposed coils are distinguished, but the transformer can invert the primary side and the secondary side, so that R1 in FIG. , L1 can be obtained as R2 and L2 on the secondary side. That is, the power transmission coil in the embodiment of the present invention may be provided on at least one of the primary side and the secondary side. For example, the same configuration as the coil 1a can be used on the secondary side (device side), and a solenoid-like coil or a honeycomb-shaped multi-layer coil described later can be used on the primary side (power transmission side). Let Rw be the effective series resistance of the coil 1a alone. Let Rs (Ω) be the effective series resistance of the coil 1a when a solenoid-like or honeycomb-like multi-layer wound coil is short-circuited to the coil 1a.

  Also in this case, the solenoid-type or honeycomb-type multi-layer wound coil as the power transmission coil 1 has an AC power supply at a frequency fd (Hz) less than the maximum frequency f1 (Hz) where the coil 1a satisfies Rs> Rw. It is driven by 30b. The output frequency fa (Hz) of the AC power supply 30b is set to a frequency less than f1 (Hz). Further, when a solenoid coil or honeycomb-shaped multi-layer coil is used as the power receiving coil and the coil 1a is used as the power transmitting coil, the frequency fj (Hz) at which the power receiving coil receives power is less than f1 (Hz). On the condition.

  Hereinafter, a specific configuration example of the coil 1a will be described.

(Description of specific configuration example 1A of coil 1a)
FIG. 9 shows effective power transmission when a 10 Ω load resistance is connected to Rw, Rn, Rs of coil 1A and a coil 1A in which a formal wire having a copper wire diameter of 1 mm is closely wound for 25 turns (T) with an outer diameter of 70 mm. It is a figure showing the relationship between efficiency (eta) and a frequency.

  The inventor of the present application refers to the coil described in Patent Document 2 (hereinafter referred to as a conventional example) as the coil 1a shown in FIG. A formal wire was used, and the coil 1A was formed by winding so that adjacent conducting wires were in close contact with each other. As a result, it has been found that when the coil 1A is used for both the power transmission coil 1 and the power reception coil 2, only predetermined power transmission performance can be achieved.

  Therefore, the inventor of the present application has found coils 1B to 1G shown in FIGS. 10 to 17 that have improved transmission performance as compared with the coil 1A. Each of the coils 1B to 1G is configured by using a formal wire in the form of a flat single-layer spiral with an air core like the coil 1a in FIG.

(Description of specific configuration example 1B of coil 1a)
FIG. 10 is a characteristic diagram for explaining the coil 1B.

  The coil 1B is obtained by closely winding a formal wire having a copper wire diameter of 0.6 mm with an outer diameter of 70 mm for 40 turns. FIG. 10 shows the relationship between Rw, Rn, Rs, kr, ki and the frequency of the coil 1B.

(Description of Specific Configuration Example 1C of Coil 1a)
FIG. 11 is a characteristic diagram for explaining the coil 1 </ b> C.

  The coil 1C is obtained by closely winding a formal wire having a copper wire diameter of 0.3 mm with a diameter of 70 mm for 70 turns. FIG. 11 shows the relationship between Rw, Rn, Rs of the coil 1C, the phase angle θ described later, and the frequency.

(Description of Specific Configuration Example 1D of Coil 1a)
FIG. 12 is a characteristic diagram for explaining the coil 1D.

  The coil 1D is obtained by closely winding a formal wire having a copper wire diameter of 0.3 mm for 31 turns with a diameter of 30 mm. FIG. 12 shows the relationship between Rw, Rn, Rs of the coil 1D and the frequency.

(Description of specific configuration example 1E of coil 1a)
FIG. 13 is a characteristic diagram for explaining the coil 1E.

  The coil 1E is formed by sparsely winding a formal wire having a copper wire diameter of 1 mm with an outer diameter of 70 mm and a gap of about 1 mm for 14 turns. The relationship between Rw, Rn, Rs, kr and the frequency of the coil 1E is shown in FIG.

(Description of Specific Configuration Example 1F of Coil 1a)
FIG. 14 is a characteristic diagram for explaining the coil 1F.

  The coil 1F is obtained by closely winding an electric wire (Litz wire) bundled with 75 formal wires having a copper wire diameter of 0.05 mm to an outer diameter of 70 mm for 30 turns. The relationship between Rw, Rn, Rs, kr, ki and frequency of the coil 1F is shown in FIG.

(Description of Specific Configuration Example 1G of Coil 1a)
FIG. 15 is a characteristic diagram for explaining the coil 1G.

  The coil 1G is obtained by closely winding an electric wire (Litz wire) bundled with 75 formal wires having a copper wire diameter of 0.05 mm to an outer diameter of 50 mm for 20 turns. FIG. 15 shows the relationship between Rw, Rn, Rs, kr, ki and the frequency of the coil 1G.

(Examination of each coil)
10 to 15 show the maximum frequency f1 (Hz) satisfying Rs> Rw and the maximum frequency satisfying Rs> Rn ≧ Rw for all of the coils 1B to 1G. f2 (Hz) is shown in common. However, the maximum frequencies f1 (Hz) and f2 (Hz) differ depending on each of the coils 1B to 1G.

  Moreover, the characteristic diagrams shown in FIGS. 10 to 15 are all measured by measuring the distance between the opposing coils at zero. Even if the facing distance between the coils is separated, Rs (Ω) and Rn (Ω) are slightly lower than when the facing distance is zero, but the facing distance is up to about 1/10 of the coil outer diameter D. Almost no change. Actually, when the facing distance increases, the coupling coefficient between the two coils decreases, the reactance of the primary coil increases, and the apparent power increases, so the power factor decreases.

The power loss due to the effective series resistance of the coil can be suppressed by the definition of Rw ≦ (Tw−Ta) / (Ia ² × θi) described later. As will be described later, the values of R1 and R2 in FIG. Since the unknown point and Tw (° C.) and Ta (° C.) differ depending on the use conditions of the coil, in the embodiment of the present invention, the aforementioned Rw, Rs, and Rn are set to zero or the actual facing distance. What is necessary is just to obtain the maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw and the maximum frequency f1 (Hz) satisfying Rs> Rw by measuring at the opposing distance of the coil to be used.

  First, the difference between the case where Rs> Rw is satisfied and the case where Rs> Rw is not satisfied will be described. As described above, it is known that the effective series resistance Rw (Ω) of a single coil increases as the frequency increases, and the skin effect and eddy current loss are known as the cause.

Further, according to the above circuit theory, as shown in FIG. 7, it is known that when the secondary coil is short-circuited, the pure resistance value on the primary side increases to (R1 + A ² R2) Ω. R2 is an effective series resistance value of the secondary coil, M is a value of mutual inductance between the primary coil and the secondary coil, ω is an angular frequency (ω = 2πf, f is a frequency, Hz), and L2 is 2 When the value of the self-inductance of the secondary coil is A ² = ω ² M ² / (ω ² L2 ² + R2 ² ), ω ² > 0, M ² ≧ 0, L2 ² > 0, R2 ² > 0, Therefore, clearly, A ² ≧ 0. As for the primary side inductance, if L1 is the value of the self-inductance of the primary side coil, as shown in FIG. 7, when the secondary side coil is short-circuited, the primary side inductance is (L1-A ² L2) H, is known to decrease.

  However, referring to FIG. 9 to FIG. 11, it can be seen that Rs (Ω) is smaller than Rw (Ω) in the high frequency region. The frequency at which Rs <Rw is about 67 kHz or more in the coil 1A as a comparative example, whereas it is about 208 kHz or more in the coil 1B. In the coil 1C, it becomes about 820 kHz or more. In a coil in which a formal wire is wound in close contact with a flat plate spiral, the maximum frequency f1 (Hz) that satisfies Rs> Rw decreases as the diameter of the formal wire increases. Further, from FIG. 12, in the coil 1D that uses the same single conductor as the coil 1C and is wound 31 turns with an outer diameter of 30 mm, the maximum frequency f1 (Hz) that satisfies Rs> Rw is higher than that of the coil 1C. Yes.

(Explanation of fluctuations in frequency characteristics due to wire diameter)
FIG. 16 shows the frequency and effective series resistance Rw (Ω) of a coil in which each formal wire having a copper wire diameter of 0.2 mm, 0.4 mm, 0.8 mm, and 1 mm is closely wound in a flat plate shape for 25 turns. Showing the relationship.

  As is apparent from FIGS. 9 to 12, a coil having a low maximum frequency f1 (Hz) that satisfies Rs> Rw has a high increase rate of the effective series resistance Rw (Ω) as the frequency increases. From FIG. 16, this characteristic is the same even with coils having different outer diameters in which formal wires having different wire diameters of 0.2 mm, 0.4 mm, 0.8 mm, and 1.0 mm have the same number of turns of 25 times. . That is, it can be seen that as the diameter of the formal wire increases, the rate of increase in the effective series resistance Rw (Ω) accompanying an increase in frequency increases. Further, in a coil wound with the same wire diameter, the smaller the number of turns and the smaller the outer shape, the higher the maximum frequency f1 (Hz) that satisfies Rs> Rw, and the effective series resistance Rw (Ω) due to the increase in frequency. ) Increase rate is also small.

  That is, if the circuit theory is followed, the relationship of Rs> Rn = Rw must be satisfied. However, in the transformer using coils 1A to 1D and configured as shown in FIGS. Then, the relationship of Rs> Rw is not satisfied. For example, in the coil 1B, it can be seen from FIG. 10 that Rs <Rw at a frequency of 208 kHz or higher.

In a frequency region where the relationship between Rw and Rs is Rs <Rw, A ^{2 that} is not positive becomes negative. 9 to 12, the actual values of the effective series resistances R1 and R2 shown in FIG. 8 cannot be obtained in the frequency region where Rs <Rw. An example is shown below. Here, since the coupling coefficient is approximately obtained from the effective series resistance, the coupling coefficient is expressed as kr. As will be described later, the coupling coefficient obtained from the inductance is expressed as ki.

According to the known circuit theory, when the coupling coefficient is kr, the mutual inductance is M (H), the self-inductance of the primary side coil is L1 (H), and the self-inductance of the secondary side coil is L2 (H). Then, the relationship of M ² = kr ² · L1 · L2 is established. If the same coil is used for the primary side coil and the secondary side coil, R1 = R2 = Rw and L1 = L2 = Lw. Therefore, when ω ² L2 ² >> R2 ² is satisfied,
A ² = ω ² M ² / (ω ² L2 ² + R2 ² ) ≈ω ² M ² / (ω ² L2 ² )
= Kr ² · L1 · L2 / L2 ² = kr ² · L1 / L2. Therefore,
From (R1 + A ² R2), (Rw + kr ² Rw) = Rs,
^{kr 2 ≒ (Rs-Rw)} / Rw, approximately prompted kr ² as,
kr = √ ((Rs−Rw) / Rw).

When both coils are the same, R1 = R2 = Rw and L1 = L2 = Lw. Therefore, whether ω ² L2 ² >> R2 ² is satisfied is calculated by calculating the value of ω ² Lw ² / Rw ² , and the value of the coupling coefficient obtained when this value is 50 or more has an error of about 2%. Judgment is as follows. 9 to 15, when 10 kHz to 30 kHz or more, ω ² Lw ² / Rw ² > 50. In the frequency region satisfying Rs> Rw, the coupling coefficient kr can be approximately obtained from Rw and Rs in this way.

However, in the frequency region where Rs <Rw, A ^{2 that} is not positive becomes negative, and kr ² that is the square of the coupling coefficient kr that should be positive also becomes negative. It cannot be obtained from the effective series resistances Rw and Rs. As is apparent from the equation (4), the actual values of R1 and R2 cannot be obtained in FIG. When Rs = Rw, the coupling coefficient kr becomes zero, and when Rs <Rw, the coupling coefficient kr is mathematically an imaginary number. Although two coils are actually facing each other and the mutual inductance M is M ≠ 0, it is theoretically impossible that the coupling coefficient between the two coils becomes zero or becomes an imaginary number. .

In the frequency region where the condition of Rs> Rw is not satisfied, the values of the effective series resistances R1 and R2 in FIG. 8 are unknown as described above. Furthermore, the effective series resistance Rw (Ω) of the coil increases, and even if the current I is passed through either the primary side or the secondary side coil, R1 × I ² (W), R2 × I ² (W), The power loss due to is excessive, and the coil generates heat. Due to the power loss, the effective power transmission efficiency η decreases. When the same coil is used on both the primary side and the secondary side, when 2 × Rw = Rs (Ω), the coupling coefficient kr is 1, so that Rs is 2 × Rw (Ω), The closer it is to the better.

(Description of combination of coil 1A and coil 1F)
FIG. 17 shows effective power transmission when a load resistance of 10Ω is connected to Rw, Rn, Rs of coil 1A and coil 1F when coil 1A is one coil and coil 1F described later is the other coil. It is a characteristic view which shows the relationship between efficiency and a frequency.

  In FIG. 9, it has been explained that when the coil 1 </ b> A is used for both the power transmission coil 1 and the power reception coil 2, only a predetermined power transmission performance can be achieved. When the coil 1A is used for both one coil and the other coil, the maximum frequency f1 at which the coil 1A satisfies Rs> Rw is about 67 kHz from FIG. That is, f1 of the coil 1A is less than 100 kHz. Therefore, if the coil 1A using a 1 mm formal wire is used for both the power transmission coil 1 and the power reception coil 2, only the same power transmission performance as that of the conventional coil can be achieved.

  The coil 1A shown in FIG. 9 was used as one coil, and the coil 1F shown in FIG. 14 was used as the other coil. Then, in the coil 1A, the highest frequency f1 satisfying at least Rs> Rw increased from 67 kHz to 110 kHz. As a result, the power transmission performance could be improved. Therefore, even if it is the coil 1A of FIG. 9, by selecting the other coil which opposes, electric power transmission performance can be improved with an air core, without using a magnetic material etc.

According to actual measurements, the maximum frequency f1 satisfying the condition of Rs> Rw per coil 1A is about 67 kHz when the opposing coil is the coil 1A, as shown in FIG. 9, and when the opposing coil is the coil 1F. From FIG. 17, when the opposing coil is the coil 1G, it is 150 kHz although not shown. By selecting the other opposing coil, the coil 1A can increase the maximum frequency f1 (Hz) that satisfies the condition of Rs> Rw. When the coil 1A is opposed to the coil 1F, the maximum frequency f1 that satisfies the condition of Rs> Rw for the coil 1F is about 2 MHz. In such a frequency region, the effective series resistance Rw of the coil 1A alone is a high value of 10Ω or more. Therefore, the thermal condition defined by Rw, which will be described later, Rw ≦ (Tw−Ta) / (Ia ² × θi), Thus, the current that can be passed through the coil 1A as the secondary coil can be defined.

  Preferably, when the coil 1A and the coil 1F are used in combination, as described above, the output frequency fa (Hz) of the AC power supply is set to f1 (Hz) in order to transmit power in the frequency region below f1 = 110 kHz. Set to less than. Naturally, at fa (Hz), both the coil 1A and the coil 1F satisfy Rs> Rw. The maximum frequency f1 at which the coil 1A satisfies Rs> Rw is about 67 kHz. However, by using the coil 1A and the coil 1F in combination, even if the coil 1A is used for either the power transmission coil 1 or the power reception coil 2, power can be transmitted at 67 kHz or higher.

  In the embodiment of the present invention, when f1 (Hz) of one coil is low, f1 (Hz) of one coil is set to a predetermined frequency as the other coil, and the margin of about 10% shown in FIG. And select a coil that is higher than 100 kHz. The power transmission device is configured by combining one coil selected in this way and the other coil. With such a configuration, the coil can be used at a high frequency. Then, the power transmission performance of the power transmission device can be improved.

  That is, first, one coil and the other coil are selected. In one coil, each frequency characteristic of Rw, Rs, Rn is measured. Based on the measurement data, one coil obtains the maximum frequency f1 (Hz) that satisfies Rs> Rw. It can be seen from FIG. 17 that the frequency characteristics of the power transmission performance are good in the combination of coils having a high f1 (Hz), compared with FIG. And the output frequency fa (Hz) of AC power supply 30b is set to less than f1 (Hz). In this way, a power transmission device with good power transmission performance can be realized.

(Description of combination of coil 1B to coil 1D)
In each of the coils 1B to 1D using the single conducting wire, the maximum frequency f1 satisfying Rs> Rw exceeds 100 kHz. The coils 1B to 1D are set as one coil, and the other coil is set as any one of the coils 1B to 1D. In one coil, a maximum frequency f1 (Hz) satisfying Rs> Rw is obtained. The output frequency fa (Hz) of the AC power supply 30b included in the power transmission device is set to be less than f1 (Hz). In this way, a power transmission device with good power transmission performance can be realized.

(Explanation when Rs> Rn ≧ Rw is satisfied)
Next, the difference between when Rs> Rn ≧ Rw is satisfied and when it is not satisfied will be described. As described above, in the single coil, this effective series resistance Rw can be obtained accurately by measurement. However, in the transformer configured as shown in FIG. 6, as shown in FIGS. R1 rises from Rw to Rn in the high frequency region just by facing the side coils. R1 is the effective series resistance of the primary coil, but the frequency characteristic of R1 (same as Rw) in FIG. 5 is different from the frequency characteristic of R1 (same as Rn) in FIG. It can be seen from the frequency characteristic chart of Rw and Rn plotted in FIG.

Furthermore, as described above, the coupling coefficient kr can be obtained approximately by obtaining A ² from Rw and Rs and taking the square root of A ² .

  FIG. 13 plots the coupling coefficient kr obtained from Rw and Rs of the coil 1E and FIG. 14 of the coil 1F. In the coil 1E, as shown in FIG. 13, the rate at which Rn (Ω) increases with increasing frequency is low, and Rs> Rn ≧ Rw is satisfied up to about 3.7 MHz. In the coil 1F, as shown in FIG. 14, Rn (Ω) increases rapidly with increasing frequency, and Rs <Rn when the frequency region is 780 kHz or higher.

  Looking at the relationship between the coupling coefficient kr and the frequency between the two coils obtained approximately from Rw and Rs, the coil 1E has a value of the coupling coefficient kr of approximately 0.8 or more up to about 2 MHz. On the other hand, in the coil 1F, it can be seen that the coupling coefficient kr decreases from about 0.9 at 100 kHz as the frequency increases, and decreases to about 0.65 at 1 MHz. Therefore, the highest frequency f2 that satisfies Rs> Rn ≧ Rw is preferably as high as possible.

  By using the coil in the frequency region satisfying the condition of Rs> Rn ≧ Rw, both the coil alone of FIG. 5 and the transformer configured as shown in FIG. 6 are theoretically ideal characteristics. Therefore, the power transmission performance can be improved as compared with the conventional case.

(Explanation when Rs> Rn ≧ Rw is not satisfied)
However, depending on the frequency domain, Rn = Rw is not satisfied and Rn> Rw, and is affected by Rn. Therefore, the values of R1 and R2 cannot be accurately obtained in FIG. R1 and R2 vary depending on the value of RL shown in FIG. That is, R1 and R2 fluctuate due to currents flowing through R1 and R2, and naturally fluctuate depending on the frequency. Therefore, in FIG. 8, the actual accurate values of R1 and R2 at the time of power transmission cannot be obtained.

  In this embodiment, the measurement of whether two conditions of Rs> Rw and Rs> Rn ≧ Rw are satisfied describes the case where the same coil is opposed. However, as shown in FIG. 17, two arbitrary coils having different structures, configurations, outer diameters, and the like may be opposed to each other, and measurement may be performed using either the primary side coil or the secondary side coil. It is not necessary to measure it facing each other.

  Further, detailed operational effects regarding the relationship of Rs> Rn ≧ Rw will be described later with reference to the coils 1F and 1G.

(Description of thermal resistance θi (° C / W), temperature Tw (° C), ambient temperature Ta (° C))
Next, the relationship of Rw ≦ (Tw−Ta) / (Ia ² × θi) will be described. As described above, in FIG. 8, the values of the effective series resistances R1 and R2 of the coil when the power is actually transmitted to the load resistance RL are unknown, and in FIG. R1> Rw. In other words, the thermal condition of the coil cannot be defined except that Rw is used as a reference at the minimum. Therefore, it is necessary to define the thermal conditions of the coil based on Rw as a minimum.

  In implementing this invention, the thermal resistance θi (° C./W) of the coil is determined by the coil structure and installation conditions. For example, when the coil is a single air core, θi is high, and when the coil is fixed in a resin with low thermal resistance and installed in water, θi is low. The temperature Tw (° C.) at which the coil can operate is determined by the structure and application of the coil, and is 50, for example, in a case where it is incorporated in a case with good heat insulation or in a device such as a transformer. In a case where it is installed at a place where a human body, an animal, or the like touches, for example, about 80 ° C. to 80 ° C., the temperature is about 40 ° C. The ambient temperature Ta (° C.) of the place where the coil is installed is, for example, −20 ° C. to 40 ° C. outdoors, for example, 15 ° C. to 30 ° C. indoors, and 40 ° C. to 50 ° C. Become.

  Normally, the higher the temperature, the more heat is dissipated to the surroundings, so it is necessary to solve the heat diffusion equation accurately. However, since it is difficult to solve the thermal diffusion equation in consideration of thermal constants such as specific heat for coils having various structures, the thermal resistance θi (° C./W) is simply obtained by the following method.

  First, the coil temperature T1 (° C.) in the initial state is obtained at the place where the primary side or secondary side coil is installed. A DC constant current Id (A) is passed through the coil, the voltage Vd (V) across the coil is measured, and the power consumption of the coil is determined as Pd = Vd × Id (W). When the temperature of the metal conductor rises, the resistance value increases and the voltage Vd across the coil rises. Therefore, Vd is recorded with a pen recorder or the like to obtain an average value, or the Vd is monitored successively with an A / D converter or the like. It is desirable to take an average value. When the thermal equilibrium is reached, the coil temperature T2 (° C.) is measured. The thermal resistance θi (° C./W) is obtained as θi = (T2−T1) / Pd (° C./W). This measurement is preferably performed as an average value by measuring several times while changing the current value of Id.

The thus obtained thermal resistance θi (° C./W) is determined by an effective series resistance Rw (determined by an effective series resistance Rw (Ω) of the coil under actual use conditions and a current Ia (A) flowing through the coil. When the power consumed by Ω), Rw × Ia ² (W), is multiplied, the temperature rise value of the coil under actual use conditions, Tr (° C.), is obtained. Tr = θi × Rw × Ia ² (° C.) If the temperature at which the coil can be operated is Tw (° C.) and the ambient temperature of the place where the coil is installed is Ta (° C.), Tr = Tw−Ta. If (Tw−Ta) ≧ θi × Rw × Ia ² (° C.) is not satisfied, the usable temperature of the coil will be exceeded, making it difficult to implement the present invention.

The condition relating to the effective series resistance Rw (Ω), Rw ≦ (Tw−Ta) / (Ia ² × θi), transforms the inequality and defines the condition of Rw (Ω) or Ia (A). At the frequency at which power is transmitted, the effective series resistance Rw (Ω) is a variable obtained by actually measuring the primary side or secondary side coil alone, and the current Ia (A) flowing through the primary side or secondary side coil is also calculated. It is obtained by actual measurement or is determined by the power supply condition on the primary side and is determined by the load condition on the secondary side, and other Tw (° C), Ta (° C), and θi (° C / W) are known. Constant. Therefore, if Rw (Ω) is obtained, the upper limit value of Ia (A) is defined. Conversely, if Ia (A) is determined, the upper limit value of Rw (Ω) is defined.

  Rw (Ω) is the sum of DC resistance Rd (Ω) and AC resistance Ra (Ω), and Rd and Rw can be directly measured, so by determining Ia (A), it increases with the number of turns. The upper limit value of the effective series resistance Rw (Ω), which is the sum of Rd and Ra, can be defined, and the frequency range in which power can be transmitted can be defined from the relationship between the effective series resistance Rw (Ω) and the frequency.

  Both 1V × 10A and 10V × 1A have the same power of 10 W, but the power loss due to the effective series resistance of the coil is 100 times that of 1A in the case of 10A. Power transmission performance between two coils if the power loss due to the effective series resistance of the coil is not specified considering the current Ia (A) flowing through the coil, regardless of the primary side or the secondary side. Cannot be improved.

(Explanation of the relationship of Rs> Rn ≧ Rw)
Here, with reference to the coil 1F and the coil 1G, the detailed effect regarding the relationship of Rs>Rn> = Rw is demonstrated. The Litz wire is considered to have an equivalent circuit as shown in FIG. 18 in which the self-inductances of the respective strands constituting the Litz wire are connected in parallel and the mutual inductance is provided between the respective strands. Even if a litz wire is wound with a flat single-layer spiral, the frequency characteristics of the effective series resistance Rw (Ω) of the coil itself will not be improved, and the self-inductance of the coil itself will decrease. It is considered that the self-inductance when the wire is formed as a coil changes due to the mutual inductance between the formal wires and between the conducting wires. That is, the characteristics when formed as a coil vary depending on the twisting method, twisting pitch, winding method (close winding, loose winding, multilayer winding), number of turns, outer shape, and the like.

  The coil 1F shown in FIG. 14 and the conductor used in the coil 1G shown in FIG. 15 are the same, the conductor outer diameter is 0.05 mm, the insulation coating thickness is 5 μm, and the conductor outer diameter is 0.1 mm. A coil 1F is tightly wound 30 turns around an outer shape 70 mm, and the coil 1G is tightly wound 20 times around an outer diameter 50 mm.

  When the frequency characteristics of Rw, Rn, and Rs of the coil 1F and the coil 1G are compared in FIG. 14 and FIG. 15, the frequency region in which Rs <Rn exists in the coil 1F exists at 780 kHz or more. Then, the condition of Rs> Rn ≧ Rw is satisfied up to about 2.1 MHz. It cannot be determined whether this cause is related to the twisting method, the twisting pitch, or the number of turns, the outer diameter, and the winding method. However, if at least the frequency characteristics of Rw, Rn, and Rs of the coil are measured, it can be determined whether or not the coil is suitable for the power transmission device. The specific method is described below.

(Description of inductance and coupling factor)
Table 1 shows the inductance Ls when the single inductance Lw (μH) of the coil 1B, the coil 1F, and the coil 1G and the same short-circuited coil face each other at a distance of zero at frequencies of 5.0 kHz to 1.0 MHz. The value of (μH) and the coupling coefficient ki approximately obtained by the calculation method shown below are described. Each ki in this table is a ki plotted in FIG. 10, FIG. 14, and FIG.

First, a method for approximately obtaining the coupling coefficient ki from the coil inductance change will be described. As described above, when the self-inductance of the coil in FIG. 5 is Lw (H) and the inductance of the primary coil in FIG. 6 is Ln (H), L1 = Lw in FIGS. = Ln (H). Further, as shown in FIG. 7, when the primary side inductance when the secondary side coil facing the primary side coil is short-circuited is Ls (H), Ls = (L1−A ² L2 ) The relationship of H holds. Unlike the effective series resistances Rw (Ω) and Rn (Ω), L1 = Lw = Ln (H) in actual measurement. L1, L2, and A ² are as described above.

When the same coil is used for the primary side and the secondary side, since L1 = L2 = Lw and R1 = R2 = Rw, the relationship of Ls = (Lw−A ² Lw) is established. As described above, since the value of ω ² L2 ² / R2 ² = ω ² Lw ² / Rw ² is 50 or more at 10 kHz to 30 kHz or more, it can be regarded that A ² ≈ki ² . Therefore, the coupling coefficient ki is approximately obtained as ki ² = (Lw−Ls) / Lw and ki = √ ((Lw−Ls) / Lw). As described above, the coupling coefficient obtained from the inductance change, Lw, and Ls in this way is expressed as ki. Comparing kr and ki plotted in FIG. 14 and FIG. 15, it can be seen that kr and ki are almost the same in FIG.

  However, in FIG. 14, kr and ki do not match. Furthermore, in the coil 1B, kr and ki are plotted in FIG. 10, but in FIG. 10, it can be seen that kr sharply decreases at a frequency where Rn> Rs. Actually, when two coils 1G shown in FIG. 15 are used, Rs> Rn ≧ Rw is satisfied up to 2.1 MHz, and Rs> Rw is satisfied up to 10 MHz or higher. It can transmit power with high power factor and high effective power efficiency, and power transmission performance is very good.

  That is, the higher the maximum frequency f2 (Hz) that satisfies Rs> Rn ≧ Rw, and the higher the frequency, the closer the value of Rn / Rw is to 1, the better the coil performance, and the higher the frequency, Rw (Ω) There is little increase. As described above, the frequency characteristic of the coupling coefficient kr obtained from Rw and Rs is compared with the frequency characteristic of the coupling coefficient ki obtained from Lw and Ls by observing the relationship between the frequency and Rw, Rn, and Rs. As a result, it is possible to predict the performance as a transformer, which is a power transmission means with the coils facing each other, which cannot be determined only by the frequency characteristics of the effective series resistance of the single coil.

  Therefore, the proper twisting method, twisting pitch, and winding method of the Litz wire constituting the coil are to form a plurality of coils and measure the frequency characteristics of the coils Rw, Rn, Rs, and preferably the frequency of Lw, Ls By measuring the characteristics and comparing the frequency characteristics of kr and ki, it is possible to find the optimum coil. This technique is applicable not only to litz wires but also to copper wires, vinyl wires, and other electric wires of other embodiments described later, and a coil suitable for power transmission can be selected. In other words, by changing the wire material, wire diameter, dimensions, shape, winding method, etc., it is possible to determine the performance as a transformer that is a power transmission means facing the coil, which cannot be determined only by the frequency characteristics of the effective series resistance of the coil alone. Therefore, it is possible to provide a coil with good power transmission performance that could not be realized by the conventional technology.

  For example, a coil 1E wound with a 1 mm single conductor and provided with a gap has a maximum frequency f2 satisfying Rs> Rn ≧ Rw of 3.7 MHz and a maximum frequency f1 satisfying Rs> Rw. , 7.7 MHz, there is not much difference in the definition of Rs> Rn ≧ Rw compared to the coil 1G. However, the Rw of the coil 1E alone at 4 MHz is 0.87Ω, the Rw of the coil 1G alone is about 2Ω, the Rw of the coil 1E alone at 2.9MHz is 2.9Ω, and the Rw of the coil 1G alone is 17Ω. Thus, the coil 1E has better high-frequency characteristics of the effective series resistance Rw (Ω) of the coil alone than the coil 1G.

Therefore, according to the definition of Rw ≦ (Tw−Ta) / (Ia ² × θi), the coil 1E formed with a single conductor can be used at a higher frequency than the coil 1G formed with a litz wire. Thus, in the embodiment of the present invention, the highest frequency f1 (Hz) satisfying Rs> Rw, the highest frequency f2 (Hz) satisfying Rs> Rn ≧ Rw, and Rw ≦ (Tw−Ta) / ( Ia ² × θi), by realizing a coil that cannot be realized by the conventional technology, and by specifying an optimum frequency region for using the coil, power transmission performance compared to the conventional technology The present invention has an excellent effect that a good power transmission device can be realized.

  In the prior art including the above cited reference, only a specific configuration of the coil is defined. Then, by showing only one embodiment of a specific configuration, it is claimed that the characteristics of interest, for example, power transmission performance can be improved. However, as described above, even if the outer diameter and inner diameter are the same, the coil characteristics are completely different depending on the wire diameter and the number of turns. Even if the same conducting wire is used, if the configuration (outer diameter, number of turns, etc.) is different, the coil characteristics will be different. That is, even if a specific configuration such as a wire rod and a winding method is defined, coils actually produced have various configurations, and it is not guaranteed at all that they have the same effect.

  Therefore, it is impossible to realize a coil that satisfies the requirements as a coil of the power transmission device only by defining a specific configuration of the coil. Actually, a power transmission apparatus capable of transmitting 20 W of power with an effective power transmission efficiency of 80% as described in the prior art has not been implemented to date.

  As in the present application, a coil having good power transmission performance and a power transmission device having good power transmission performance are defined as long as the characteristics change when the configuration other than the specific configuration of the coil is changed, and the operating conditions of the coil are not defined. Cannot be realized. On the other hand, the embodiment of the present invention can realize a power transmission device with good power transmission performance by defining operating conditions of each coil in coils having various configurations capable of inductive coupling. As described above, the embodiment of the present invention exhibits extremely excellent effects that could not be realized by the conventional technology.

(Explanation of coil power factor)
In each embodiment of the present invention, a coil that is not equipped with a magnetic material transmits a large amount of power, which was conventionally difficult, between two coils in a loosely coupled state with a coupling coefficient of about 0.9 or less. The coil which can be realized is realized. As described above, although the power factor is 0.5 or more, the reactive power input to the primary coil may exceed the effective power in the loosely coupled state.

When the power factor decreases from 1 to 0.5, the current flowing in the primary coil due to the apparent power is doubled, and the power loss due to the effective series resistance Rw (Ω) of the primary coil is doubled. . In addition, when a current flows through the load resistance connected to the secondary coil, the magnetic flux generated by the current flowing through the secondary coil passes through the conducting wire forming the primary coil, generating eddy current loss and causing the primary The side coil generates heat. Therefore, the above inequality, Rw ≦ (Tw−Ta) / (Ia ² × θi), is preferably satisfied to implement the embodiment of the present invention, otherwise the implementation of the present invention is difficult. become.

If the frequency fd (Hz) at which one coil is driven satisfies Rs> Rn ≧ Rw, the value of the internal resistance R3 of the power source shown in FIG. 117 is equal to or less than Rw (Ω). Since the secondary side coil viewed from the load resistance RL can be regarded as the primary side being short-circuited, R2 (Ω) is substantially equal to Rs (Ω). Therefore, in the secondary side coil, at the frequency fj (Hz) for receiving power,
It is more preferable if the relationship of Rs ≦ (Tw−Ta) / (Ia ² × θi) is satisfied. In addition, in FIG. 117, the value of R1 is unknown, but also in the primary side coil,
It is more preferable if the relationship of Rs ≦ (Tw−Ta) / (Ia ² × θi) is satisfied.

However, in a general transformer, the relationship between the flux linkage Φc, the leakage flux Φg, and the coupling coefficient k is k ² = Φc / (Φc + Φg), 1-k ² = Φg / (Φc + Φg), which is known. As shown, the flux linkage Φc transmits effective power. As is known, the leakage magnetic flux Φg provides reactive power that is the product of the voltage V applied to the reactive element and the flowing current I.

  In the coil, since the phase of I is 90 degrees behind the phase of V, multiplying the instantaneous value of V by the instantaneous value of I and integrating for one period makes the power zero, so that it is a reactive element. The coil does not consume power. In this field, there are many documents that specify that the magnetic flux leakage causes energy loss and specify the coil shape to increase the flux linkage ratio, but as mentioned above, the magnetic flux leakage does not consume power. .

  Therefore, if the effective series resistance Rw (Ω) is small enough to be ignored, large power can be transmitted regardless of the ratio of leakage magnetic flux. However, although the effective series resistance Rw (Ω) is small in the coil having the configuration disclosed in Patent Document 1, the power factor is remarkably small because the self-inductance and coupling coefficient of the coil are small. For this reason, a large apparent power must be supplied to the primary coil. To realize a coil suitable for power transmission, the coil configuration is determined, all parameters are set appropriately, and the effective series resistance is set. Rw (Ω) must be made as small as possible.

(Description of frequencies that can be used for power transmission)
The upper limit of the frequency at which the coil of the embodiment of the present invention can be used for power transmission is the highest frequency that satisfies fs (Hz) that satisfies Rs> Rw and Rs> Rn ≧ Rw. The lower limit of the frequency at which the coil can be used for power transmission can be determined by the regulation of f2 (Hz). The phase difference between the voltage V applied to the coil alone and the current I flowing through the coil alone is 80 degrees. It is obtained by defining the above.

  Although not shown, in the coil 1B having a low maximum frequency f1 (Hz) that satisfies Rs> Rw, the phase difference between V and I is 80 degrees or more up to less than 5 kHz, but Rs> Rw In the coil 1G having a maximum frequency f1 that satisfies 10 MHz exceeding 10 MHz, the phase difference between V and I is 80 degrees or less when it is less than 20 kHz.

  As described above, referring to FIG. 10, the highest frequency f1 at which the coil 1B satisfies Rs> Rw is about 210 kHz, and the maximum frequency f2 at which Rs> Rn ≧ Rw is about 75 kHz. The frequency range in which the coil 1B according to the definition of the maximum frequency f1 (Hz) satisfying Rs> Rw can be used is 5 to 210 kHz, and the coil according to the specification of the maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw. The frequency region where 1B can be used is 5 to 75 kHz. In this manner, the coil according to the embodiment of the present invention can be used in a frequency region that is close to a theoretical ideal characteristic. In FIG. 11, the phase angle θ of the coil 1C is plotted. In the coil 1C having a higher f1 (Hz) than the f1 (Hz) of the coil 1B, the frequency at which the phase angle θ is 80 degrees is about 8 kHz, which is slightly higher than 5 kHz.

  As described above, according to this embodiment, the necessary self-inductance and coupling coefficient k can be ensured by defining the wire diameter of the conducting wire 11 of the coil 1a, the outer diameter of the coil, and the number of turns. In addition, the upper limit of the current value Ia of the coil 1a or the upper limit of the number of turns that determines the effective series resistance Rw (Ω) of the coil 1a can be defined, and the ratio of the reactance X to the pure resistance R when a load resistance is connected, X / By using the coil 1a in the vicinity of the frequency where R and the phase difference φ between the AC voltage applied to the coil and the AC current flowing through the coil are minimum, the power factor cos φ is maximum, and the effective series resistance Rw (Ω) is small. Reactive power and apparent power during power transmission can be reduced. Furthermore, the effective power efficiency can be increased to, for example, 85% or more.

(Description of comparison of frequency characteristics of coils 1A and 1E)
19 compares the frequency characteristics of the effective series resistance Rw (Ω) of the closely wound coil 1A shown in FIG. 9 and the effective series resistance Rw (Ω) of the loosely wound coil 1E shown in FIG. FIG. As shown in FIG. 19, when the frequency is increased, the loosely wound coil 1E can suppress an increase in the effective series resistance Rw (Ω) of the coil compared to the closely wound coil 1A. Further, in the case of a coil having the same outer diameter, the total extension of the winding is shortened, so that the direct current resistance can be kept low.

(Description of an example in which the frequency characteristics of the effective series resistance change depending on the width of the air gap)
FIG. 20 is a diagram showing how the frequency characteristic of the effective series resistance of the coil changes depending on the width of the gap when a 0.4 mm formal wire is wound for 25 turns. The width of the gap is set to 0 mm, 0.2 mm, and 0.4 mm, but it can be seen that a wider gap can suppress an increase in effective series resistance with an increase in frequency. Since the number of turns is the same, the outer diameter of the coil increases as the width of the gap increases, and the total length of the copper wire that constitutes the coil increases. However, the effective series resistance is low.

  However, since the eddy current loss is proportional to the volume of the conductor through which the magnetic flux penetrates, if the maximum diameter of the single conductor is not 0.2 mm or more, even if the gap t (mm) is provided between the conductors, The increase rate of the effective series resistance Rw does not decrease so much. In view of the relationship between the frequency of a single coil in which a single conducting wire having a wire diameter of 0.2 mm is closely wound and the effective series resistance Rw in FIG. 15, the increase rate of the effective series resistance due to the increase in the frequency is 0.2 mm. It can be seen that the frequency characteristic of the effective series resistance Rw cannot be improved much even with a single conductor having a wire diameter of 0.2 mm, even if a gap is provided.

  The self-inductance of the coil 1D shown in FIG. 12 is about 19 μH. The self-inductance of the coil in which the coil 1D is wound in two layers is about 76 μH, and a result almost equivalent to the theory that the self-inductance is proportional to the square of the number of turns is obtained. The frequency characteristic of the effective series resistance of the coil wound in two layers is worse than that in the single layer winding, and the maximum frequency f1 (Hz) that satisfies Rs> Rw is also low. However, in a low frequency region where the effective series resistance is low, reactance can be ensured, so it may be advantageous to use a two-layer winding and use at a low frequency.

  A coil in which the coil 1D is wound in two layers is used for one coil and the other coil. The maximum frequency f1 in which the coil 1D is wound in two layers satisfies Rs> Rw is 550 kHz, and the coil in which the coil 1D is wound in two layers has a high inductance. However, the required reactance can be secured.

In FIG. 16, the effective series resistance Rw at 5 kHz when a single conducting wire having a wire diameter of 0.2 mm is closely wound is 0.83Ω. The effective series resistance at 1 MHz is 2.16Ω, and the increase rate of the effective series resistance Rw is 2.16 / 0.83 = 2.60. The increase rate of the coil 1E provided and wound is smaller than 7.6. However, in a coil having a wire diameter of 0.2 mm, the absolute value of Rw (Ω) increases and the thermal resistance θi (° C./W) decreases, so that Rw ≦ (Tw−Ta) / (Ia ² × θi) In order to satisfy the relationship, it is necessary to select a wire diameter suitable for the power value to be transmitted.

(Explanation of coil turns and inductance)
Next, a description will be given of a coil having 8 turns and a minimum inductance value of 2 μH. The coil of the conventional example has 5 turns, and at 1 MHz, the coil Lw is 0.79 μH, Ls is 0.45 μH, Lw, and the coupling coefficient ki approximately calculated from Lw and Ls is 0.66. Power transmission performance is also very bad. A coil wound eight times in the same shape using the same wire as the coil has an Lw of about 2.1 μH, an Ls of about 0.7 μH, and an approximately calculated coupling coefficient ki of about 0.83. It has become.

  As described above, the coil in which the conductive wire of the conventional example is wound eight times is actually a high frequency that has an effective series resistance that is too low, a poor frequency characteristic of Rw (Ω), and a sufficient reactance. In the region, Rs> Rw is not satisfied. For this purpose, it is necessary to select an appropriate twisting and winding method for the conducting wire, but since the minimum inductance and coupling coefficient used in the high frequency region can be secured, a minimum number of windings of 8 is specified from the above measurement results. In addition, 2 μH is defined as the minimum value of inductance.

  And as mentioned above, the diameter of the conducting wire of the coil of the conventional example is 1.5 mm, and the conducting wire is further wound 3 times on the outermost peripheral part of the coil wound 5 times, and the number of turns is 8 times. The outer diameter is 3 times × 2 × 1.5 mm + 30 mm = 39 mm. Therefore, in a coil configured using the conventional coil conductor, in order to ensure the minimum inductance value of 2 μH and the coupling coefficient, the ratio of the coil outer diameter D to the wire diameter d3 is 39 / 1.5 = 26. The coil outer diameter D is required to be at least 25 times the wire diameter d3.

  However, as described above, the specific configuration that “D is required to be at least 25 times d3” can secure a minimum inductance value of 2 μH and a coupling coefficient by changing another configuration factor such as a wire and the number of turns. It can be lost. For example, it is conceivable that the diameter of the wire is reduced to provide a gap between the wires. Therefore, in order to secure the minimum value of 2 μH of inductance, there is a possibility that the number of turns of 8 times or more is required. The wire used and the number of turns are selected so as to ensure the minimum inductance value of 2 μH, and finally the frequency characteristics of Rw, Rs, and Rn are measured in the coil whose configuration is uniquely specified. It goes without saying that a coil whose configuration is uniquely specified means one actually created as a coil. Therefore, the frequency fa (Hz) of the AC power source, which is the aforementioned operating condition of the coil, derived from the characteristics obtained by measuring what is actually created as a coil is defined.

  To reiterate, a coil has a virtually infinite configuration by changing other components, for example, simply by defining a specific configuration. It has not been proved that a coil having a specific configuration exhibits an effect of always exhibiting an excellent power transmission performance as compared with an invention based on other specific configuration rules. It is virtually impossible to prove.

  Only by the embodiment of the present invention, the configuration is defined so as to ensure the above-described minimum inductance value of 2 μH and the coupling coefficient, and a coil suitable for power transmission can be selected from the coils satisfying those characteristic conditions. become able to. As described above, the embodiment of the present invention shows measured characteristic data in various embodiments, unlike the conventional coil. A coil having a configuration capable of inductive coupling has a variation that cannot be specified. Therefore, it is impossible to ensure power transmission performance in a coil having an arbitrary configuration. Further, in the conventional technique, it cannot be determined that the coil whose configuration is uniquely specified can ensure the power transmission performance.

  A power transmission device with good performance using the coils of the power transmission device having various configurations only by defining the operating condition based on the characteristic definition which is the gist of the embodiment of the present invention for the coil selected by the above-described method. Can be realized. This extremely excellent effect could not be realized with a conventional coil that defines only a specific configuration of the coil.

  Moreover, it is preferable that the maximum frequency f1 with which one coil satisfies Rs> Rw is 500 kHz or more. The same coil is used, and a coil having a high maximum frequency f1 (Hz) satisfying Rs> Rw is used at a frequency at which reactance can be secured. For example, it has been confirmed that power transmission performance can be ensured by driving at less than 250 kHz. Alternatively, it is more preferable that the maximum frequency f2 at which one coil satisfies Rs> Rn ≧ Rw is 500 kHz or more.

(Explanation regarding load resistance and power factor)
21 and 22 are diagrams showing the relationship between the power factor and the frequency when the load resistance RL is varied. 9 also illustrates the relationship between the effective power transmission efficiency η and the frequency when the coil 1A is used for both the power transmitting coil 1 and the power receiving coil 2. FIG. 17 also shows the relationship between the effective power transmission efficiency η and the frequency when the coil 1A is used for the power transmission coil 1 and the coil 1F is used for the power reception coil 2. Both are frequency characteristics when the load resistance RL = 10Ω. The power factor is calculated from cos φ by measuring the impedance on the primary side to obtain the phase angle φ. cos 60 degrees = 0.5. In the frequency region where φ <60 degrees, the power factor is 50% or more.

  As can be seen from FIGS. 21 and 22, when the load resistance value is low, the frequency at which the power factor is highest is low. When the load resistance is high, the frequency at which the power factor is highest is high. Further, when the load resistance value is low, the maximum value of the power factor is large. When the load resistance value is high, the maximum value of the power factor is small. Below 5Ω, which is the minimum load resistance value that is generally used, the frequency at which the power factor is highest is less than f2 (Hz) of one coil.

  In FIG. 21, when two coils 1A are used, the load resistance value satisfying a power factor of 50% or more is 10Ω or less at the maximum frequency f1 = 67 kHz that satisfies Rs> Rw. In FIG. 22, when the coil 1A is used for the power transmission coil 1 and the coil 1F is used for the power reception coil 2, the load resistance value satisfying a power factor of 50% or more at the maximum frequency f1 = 110 kHz satisfying Rs> Rw is Supports up to 50Ω. Further, as can be seen from a comparison between FIG. 21 and FIG. 22, in FIG. 22, the maximum value of the power factor increases as f1 (Hz) increases.

  Comparing the relationship between the power transmission efficiency η and the frequency in FIG. 17 with FIG. 9, it can be seen that f1 (Hz) increases and the power transmission performance is improved. In both FIG. 9 and FIG. 17, when the frequency is equal to or higher than f1 (Hz), the power transmission efficiency η is extremely deteriorated. Therefore, it can be seen that the power transmission performance η can be improved by making the coil 1F face the coil 1A.

  In the power transmission device in which an example of the specific configuration of the coil of the conventional example is described, only an example at a specific frequency of 100 kHz is described. And it is specified that the frequency is not limited to 100 kHz. However, as described above, the power factor and the effective series resistance Rw (Ω) of the coil change depending on the frequency. If the frequency is not set at the maximum power factor at the minimum value Rm (Ω) of the load resistance RL, power loss due to the effective series resistance Rw (Ω) occurs due to reactive power. As described above, the frequency characteristics of Rw, Rs, and Rn are measured, and f1 (Hz) and f2 (Hz) are obtained. The frequency fφ (Hz) at which the power factor is maximized is preferably smaller than f1 (Hz). However, as the resistance value of the load RL increases, the ratio of the effective series resistance Rw (Ω) to RL (Ω) of the coil, Rw / RL, decreases. Therefore, the power loss due to Rw (Ω) is relatively small compared to the power consumed by the load. Therefore, even when the load resistance value is large, the power factor is small, but power transmission can be performed at a frequency less than f1 (Hz).

(Explanation of frequency characteristics of effective power transmission efficiency η)
The frequency characteristics of the effective power transmission efficiency η in FIGS. 9 and 17 will be described. A non-inductive load resistance of 10Ω is connected to the power receiving coil 2 and the impedance is measured on the power transmission side. The phase angle φ is obtained on the power transmission side by impedance measurement, and the power factor cos φ at each frequency is calculated. The voltage V (V) applied to the power transmission coil 1 is set so that a constant current Ia of 0.2 A flows through the power transmission coil 1. The effective power Pr on the power transmission side is obtained as Pr = cos φ × V × Ia (W). The secondary effective power Ps (W) determines the effective value Ve of the voltage across the non-inductive load resistance ^{10Ω, Ps = Ve 2/10} (W), obtained as. The effective power transmission efficiency η at each frequency is obtained as η = Ps / Pr. This measurement method is different from the conventional example that does not take into consideration that the power factor varies depending on the load resistance value and frequency.

  Find the load resistance value from the power required by the actual electrical equipment. Since the power required by the electrical equipment is at the lower limit of voltage Vs = 5V, current Is = 0.5A, and power 2.5W, the minimum value of the load resistance value RL is about 10Ω. In an electric device that requires electric power of 10 W or more, the voltage Vs is increased and the current Is is decreased. Even if the actual circuit voltage is about 5V, a step-down PWM step-down converter is often used. For example, a personal computer or the like that requires about 30 W of power uses a 15V, 2A power source. In this case, the minimum value of the load resistance value RL is about 15/2 = 7.5Ω. Further, when the voltage Vs is increased and the current Is is decreased to about 30 V and 1 A, the minimum value of the load resistance value RL is about 30/1 = 30Ω. As a rough guide, the minimum value of the load resistance RL is about 2 to 50Ω. Therefore, in order to suppress the power loss due to the effective series resistance of the coil to about 20% or less of the received power, assuming that the minimum value of the load RL is Rm (Ω), the effective series resistance Rw of the receiving coil 2 is Rw × 5. ≦ Rm (Ω) must be satisfied. That is, it is desirable that Rw of the power receiving coil 2 is 0.4 to 10Ω or less at the output frequency fa (Hz) of the AC power supply.

  According to actual measurement, the resistance component on the power transmission coil 1 side is usually equal to or less than the load resistance value RL in the above-described embodiment, although it depends on the frequency. Therefore, when the minimum value of the load RL is Rm (Ω), it is desirable that the effective series resistance Rw of the power transmission coil 1 and the power reception coil 2 is 0.4 to 10Ω or less.

  When the upper limit of the effective series resistance Rw (Ω) is determined, Rs (Ω) and Rn (Ω) are obtained by actual measurement. At f1 (Hz), the effective series resistance Rw is desirably 0.4 to 10Ω or less. Accordingly, it is desirable that both Rs and Rn be 10Ω or less at the frequency at which the coil is actually used.

As shown in FIG. 2 and described above, when power is actually transmitted, a loss due to eddy current loss due to the magnetic flux generated by the current flowing in the power transmitting coil 1 and the power receiving coil 2 passing through the other coil occurs. Loss increases. As described above, when power is actually transmitted, the values of R1 and R2 in FIG. 8 are unknown. Therefore, the actual effective series resistance value Rw (Ω) described above is determined by the use condition of the power receiving side device in the same manner as the definition of Rw ≦ (Tw−Ta) / (Ia ² × θi). Is.

(Description of change in inductance of power transmission coil 1)
Further, as described above, the power transmission device, the power transmission device of the power transmission device, and the power reception device of the power transmission device in the embodiment of the present invention have extremely good power transmission performance. Although depending on the current flowing through the power transmission coil 1 and the power reception coil 2, by using the coil 1G having a diameter of 5 cm described above for both the power transmission coil 1 and the power reception coil 2, it is possible to transmit power of up to about 40 W. . When such power transmission performance is achieved, the power transmission coil 1 has a metal heating action, similar to the overheating coil of the induction heater. Therefore, as described in Patent Document 5, when a metal body such as a clip approaches the power transmission coil 1, heat generation of the metal body must be prevented. Alternatively, it is necessary to determine a proper power receiving unit. Therefore, next, a two-terminal network circuit of the power transmission coil 1 will be considered.

(About the coil configuration to prevent unwanted radiation)
FIG. 23 is a diagram illustrating the structure of the coil 1 f that prevents unwanted radiation.

  In FIG. 23, at least one magnetic material plate 51 is provided on one side facing the coil 50. Thus, by providing at least one magnetic material plate 51 on one surface side of the coil 50, the coil 1f that prevents unnecessary radiation can be realized. In the conventional example, it is described that a coil having a configuration equivalent to that of the coil 1f has an effect of improving power transmission performance. However, the coil of the power transmission device of various configurations equipped with the magnetic material plate in the embodiment of the present invention to be described later improves the power transmission performance and reduces unnecessary radiation as compared with the conventional example. It has the effect of preventing fluctuations in coil characteristics when a metal body is close to the surface opposite to the coil equipment surface. Furthermore, the coils of the power transmission devices having various configurations according to the embodiments of the present invention to be described later have the same effects even when other structural factors such as the type of the conductive wire and the outer diameter change.

As described above, in the coil 1f, the coil 50 is short-circuited with the effective series resistance of one coil alone in the air-core state as Rwa (Ω) and the other coil facing the one coil in the air-core state. When the effective series resistance of one coil is Rsa (Ω), 100 kHz
Thus, one coil satisfying Rsa> Rwa is used. For example, as the coil 50, the coils 1B to 1G which are the embodiments of the present invention described above are used.

  Furthermore, when the coil 1t is formed from the coil 1f equipped with the magnetic material plate 51 or 511, 512 and the metal plate 55 shown in FIGS. 23 to 34 as one coil, the coil 1f to the coil 1t are converted to Rsa> at 100 kHz. It is preferable that Rwa is satisfied.

  Furthermore, when the coil 1f equipped with the magnetic material plate 51 is one coil, the coil 1f preferably satisfies Rsa> Rwa at 100 kHz.

  Assuming that the maximum frequency satisfying Rsa> Rwa of the coil 1f is f1 (Hz), the power transmission device 100 equipped with the coil 1f transmits power at a frequency less than f1 (Hz).

  When the coil 1f is the power transmission coil 1 of the power transmission device 100, the coil 1f is driven at a frequency fa (Hz) less than f1 (Hz) by the AC power supply 30b shown in FIG.

  When the power transmission unit 30 of the power transmission device 100 includes the coil 1f, the power transmission unit 30 including the coil 1f is a power transmission device of the power transmission device of the present invention.

  When the power receiving unit of the power transmission device 100 includes 40 and the coil 1f, the power receiving unit 40 including the coil 1f is a power receiving device of the power transmission device of the present invention.

(Other coil configurations that prevent unwanted radiation)
FIG. 24 is a diagram showing a coil 1 g in which an insulating plate 52 is provided between the coil 50 and the magnetic material plate 51. In FIG. 24, the insulating plate 52 can suppress an increase in the effective series resistance Rwa (Ω) of the coil 1g when the frequency is increased. Further, it is possible to prevent the Q of the coil 1g from being lowered when the frequency is increased.

  FIG. 25 is a diagram showing a coil 1 h in which two layers of a magnetic material plate 511 and a magnetic material plate 512 are provided on one side of the coil 50. In FIG. 25, the coil 1h is provided with the magnetic material plate 511 and the magnetic material plate 512, so that the coil inductance can be increased and the coil Q can be increased compared to the configuration of the coil 1f.

  FIG. 26 is a diagram showing a coil 1j in which an insulating plate 52 having a thickness of I (mm) is provided between two magnetic material plates 511 and 512 constituting the coil 1h shown in FIG. In FIG. 26, the insulating plate 52 can suppress an increase in the effective series resistance Rwa (Ω) of the coil 1j when the frequency is increased. Further, it is possible to prevent the Q of the coil 1j from being lowered when the frequency is increased. Such a configuration increases the thickness of the coil but is suitable for use in a power transmission coil. In particular, the power transmission coil converts electrical energy into magnetic energy, and it is preferable to use a power transmission coil to eliminate the cause of unnecessary radiation. The thickness I (mm) of the insulating plate 52 is preferably at least half of the thickness of the magnetic material plate 511 or 512. The effect of the insulating plate 52 is as described above.

(Regarding the coil configuration that prevents fluctuations in coil characteristics when a metal object is in close proximity)
FIG. 27 is a diagram illustrating a structure of a coil 1k that prevents fluctuations in coil characteristics when a metal body is close to the coil. In FIG. 27, a metal plate 55 has a thickness M (mm), and an insulating plate 54 is provided on one side of the coil 50 so as to face the coil 50 via a predetermined distance G (mm). The insulating plate 54 is formed thicker than the insulating plate 52 shown in FIG. The dimension of the metal plate 55 is equal to or greater than the dimension of the coil 50 and is disposed so as to face the entire surface of the coil 50. For the coil 50, for example, the coil 1B to the coil 1G which are the embodiments of the present invention described above are used. The predetermined distance G (mm) is the same as the thickness of the insulating plate 54 and is selected to be 10% or more of the coil outer diameter D. Since the fluctuation of the characteristics of the coil 50 is small, the longer the predetermined distance G (mm) is, the better. When one metal is provided with a metal plate 55 as shown in FIG. 27, a coil capable of preventing characteristic variation when another metal body is close to the coil 50 can be realized.

(Concerning the configuration of the coil that prevents both the proximity effect of the metal body and unnecessary radiation)
FIG. 28 is a diagram illustrating a configuration of the coil 1m that prevents both the proximity effect of the metal body and unnecessary radiation. 28, in FIG. 27, the magnetic material plate 51 is replaced with a predetermined distance G (mm) provided between the insulating plate 54 and the metal plate 55 disposed on the opposite side of the surface facing the coil 50. Is provided. For the coil 50, for example, the coil 1B to the coil 1G which are the embodiments of the present invention described above are used. The metal plate 55 is a metal or alloy having diamagnetic or paramagnetic magnetic properties and has a thickness of 0.1 mm or more. The metal plate 55 has the same dimensions as the magnetic material plate 51. As described in Patent Document 3, if the metal plate 55 is made larger than the dimension of the magnetic material plate 51, there is a problem when the wire winding outer diameter of the opposing coil is larger than the wire winding outer diameter of the coil 1 m. Arise. The coil configuration in the case where the outer diameter of the winding of the opposing coil is larger than the outer diameter of the winding of the coil 1m will be described later.

  FIG. 29 is a diagram showing a configuration of a coil 1n that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 29, the insulating plate 52 is provided between the magnetic material plate 51 and the metal plate 55. The effect of the insulating plate 52 is as described above. The metal plate 55 is a metal or alloy having diamagnetic or paramagnetic magnetic properties and has a thickness of 0.1 mm or more.

  FIG. 30 is a diagram showing a configuration of a coil 1p that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 30, an insulating plate 52 is provided between the coil 50 and the magnetic material plate 51. The effect of the insulating plate 52 is as described above. The metal plate 55 is a metal or alloy having diamagnetic or paramagnetic magnetic properties and has a thickness of 0.1 mm or more.

  FIG. 31 is a diagram showing a configuration of a coil 1q that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 31, a magnetic material plate 511 and a magnetic material plate 512 are provided on one surface side of the coil 50. By providing the two magnetic material plates 511 and 512, the inductance of the coil 50 can be increased and the Q of the coil 50 can be increased. Also, by providing the two magnetic material plates 511 and 512, fluctuations in coil characteristics due to the type and thickness of the metal plate 55 provided on the magnetic material plate 512 side can be reduced. Therefore, unlike the coil 1k to the coil 1p described above, the metal plate 55 can have any magnetic property and thickness. It is more preferable to use a metal or alloy having a diamagnetic or paramagnetic magnetic property and having a thickness of 0.01 mm or more as the metal plate 55. For the coil 50, for example, the coil 1B to the coil 1G which are the embodiments of the present invention described above are used.

FIG. 32 is a diagram showing a configuration of a coil 1r that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 32, a magnetic material plate 511 is provided on one side of the coil 50. The insulating plate 52 is provided between the magnetic material plate 511 and the magnetic material plate 512, and the metal plate 55 is provided on the magnetic material plate 512 side. Regarding the function and effect of the insulating plate 52,
As described above. The metal plate 55 has the same dimensions as the magnetic material plates 511 and 512. The thickness I (mm) of the insulating plate 52 is selected similarly to the coil 1j.

  FIG. 33 is a diagram showing a configuration of a coil 1s that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 33, the insulating plate 52 is provided between the magnetic material plate 512 and the metal plate 55. The effect of the insulating plate 52 is as described above. The metal plate 55 has the same dimensions as the magnetic material plate 512.

  FIG. 34 is a diagram showing a configuration of a coil 1t that prevents both the proximity effect of a metal body and unnecessary radiation according to another embodiment of the present invention. In FIG. 34, the insulating plate 52 is provided on one side of the coil 50. The insulating plate 52 is provided with magnetic material plates 511 and 512 and a metal plate 55. The effect of the insulating plate 52 is as described above. The metal plate 55 has the same dimensions as the magnetic material plates 511 and 512.

  Even when three or more magnetic material plates are used, the place where the insulating material is provided is between the plurality of magnetic material plates 511 and 512, between the coil 50 and the magnetic material plate 51, as shown in FIG. There are various embodiments, such as between the plate 51 and the metal plate 55. The effect of the insulating plate 52 is as described above.

  As described above, in the coils 1g to 1t, the coil 50 includes the effective series resistance of one coil alone in the air-core state as Rwa (Ω), and the other coil facing the one coil in the air-core state. Assuming that the effective series resistance of one coil when short-circuited is Rsa (Ω), one coil satisfying Rsa> Rwa at 100 kHz is used. For example, as the coil 50, the coils 1B to 1G which are the embodiments of the present invention described above are used.

  Furthermore, when the coil 1g is equipped with the magnetic material plate 51 or 511, 512 and the metal plate 55, and the coil 1t is one coil, the coil 1g to the coil 1t satisfy Rsa> Rwa at 100 kHz. Is preferred.

  Assuming that the maximum frequency satisfying Rsa> Rwa from coil 1g to coil 1t is f1 (Hz), power transmission device 100 equipped with coil 1t to coil 1t transmits power at a frequency less than f1 (Hz). .

  When the coils 1g to 1t are the power transmission coil 1 of the power transmission device 100, the coils 1g to 1t are driven at a frequency fa (Hz) less than f1 (Hz) by the AC power supply 30b.

  When the power transmission unit 30 of the power transmission device 100 includes the coil 1g to the coil 1t, the power transmission unit 30 including the coil 1g to the coil 1t is a power transmission device of the power transmission device of the present invention.

  When the power reception unit 40 of the power transmission device 100 includes the coil 1g to the coil 1t, the power reception unit 40 including the coil 1g to the coil 1t is a power reception device of the power transmission device of the present invention.

  Detailed actions and effects of the coils 1g to 1t, which are coils equipped with at least one of the metal plate 55 and the magnetic material plates 51, 511, and 512, will be described in detail later.

(When the opposing coil outer diameter is different)
FIG. 35 is a three-dimensional view of elements constituting both opposing coils. As shown in FIG. 35, the opposing coils 1ma and 1mb are equivalent to the coil 1m shown in FIG. 28, and the coils 50a and 50b, the magnetic material plates 51a and 51b, and the metal plates 55a. 55b.

  FIG. 36 and FIG. 37 are diagrams showing the opposed state when the outer diameters of the coil 50a and the coil 50b are different from each other and are configured in the same manner as the coil 1m shown in FIG. The coil 50b having a large conductor winding outer diameter is referred to as a power transmission coil, and the coil 50a having a small conductor winding outer diameter is referred to as a power receiving coil.

  FIG. 36 shows a case where the centers of the opposing power transmission coil 1mb and power reception coil 1ma are coincident. In FIG. 36, Da represents the maximum distance from the center of the winding surface of the power receiving coil conducting wire constituting one coil to the end of the winding surface, and the winding of the power transmitting coil conducting wire constituting the other coil facing one coil. The minimum distance from the center of the winding surface to the end of the winding surface is Db, and when Db> Da, the minimum distance from the center of the winding surface of the receiving coil conducting wire to the end of the magnetic material plate 51b is at least , Db + (Db−Da) × 2.

  In FIG. 37, the conductive wire winding surface of the opposing power receiving coil 1 ma is in the conductive wire winding surface of the power transmission coil 1 mb indicated by a dotted line, and the end of the conductive wire winding surface of the power receiving coil 1 ma and the power transmission coil 1 mb The state which the end of conducting-wire winding surface exists in the same position is shown. As shown in FIG. 37, the relative positions of the opposing power transmission coil 1mb and power reception coil 1a are permissible until the end of the power reception coil conductor winding surface is within the end of the power transmission coil conductor winding surface. As described in Patent Document 3, if a magnetic material plate having the same dimensions as the conductive wire winding surface of the power receiving coil 1ma is equipped and a metal plate larger than the magnetic material plate is equipped, in the case of FIG. The metal plate faces the conductive wire winding surface of the power transmission coil 1mb. When the metal plate faces the conductive wire winding surface of the power transmission coil 1mb, the characteristics of the power transmission coil 1mb naturally change. Furthermore, even if a magnetic material plate having the same dimensions as the conductive wire winding surface is provided, if a metal plate exists on the back surface of the magnetic material plate, depending on the relative position of the power transmission coil 1mb and the power reception coil 1ma, the power transmission coil 1mb The metal plate faces the conductive wire winding surface. Therefore, the shape and size of the magnetic material plate must be determined so that the magnetic material of the other coil faces the conductive wire winding surface.

  In the case of FIG. 37, if the outer diameter of the power transmission coil 1mb is D1 and the outer diameter of the power reception coil 1ma is D2, the outer diameter of the magnetic material plate of the power reception coil 1ma is Y2 + D1-D2, and D2 + 2 × Y The dimensions are required. Alternatively, the dimension of 2 (D1-D2) + D2 = 2 × D1-D2 is required.

  FIG. 38 is a diagram illustrating a relative positional relationship between the power transmission coil 1mb and the power reception coil 1ma when both the power transmission coil 1mb and the power reception coil 1ma are elliptical. 38, the minimum outer diameter of the power transmission coil 1mb is D11, the maximum outer diameter of the power transmission coil 1mb is D12, the minimum outer diameter of the power reception coil 1ma is D21, the maximum outer diameter of the power reception coil 1ma is D22, Y1 = D11-D22, Y2 = Assuming D12-D21, the magnetic plate must have a minimum outer shape that is at least 2 × Y1 larger than the minimum outer diameter of the coil 50. The magnetic material plate has a maximum outer diameter of 2 × Y2 or less than the maximum outer diameter of the coil 50, and the power receiving coil 1ma has a conductor winding surface within the conductor winding surface of the power transmission coil 1mb. The size and shape of the magnetic material plate 51a are selected so that the magnetic material plate 51a mounted on the coil 1ma always faces the conductive wire winding surface of the power transmission coil 1mb.

  The above can be generalized as follows. The “coil” refers to a conductive wire winding surface. When the size and shape of the opposing coils are different, the maximum distance from the center or center of gravity of the larger coil to the coil end surface is La, and the minimum dimension from the center or center of gravity of the smaller coil to the coil end surface is Lb. La and Lb correspond to radii with respect to the diameters of D1 and the like described above. Therefore, the smaller coil has a dimension of 4 (La−Lb) + 2 × Lb = 4 × La−2 × Lb or less, and the smaller coil facing surface enters the larger coil facing surface. The smaller coil must be equipped with a magnetic material plate having a size and shape so that the magnetic material plate faces the larger coil facing surface.

  FIG. 39 is a diagram illustrating an example of the opposing power transmission coil 1 and power reception coil 2 when the coil facing surface is not circular. For example, as shown in FIG. 39A, it is assumed that the power receiving coil 2 is a square having a side of 40 mm. The power receiving coil 1ma is completely in the plane of the power transmitting coil 1mb. In this case, the minimum outer diameter D2 of the power reception coil 1ma is 40 mm, and the maximum outer diameter D1 of the power transmission coil 1mb is 57 × √2≈80 mm. Therefore, the magnetic material plate 51 provided in the power receiving coil 1ma may be a square having a side of 80 mm as shown in FIG.

  When Y in FIG. 37 is applied to FIG. 39A, Y = D1−D2 = 80−40 = 40 mm. The power receiving coil 1ma is equipped with a magnetic material plate having an outer diameter of D2 + Y × 2. Accordingly, in FIG. 37, the receiving coil 1ma is a square magnetic material plate whose diagonal is D2 + Y × 2 = 40 + 40 × 2 = 120 mm (maximum outer dimension) and whose one side is 120 / √2≈85 mm (minimum outer dimension). You must equip at least 51. However, as shown in FIG. 39B, even when the relative angle between the power receiving coil 1ma and the power transmitting coil 1mb is changed and the power receiving coil 1ma is completely within the plane of the power transmitting coil 1mb, the power transmitting coil 1mb is magnetically coupled. It can be seen that the material plates 51 are completely opposed.

  36 to 39, the area of the magnetic material plate 51 of the power transmission coil 1mb is equal to or larger than the area of the coil, the coil is within the area of the magnetic material plate 51, and the magnetic material of the power receiving coil 1ma. The plate is within the area of the power transmission coil 1mb and is not related to the magnetic material plate on the power transmission coil 1mb side. Further, the outer diameters of the power transmission coil 1mb and the power reception coil 1ma are the same or the outer diameter of the power reception coil 1ma is smaller, and the conductive wire winding surface of the power reception coil 1ma does not protrude from the conductive wire winding surface of the power transmission coil 1mb. As long as these conditions are satisfied, the outer diameter shape of the coil and the magnetic material plate is not limited.

  However, when the smaller coil end is completely within the larger coil end, the magnetic plate is sized and shaped to face the larger coil as the magnetic plate to be equipped on the smaller coil. And the mounting position of the magnetic plate to be mounted on the smaller coil must be selected.

(Characteristics of coil 1f to coil 1t)
FIG. 40 shows the effective in each coil when the coil 1G shown in FIG. 15 is used as the coil 1f shown in FIG. 38, the coil 1g shown in FIG. 24, the coil 1h shown in FIG. 25, and the coil 1j shown in FIG. It is a figure which shows the relationship with the frequency of series resistance Rw ((ohm)).

  FIG. 41 is a diagram illustrating a relationship between the Q frequency of each coil when the coil 1G is used as each of the coils 1f, 1g, 1h, and 1j.

  As is apparent from FIG. 40, when comparing the coil 1g and the coil 1f, in which the coil 50 is provided with the insulating plate 52 and the magnetic material plate 51, for example, the effective series resistance Rw (Ω) at 1 MHz is greater for the coil 1g. Low. As described above, by providing the coil 50 with the insulating plate 52, the effective series resistance Rw (Ω) of the coil in the high frequency region can be lowered. This tendency is the same even when the coil 1j in which the coil 50 is equipped with the magnetic material plate 511, the insulating plate 53, and the magnetic material plate 512 is compared with the coil 1h in which the coil 50 is equipped with the magnetic material plates 511 and 512. .

  As is clear from FIG. 41, the coil 1g equipped with the insulating plate 52 in the coil 50 has a higher Q at 1 MHz than the coil 1f equipped with only the magnetic material plate 51, for example. As is apparent from FIGS. 40 and 41, even if the insulating plate 52 is provided in the coil 50, there is almost no difference in effective series resistance Rw (Ω) and Q at a frequency of 100 kHz. Thus, by providing the coil 50 with the insulating plate 52, the Q of the coil in the high frequency region can be increased. The effect of this insulating plate is the same in the coils 1m to 1t equipped with the magnetic material plates 51, 511, and 512 described above. Therefore, the description regarding an insulating board is abbreviate | omitted about the coil 1m to the coil 1t.

(Explanation when various metal plates face the coil)
FIG. 42 is a characteristic diagram showing the effective series resistance Rw (Ω) of the single coil 1G and the inductance Lw (μH) of the single coil 1G when various metal plates are brought close to each other using the coil 1G. .

  Although shown in FIG. 42, the coil of the configuration (1) represents the effective series resistance Rw (Ω) and the inductance Lw (μH) of the coil 1G alone. The coil of the configuration (2) is left blank because it is used when a magnetic material plate described later is provided. The coil of the configuration (3) is in a state where an aluminum (Al) foil having a thickness of 12 μm is brought close to the coil 1G. The coil of the configuration (4) is in a state where an aluminum plate having a thickness of 0.1 mm is brought close to the coil 1G. The coil of the configuration (5) is in a state in which an aluminum plate having a thickness of 0.5 mm is brought close to the coil 1G. The coil of the configuration (6) is in a state in which an aluminum plate having a thickness of 3 mm is brought close to the coil 1G. The coil of the configuration (7) is in a state in which a copper foil (Cu) having a thickness of 35 μm is brought close to the coil 1G. The coil of the configuration (8) is in a state in which a copper plate having a thickness of 0.1 mm is brought close to the coil 1G. The coil of the configuration (9) is in a state in which a copper plate having a thickness of 0.5 mm is brought close to the coil 1G. The coil of the configuration (10) is in a state in which an iron plate (Fe) having a thickness of 0.5 mm is brought close to the coil 1G. In FIG. 42, the effective series resistance Rw (Ω) and the inductance Lw (μH) at 100 kHz of the coils of the configurations (3) to (10) are compared with the characteristics of the coil 1G alone of the configuration (1). It is shown as a bar graph.

(Construction and characteristics of the coil excluding the influence of proximity to the metal plate)
FIG. 43 is a characteristic diagram showing the effective series resistance Rw (Ω) of the single coil 1Ga and the inductance Lw (μH) of the single coil 1Ga when various metal plates are brought close to the coil 1G with a distance of 10 mm. It is. In other configurations, the metal plate provided in the coil 1Ga is the same as described above. Coil 1Ga is an embodiment of coil 1k shown in FIG.

  44 shows the effective series resistance Rw (Ω) of the coil 1Gb alone and the inductance Lw (μH) of the coil 1Gb alone when various metal plates are brought close to the coil 1G having the single magnetic material plate 51. FIG. The coil of the configuration (2) is in a state where the coil 1Gb is equipped with one magnetic material plate 51 and no metal plate is brought close to the coil 1Gb. In other configurations, the metal plate provided in the coil 1G is the same as described above. Coil 1Gb is an embodiment of coil 1f shown in FIG.

  FIG. 45 shows the effective series resistance Rw (Ω) of the coil 1Gc alone and the coil 1G alone when various metal plates are brought close to the coil 1Gc provided with the two magnetic material plates 511 and 512 in the coil 1G. It is a characteristic view which shows inductance Lw ((micro | micron | mu) H). The coil of the configuration (2) is in a state where the coil 1Gc is equipped with the two magnetic material plates 511 and 512 and the metal plates are not brought close to each other. In other configurations, the metal plate provided in the coil 1Gc is the same as described above. Coil 1Gc is an embodiment of coil 1q shown in FIG.

  FIG. 46 is a characteristic diagram showing the Q of each coil at 100 kHz of the coil 1Ga, the coil 1Gb, and the coil 1Gc described above.

42 to 46 refer to the frequency characteristics of the effective series resistance Rw (Ω) of the coil 1G alone shown in FIG. 15, and the effective series resistance Rw (Ω) is substantially equal to the DC resistance of the coil 1G.
It is measured by selecting Hz.

  First, the characteristic diagram shown in FIG. 42 will be examined. 42 that the effective series resistance Rw is about 0.2Ω and the inductance Lw is about 14 μH in the configuration of the coil 1G alone. In the coil 1G having the configuration (3) in which 12 μm aluminum foils are closely opposed to each other, the effective series resistance Rw is 3Ω or more, and the inductance Lw is reduced to about 5.5 μH. Looking at the characteristic diagrams of configurations (4) to (6) in which an aluminum plate of various thicknesses, which is a paramagnetic metal, is opposed to the coil 1G, when the aluminum thickness is 0.1 mm or more, the thickness is As the value increases, the effective series resistance Rw (Ω) of the coil decreases and the inductance Lw (μH) increases. This tendency is the same in the coils of configurations (8) and (9) in which copper, which is a diamagnetic metal, is opposed to the coil 1G. When the copper plate is thin, the effective series resistance Rw (Ω) of the coil decreases and the inductance Lw (μH) also decreases. When the thickness of the copper plate is about 0.5 mm, the effective series resistance Rw (Ω) of the coil has characteristics that are not significantly different from the 0.5 mm aluminum plate of the configuration (5). When a 0.5 mm iron plate, which is a ferromagnetic metal, is placed close to and opposed to the coil 1G, the effective series resistance Rw (Ω) is the same as when the aluminum foil in the configuration (3) is placed close to the coil 1G. Is 10 times or more, and the inductance Lw is reduced to about 8.5 μH. That is, the coil characteristics do not fluctuate due to the magnetic properties of the metal as described in Patent Document 4, but the coil thickness depends on the thickness of the metal plate that is in close proximity to the coil wound in a planar air-core spiral shape. Characteristics vary.

  Each coil of configurations (3) to (10) shown in FIG. 42 has an effective series resistance Rw (Ω) that is excessive and an inductance Lw (μH) that is excessive compared to the air-core state. Therefore, the coils of configurations (3) to (10) shown in FIG. 42 cannot actually be used as coils of the power transmission device. FIG. 42 shows data to be compared with FIGS. 43 to 46 shown below. Incidentally, in the aluminum foil having a thickness of 12 μm, although the decrease rate of the inductance is small, the increase rate of the effective series resistance Rw (Ω) is 15 times or more that of the coil 1G alone. When the thickness of the aluminum plate of the configuration (4) is 0.1 mm, it can be inferred that some transition point exists between 10 μm and 100 μm (0.1 mm) when the inductance increases. .

  Looking at the data of the copper plates of the configuration (7) to the configuration (9), a tendency almost the same as that of the configuration (4) to the configuration (6) can be seen. Since copper is a diamagnetic metal, it cannot be determined, but with a thickness of about 30 μm as a boundary, a metal plate thinner than that has a reduced rate of decrease in inductance Lw (μH) and has an effective series resistance Rw (Ω). It is estimated that the rate of increase will rise rapidly. This is because the 0.1 mm thick aluminum plate and the copper plate in the configuration (4) and the configuration (8) are both close to the 0.5 mm thick aluminum plate and the copper plate in the configuration (5) and the configuration (9). It can be inferred from the fact that the characteristics are shown. The increase rate of the effective series resistance Rw (Ω) of the 35 μm-thick copper foil of the configuration (7) is smaller than the increase rate of the effective series resistance Rw (Ω) of the aluminum foil of the configuration (3) of 12 μm. Is presumed to be due to the thickness.

  FIG. 43 shows the characteristics of the coil 1Ga shown in FIG. 27 in which the various metal plates shown in FIG. 42 are opposed to the coil 1G via an insulator of 10 mm. As is apparent from comparison with FIG. 42, in FIG. 43, the increase rate of the effective series resistance Rw (Ω) and the decrease rate of the inductance Lw (μH) are also small. It can be seen that the characteristics of the coil 1Ga are greatly improved by using an insulator of 10 mm. In particular, the value of the inductance Lw has decreased only to about 11.7 μH compared to about 13.7 μH in the air-core state. The value of Lw (μH) is almost the same in each configuration, and can be used for a power transmission device. However, when the 12 μm thick aluminum foil of the configuration (3), the 35 μm thick copper foil of the configuration (7), and the 0.5 mm iron plate as the ferromagnetic material of the configuration (10) face the coil 1G, the coil 1G Since the increase rate of the effective series resistance Rw (Ω) is large and power loss due to Rw (Ω) occurs, such a configuration is not suitable for use in a power transmission device.

  That is, referring to FIG. 43, a means for providing a predetermined distance G (mm) between the coil and the metal is provided, and a metal having a diamagnetic or paramagnetic magnetic property of 0.1 mm or more as a metal plate. Alternatively, the use of an alloy can eliminate the influence of a metal body close to the air-core coil. In the configurations other than the configurations (1) and (2) shown in FIG. 43, various metals including a ferromagnetic metal such as iron are brought close to the opposite side of the metal plate to the coil facing surface. Neither change in μH) nor change in effective series resistance Rw (Ω) is observed. In addition, there is no change in power transmission performance.

  In paragraph No. 0022 of Patent Document 4, it is described that a metal plate having a size smaller than that of the magnetic field type antenna (coil) may be used. However, in order to eliminate the influence of the proximity of the metal body of the coil formed by winding the conducting wire in a plane spiral shape, the above-mentioned predetermined distance G (mm) is provided, and the thickness is 0. It must be equipped with a metal or alloy plate of the same size as the coil that is 1 mm or more.

  Note that paragraph number 0022 of Patent Document 4 describes that the metal plate is divided. The inventor of the present application measured the characteristics by dividing a 0.1 mm copper foil with the coil of the configuration (8) using the coil 1Ga, and Lw = 12.3 μH and Rw = 0.29Ω. This configuration has better characteristics than Lw = 11.7 μH and Rw = 0.23Ω when the copper plate is not divided. As described above, this is presumably because the eddy current loss that increases in proportion to the volume of the metal body decreases. This is also described in paragraph No. 0022 of Patent Document 4.

  However, in the configuration in which the 0.1 mm copper foil is divided and installed, when a 0.5 mm thick iron plate is brought close to the opposite surface of the copper plate coil, the Lw decreases to 12.3 μH and the Rw becomes 0.1. Increased to 38Ω. If the copper plate is not divided, the inductance Lw (μH) is smaller than the divided copper plate, but the effective series resistance Rw (Ω) is small. There was no change in both Lw (μH) and Rw (Ω) even when they were close to each other. Therefore, in an embodiment in which the metal plate described in paragraph No. 0022 of Patent Document 4 is divided, and in an embodiment in which the size of the metal plate is made smaller than that of the coil, it is described in Paragraph No. 0022 of Patent Document 4. The effect of eliminating the fluctuation of the coil characteristics when the metal body is close to the coil being coiled cannot be expected.

  As described above, in the embodiment of the coil 1k shown in FIG. 27, by providing means for making the distance between the coil and the metal plate constant between the coil and the metal plate, another metal body comes close to the back surface of the metal plate. However, fluctuations in the inductance Lw and effective series resistance Rw of the coil 1k can be eliminated. A coil 1k illustrated in FIG. 27 is suitable for a power transmission unit in which metal plates can be installed at regular intervals on the back surface of the coil. When the power transmission unit is placed on a steel desk, fluctuations in the inductance Lw and effective series resistance Rw of the coil 1k can be eliminated, and predetermined power transmission performance can be maintained. The predetermined distance G is 10 mm in the coil 1G, and good results are obtained. However, the predetermined distance G (mm) varies depending on the outer diameter D of the coil. Since the outer diameter D of the coil 1G is 50 mm, a predetermined distance G (mm) is determined by considering a margin, for example, G ≧ D / 10 = 5 mm.

Next, consider FIG. 44, which is a characteristic diagram of the embodiment of the coil 1m shown in FIG. The configuration (2) in FIG. 44 shows the effective series resistance Rw (Ω) and the inductance Lw (μH) of the coil 1Gb alone in which a magnetic material plate having a thickness of 0.3 mm is attached to the coil 1G. In the coil 1Gb alone, the inductance Lw (μH) is increased and the effective series resistance Rw (Ω) is hardly changed compared to the coil 1G alone in the air-core state. The effective series resistance Rw (Ω) and inductance Lw (μH) of each of the coils in the configurations (3) to (10) in which the same metal plate as that shown in FIGS. 42 and 43 is provided on the opposite surface of the coil 1Gb are shown. ing. In the configurations (3) to (10), the value of the inductance Lw is almost the same. However, as is apparent from FIG. 44, as in FIG. 43, a configuration (3) equipped with an aluminum foil having a thickness of 12 μm, a configuration (7) equipped with a copper foil having a thickness of 35 μm, 0 The coil of the configuration (10) equipped with a steel plate having a thickness of 0.5 mm has a high effective series resistance Rw (Ω).

  Furthermore, the characteristics of each component of the coil 1q having the configuration shown in FIG. 31, which is another embodiment of the present invention, will be examined with reference to FIG. The coil 1q having the configuration shown in FIG. 31 is equipped with two magnetic material plates 511 and 512. The two magnetic material plates 511 and 512 are preferably stacked with an insulating layer. Referring to FIG. 45, the characteristics of a single coil 1Gc equipped with two magnetic material plates in the coil 1G are shown in the configuration (2), and the inductance Lw is about 22.5 μH, which is about 14 μH in an air-core state. , About 1.6 times.

  Compared to FIG. 44, in all of the configurations (3) to (10), the inductance Lw exceeds 20 μH. Further, in all of the configurations (3) to (10), the value of the inductance Lw (μH) is substantially the same. Compared with FIG. 44, the 12 μm-thick aluminum foil of the configuration (3), the 35 μm-thick copper foil of the configuration (7), and the 0.5 mm iron plate as the ferromagnetic material of the configuration (10) are used as the coil 1Gc. Even when facing each other, there is a feature that there is almost no change in the effective series resistance Rw (Ω). That is, by mounting two magnetic material plates in an overlapping manner, the coil is not affected by the magnetic properties or thickness of the metal provided on the opposite side of the coil facing surface of the magnetic material plate. By adopting a configuration like the coil 1q having the configuration shown in FIG. 31, the above-described proximity effect of the metal body can be eliminated with a thin metal such as an aluminum foil.

  In addition, as the magnetic material plate, a thickness of 0.01 mm to 1.5 mm and various configurations such as a magnetic material powder solidified with a binder, an amorphous type, and a ferrite type were tested. The characteristics due to the difference in the metal plates were the same as those shown in FIGS. Further, in the configuration of the coil 1f and the coil 1q, the increase in effective series resistance was large when the increase in inductance was large. At 100 kHz, the increase rate of the inductance and the increase rate of the effective series resistance were almost equal for any magnetic material plate. As will be described later, these actual measurement results indicate that this configuration rule has generality.

  The above results are summarized in FIG. First, the configurations (5) and (9) of the coil 1Ga that is an example of the coil 1k and the coil 1Gb that is an example of the coil 1f will be compared. The value of Q is almost the same when the coil 1Ga is equipped with a 0.5 mm-thick paramagnetic metal aluminum plate and when the coil 1Ga is equipped with a diamagnetic metal copper plate. Also in the coil 1Gb, when the coil 1Gb is equipped with a 0.5 mm-thick paramagnetic metal aluminum plate and when the coil 1Gb is equipped with a diamagnetic metal copper plate, the Q value is almost does not change. That is, as described in Patent Document 4, the diamagnetic metal is not suitable for eliminating the influence of the coil characteristics due to the metal body close to the coil. It can be seen from FIG. 46 that the metal mounted on the coil deteriorates the coil characteristics only in the thickness of the metal plate in the case of the ferromagnetic metal and the metal other than the ferromagnetic metal. Furthermore, in the coil 1Gc having the configuration of the coil 1q equipped with two or more magnetic material plates, the coil characteristics due to the metal body close to the opposite side of the opposing surface of the coil regardless of the magnetic properties of the metal and the thickness of the metal plate. It can be seen that the fluctuation of the can be prevented.

  In addition, although illustrated in FIG. 46, Q of the coil 1G single core of the configuration (1) is about 42.5. The coils having the respective configurations shown in FIG. 46 that are not suitable as the power transmission coil are the coil 1k and the coil 1m, which are the configuration (3), the configuration (7), and the configuration (10). This rule is based on the Q of the coil 1G air core alone, which is 70% of about 42.5, Q = 30, and a coil having a Q higher than the reference value is selected. Any coil having a Q higher than the line Q = 30 shown in FIG. However, FIG. 46 compares the Qs of the coils of each configuration. Actually, the frequency characteristic of the effective series resistance Rw (Ω) shown in FIGS. 43 to 45 is measured, and the frequency characteristic of the effective series resistance Rw (Ω) is poor because the coil has an excessive effective series resistance Rw (Ω). The coil must be excluded. It is preferable to judge from both the Q of the coil and the effective series resistance Rw (Ω). The reference value Q = 30 is defined as a condition having a power transmission performance of 90% or more compared to the case where power transmission is actually performed using the air-core coil 1G.

(Explanation that the power transmission coil configuration rules are general)
As described above, a coil with good performance cannot be realized simply by defining a specific configuration of the coil. However, the specific configuration of the coil in the present embodiment exhibits the same effects regardless of the coil type, winding method, outer diameter, and the like. That is, the specific configuration of the coil in the present embodiment has an effect of suppressing fluctuations in coil characteristics when a metal body is close to the back surface of the coil regardless of the wire type, winding method, outer diameter, and the like of the coil. An example is shown below.

  47 is a characteristic diagram obtained by measuring the effective series resistance Rw (Ω) and the inductance Lw (μH) of the coil 1D shown in FIG. 12 in the same configuration as the coil 1k shown in FIG. . In FIG. 47, the distance between the coil 1D and the metal plate is set to 5 mm and measured.

  FIG. 48 is a characteristic diagram in which the effective series resistance Rw (Ω) and the inductance Lw (μH) are measured for the coil 1E shown in FIG. 13 in the same configuration as the coil 1k shown in FIG. . In FIG. 48, the distance between the coil 1E and the metal plate is set to 10 mm and measured.

  43, 47, and 48 are compared. In the coil 1k having the configuration shown in FIG. 27, the metal plate 55 is configured using an aluminum foil as an example (3), and is configured using a copper foil as an example (7). ), It can be seen that the effective series resistance Rw (Ω) of each of the coils of the example (10) configured using the iron plate is increased as compared with the coils of other configurations.

  This tendency is the same for both the coil 1D and the coil 1E in the configuration of the coil 1m shown in FIG. 28 and the configuration of the coil 1q shown in FIG. That is, when only a predetermined distance G (mm) is provided between the coil and the metal plate, or when one magnetic material plate is provided between the coil 50 and the metal plate 55, the metal plate 55 is configured using aluminum foil. In the coil of the example (3), the example (7) configured using the copper foil, and the example (10) configured using the iron plate, the effective series resistance Rw (Ω) is increased more than the coils of the other configurations. To do. When two or more magnetic plates are provided between the coil 50 and the metal plate 55, there is almost no increase in effective series resistance Rw (Ω) as shown in FIG. 45 depending on the type and thickness of the metal plate 55.

  The effect of the coil 1k having the configuration shown in FIG. 27 is to prevent fluctuations in coil characteristics when a metal body is close to the opposite side of the coil facing surface, as described in Patent Document 4. The operational effects of the coil 1m having the configuration shown in FIG. 28 and the coil 1q having the configuration shown in FIG. 31 also prevent fluctuations in coil characteristics when a metal body is close to the opposite side of the opposing surface of the coil. Another effect of the coil 1m having the configuration shown in FIG. 28 and the coil 1q having the configuration shown in FIG. 31 is prevention of unnecessary radiation. In order to prevent unnecessary radiation, the configuration of the coil 1q shown in FIG. 31 is preferable. As described above, the coil 1q is hardly affected by the material and thickness of the metal plate. Therefore, even if it is the coil 1h shown in FIG. 38, the proximity | contact effect of a metal body can be excluded.

  In FIG. 43 to FIG. 45, the data of the coil 1n, the coil 1p, the coil 1r, the coil 1s, and the coil 1t that are constituted by the metal plate, the magnetic material plate, and the insulating plate are omitted. This is because the effect of the metal plate 55 is the same as shown in FIGS. As shown in FIGS. 40 and 41, the coil having such a configuration has an effect of suppressing an increase in effective series resistance Rw (Ω) in a high frequency region and increasing Q. In actual measurement, in the high frequency region, the effective series resistance Rw (Ω) of each coil described above decreases compared to the coils 1n and 1q, and the Q of each coil increases compared to the coils 1n and 1q. Has been confirmed to do.

As described above, the frequency characteristics of Rw, Rs, and Rn of the coil 1G2 set are extremely good. According to FIG. 15, the maximum frequency f1 satisfying Rs> Rw is 10 MHz or more, and the maximum frequency f2 satisfying Rs> Rn ≧ Rw is about 2.2 MHz. However, especially in the coil 1m and the coil 1q having the above-described configuration, f2 (Hz) is lowered. Consider this point below.

(Description of coil equipped with magnetic material plate)
FIG. 49 is a diagram illustrating an example of a coil 1 </ b> H in which a conducting wire 56 is wound around a core 53. FIG. 49A is a configuration diagram of a single coil 1H, and FIG. 49B is a diagram illustrating a state in which two coils 1Ha and 1Hb are inductively coupled.

  In FIG. 49 (B), the cores 53a and 53b of the two coils 1Ha and 1Hb are not directly contacted but bonded via a double-sided tape (not shown).

  FIG. 50 shows the effective series resistance Rw (Ω) of the single coil of the coil 1H having the configuration of FIG. 49 and the two coils 1Ha and 1Hb inductively coupled and short-circuiting both ends of the other coil. The relationship between Rw, Rs, Rn and frequency when the effective series resistance Rs (Ω) of the coil and the effective series resistance of one coil when the both ends of the other coil are opened are Rn (Ω). FIG.

  In FIG. 51, the inductance Lw (μH) of the coil 1H alone and the two coils 1Ha and 1Hb are inductively coupled, and the other coil is opened when the other coil is short-circuited. It is a figure which shows the relationship between Lw, Ls, Ln, and a frequency when the inductance of one coil at this time is set to Ln (microH).

  As far as FIG. 50 is referred to, the coil 1H has a small difference between Rw (Ω) and Rs (Ω). This is because the coupling coefficient between both coils is as small as about 0.3. The coupling coefficient kr approximately obtained by the method described above is also about 0.3. Further, referring to FIG. 51, Ln> Ls> Lw in the frequency region from 1 kHz to 10 MHz.

The above-described equation (4) is expressed by Z = (R1 + A ² R2) + jω (L1-A ² L2),
In the above formula (5), Z = R1 + jωL1 and A2 ≧ 0. Therefore, when the other coil inductively coupled with one coil is short-circuited, the inductance of one coil must be reduced. . That is, the relationship Lw = Ln> Ls must be satisfied.

  FIG. 52 is a diagram illustrating the relationship between Lw, Ls, Ln, and the coupling coefficient ki approximately obtained from Lw and Ls and the frequency, measured when the coil 1G is in the air-core state.

  In FIG. 52, as described above, Lw at a frequency of 20 kHz or more except for a frequency of 20 kHz or less where the phase difference φ between the voltage V applied to the coil 1G and the current I flowing through the coil 1G is 80 degrees or less. It can be seen that the circuit theoretical relationship of = Ln> Ls is satisfied up to at least 4 MHz. The difference between FIG. 51 and FIG. 52 is estimated as follows. In FIG. 51, when the same two coils are inductively coupled as shown in FIG. 49B, compared with the case of a single coil as shown in FIG. 49A, the core 53a having a high relative permeability in both coils. , 53b are magnetically coupled, so that Ln (μH) rises. However, when the other coil is short-circuited, Ls (μH) becomes lower than Ln (μH). In this case, it is considered that Ls> Lw is determined by the core material, the relative magnetic permeability of the core, the wire and diameter of the conductive wire, and the coil configuration. When the inventor of the present application measured Lw, Ls, and Ln for various coils having the configuration shown in FIG. 49B, there was a coil satisfying Ls <Lw. However, there is no coil that satisfies Ln = Lw, and Ln> Ls. From the configuration of the transformer and the circuit theory described above, when the coil having the configuration shown in FIG. 49A is inductively coupled as shown in FIG. 49B, both coils are in the same state as the transformer that cannot be separated. It is believed that there is.

  FIG. 53 is a diagram showing a configuration of a transformer 70 in which a primary coil and a secondary coil in which a primary coil 71 and a secondary coil 72 are wound around a toroidal core 73 cannot be separated.

For confirmation, the inventor of the present invention uses a transformer 70 having the configuration shown in FIG. 53 to measure Rn (the transformer cannot measure Rw and Lw because both coils cannot be separated), Rs, Ln, and Ls. I measured at 100 kHz. Ln = 2.83 mH, Ls = 23.4 μH, Rn = 603Ω, Rs = 0.56Ω. Even in a transformer equipped with a core having a high relative permeability, a measurement result contrary to the circuit theory that Rn> Rs is obtained. When the coupling coefficient ki is approximately calculated from the above Ln and Ls, ki = √ ((Ln−Ls) / Ln), so ki = √ ((2830−0.0234) / 2830) = √ (0.99999)
= 0.999995 ≒ 1
It was almost tightly coupled. Thus, in the transformer of FIG. 53, the coupling coefficient ki may be obtained approximately from Ln and Ls. Therefore, in the coil 1m of FIG. 28 and the coil 1q of FIG. 31 described above, it is assumed that Ln> Lw even at a low frequency. It is also assumed that the maximum frequency f1 (Hz) satisfying Rs> Rw and the maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw are also decreased.

  FIG. 54 is a diagram showing a relationship between Lw, Ls, Ln, ki, ki2 and frequency when two coils 1Gb are used in the coil 1Gb which is the coil 1f shown in FIG.

  In FIG. 54, ki2 is obtained from Ln and Ls instead of Lw and Ls.

  ki2 = √ ((Ln−Ls) / Ln). This is an approximate method for obtaining the same coupling coefficient ki as that of the transformer shown in FIG. Both the coupling coefficients ki and ki2 approximately obtained from the aforementioned Lw and Ls are plotted. Ln (μH) is close to twice Lw (μH), but since it takes a square root, there is no significant difference between ki and ki2, and both are 0.9 or more. This is a large value compared to the values of kr and ki plotted in FIG. Thus, the magnetic material plate has an effect of increasing the coupling coefficient between the two coils.

  54, unlike Ln> Ls> Lw in FIG. 51, the relationship is Ln> Lw> Ls. In the present application, various coil characteristics and configurations have been defined for coils having a configuration capable of inductive coupling, focusing on coils having good power transmission performance. However, as shown in FIG. 49, if the coil has a configuration capable of inductive coupling between two identical coils and satisfies the relationship of Rs> Rw at 100 kHz, Not only the power transmission device but also the performance as a coil which is a passive component is good. Furthermore, if the coil satisfies the relationship of Lw> Ls at 100 kHz, the performance as a coil that is a passive component is good. And since the coil of the structure shown in FIG. 49 is equipped with the core, it was conventionally considered that there is no proximity effect of a metal body.

  FIG. 55 shows a coil having the configuration shown in FIG. 49. Various types of metal plates are placed close to the A surface of FIG. 49A in the coil 1H having the characteristics shown in FIGS. 50 and 51, as shown in FIG. FIG. 6 is a characteristic diagram showing an effective series resistance Rw (Ω) at 100 kHz and an inductance Lw (μH).

  When FIG. 55 is compared with FIG. 44, the decrease rate of the inductance Lw (μH) is small, but the increase rate of the effective series resistance Rw (Ω) tends to be almost the same as that of FIG. That is, as described above, the coil having a configuration capable of inductive coupling is affected by the proximity of the metal body, and the effective series resistance Rw (Ω) increases. The coil of the configuration in which the aluminum foil having a thickness of 12 μm is close (3), the configuration in which the copper foil having a thickness of 35 μm is close (7), and the configuration in which the iron plate having a thickness of 0.5 mm is close (10) is effective. The series resistance Rw (Ω) is high as in FIG. Further, the inductance Lw (μH) is reduced by the same configuration as that of FIG. Alternatively, when two coils 1H configured to be inductively coupled are used and both coils are inductively coupled, Ln> Lw. Some configurations of the coil are contrary to the circuit theory of Ln> Lw> Ls.

  FIG. 56 is a diagram showing the relationship between Rw, Rs, Rn and frequency when two coils 1Gb are used in the coil 1Gb in which the coil 1G is provided with a magnetic material plate like the coil 1f shown in FIG. is there.

  Looking at the frequency characteristics of Rw, Rs, and Rn shown in FIG. 56, it can be seen that the maximum frequency f1> 10 MHz that satisfies Rs> Rw shown in FIG. 15 has decreased to f1 = 1.35 MHz. . Furthermore, it can be seen that the maximum frequency f2 satisfying Rs> Rn ≧ Rw shown in FIG. 15 is reduced from 2.15 MHz to 150 kHz.

On the other hand, as shown in FIG. 50, the maximum frequency f2 of the coil 1H that satisfies Rs> Rn ≧ Rw is as high as 1.3 MHz. This is considered because the coupling coefficient between the two coils 1H is as low as about 0.3 and the coupling coefficient between the two coils 1Gb is as high as 0.9 or more. Therefore, the coupling coefficient between the two coils at 100 kHz is k, and the reference is 150 kHz, which is the highest frequency f2 in which two coils 1Gb satisfy Rs> Rn ≧ Rw. Then, the 150kHz, divided by ^{k 2.} In the case of the coil 1H with ki = 0.27, when the prescribed value of f2 (Hz) is calculated, f2 = 150 / 0.27 ² ≈2.05 MHz. From FIG. 50, the f2 of the coil 1H is about 1.3 Mz and does not satisfy f2 ≧ 2.05 MHz.

  Further, the coil 1H has characteristics contrary to the circuit theory that Ln> Lw> Ls, and as shown in FIG. 51, the relationship of Ln> Lw> Ls is not satisfied. Further, referring to FIG. 55, in the coil 1H, when an arbitrary metal body is close to the A surface shown in FIG. 49 (A), the decrease in the inductance Lw (μH) exceeds 15%. In other words, the decrease rate of Lw (μH) is 85% or less. Further, under the same conditions, there is a configuration in which the increase in effective series resistance Rw (Ω) is five times or more that of a single unit.

  However, as with the coil 1H, as shown in FIG. 49, even a coil having a configuration capable of inductive coupling between two identical coils may have characteristics different from those of the coil 1H. That is, there is a coil with the above-mentioned characteristics better than the coil 1H. The characteristics of the coil 1J will be described below with reference to the coil 1J which is an example of such a coil.

  FIG. 57 shows an effective series resistance Rw (Ω) of a single coil and two coils 1Ja and 1Jb inductively coupled with each other in the coil 1J configured in the same manner as FIG. 49, and the other coil is short-circuited. The figure which shows the relationship between Rw, Rs, Rn, and a frequency when the effective series resistance of one coil is set to Rn (Ω) when the effective series resistance of the coil is Rs (Ω) and the other coil is opened. It is.

  FIG. 58 shows the inductance Lw (μH) of the coil 1J alone, the inductance Ls (μH) of one coil when the two coils 1J are inductively coupled and the other coil is short-circuited, and one when the other coil is opened. It is a figure which shows the relationship between Lw, Ls, Ln, and a frequency when the inductance of the coil of this is set to Ln (microH).

  FIG. 59 shows an effective series resistance Rw (Ω) at 100 kHz and an inductance Lw (μH) when various metal plates are brought close to the A surface of FIG. 49A as shown in FIG. FIG.

Referring to FIG. 57, the maximum frequency f2 that satisfies Rs> Rn ≧ Rw is 2 MHz or more. The coupling coefficient k between the two coils at 100 kHz is ki = 0.37. Therefore, 150 / ki ² = 150 / 0.3722 = 1095 kHz. Therefore, the coil 1J satisfies f2> 1095 kHz. Referring to FIG. 58, Ln>Lw> Ls, and the relationship Lw> Ls is also satisfied. Further, referring to FIG. 59, the coil 1J has an effective series resistance Rw (when an aluminum foil having a thickness of 12 μm or an iron plate having a thickness of 0.5 mm (in this case, any thickness) is brought close at 100 kHz. Ω) increase rate is 5 times or less. In addition, the coil 1J has an inductance Lw (μH) reduction rate of 85% or more when an arbitrary metal body approaches at 100 kHz. In addition to the coil having the configuration shown in FIGS. 2 and 23 to 34, if the coil has a configuration that can be inductively coupled between two identical coils, the above measurement is performed and the characteristics are defined. A coil with good performance can be specified, and a coil with good performance can be realized.

(Explanation of relationship between coil facing distance and coupling coefficient, f1 (Hz), f2 (Hz))
From the experimental results described above, the coil 1J has a good performance as a coil, but has a low coupling coefficient and a low power factor. In addition, as shown in FIG. 55, the effective series resistance Rw (Ω) increases when the high frequency region is reached. On the other hand, the maximum frequency f2 at which the coil 1J satisfies Rs> Rn ≧ Rw is 2 MHz or more, and the coil 1Gb is far from 150 kHz, which is the maximum frequency f2 at which Rs> Rn ≧ Rw. Very expensive. Therefore, in order to lower the coupling coefficient, the distance between the two coils 1Gb was 3 mm, and the frequency characteristics of Rw, Rs, and Rn were measured.

  FIG. 60 is a diagram showing the relationship between Rw, Rs, Rn and frequency when the facing distance TK between the two coils 1Gb is 3 mm.

  As can be seen by comparing FIG. 60 with FIG. 56, the maximum frequency f2 at which the coil 1Gb satisfies Rs> Rn ≧ Rw has increased to about 500 KHz. When two coils 1Gb are used and the facing distance TK is 3 mm, the coupling coefficient kr approximately calculated from Rs and Rw at 100 kHz is 0.89, and the coupling coefficient ki approximately calculated from Ln and Ls. Is 0.91. Thus, there is no significant difference in the coupling coefficient both when the opposing distance between the two coils 1Gb is zero and when the distance between the two coils 1Gb is 3 mm. However, the f2 is greatly increased to 500 kHz.

  FIG. 61 shows the relationship between Rw, Rs, Rn and frequency when two coils 1Gd equipped with a 0.5 mm-thick aluminum metal plate are used for the coil 1Gb and the opposing distance TK is zero. FIG.

  The maximum frequency f2 at which the coil 1Gd satisfies Rs> Rn ≧ Rw is reduced to about 80 kHz.

  FIG. 62 is a diagram showing the relationship between Rw, Rs, Rn and frequency when two coils 1Gd are used and the facing distance TK is 3 mm.

  As can be seen by comparing FIG. 60 and FIG. 62, the maximum frequency f2 at which the coil 1Gd satisfies Rs> Rn ≧ Rw has increased to about 500 kHz. When two coils 1Gb not equipped with a metal plate are used and the opposing distance TK is set to 3 mm, or when two coils 1Gd are used and the opposing distance TK is set to 3 mm, f2 of both coils is the same.

  In the above-described embodiments from the coil 1A to the coil 1G, even if the facing distance TK between the coils changes, the highest frequency satisfying Rs> Rw and the highest frequency satisfying Rs> Rn ≧ Rw. As described above, there is not much change in f2 (Hz). However, in the coil 1Gb and the coil 1Gd, the maximum frequency f1 (Hz) satisfying Rs> Rw and the maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw significantly change depending on the facing distance TK. Taking the coil 1Gb, the coil 1Gc, and the coil 1Gd as an example, it is preferable to provide a facing distance of at least 2 mm between the coils. Since the facing distance is a function of the coil outer diameter D, it is preferable that the coil outer diameter is D and the facing distance TK is TK ≧ D / 50. The facing distance TK is shown in FIGS. 36 to 38 and is the distance between the conductor ends.

  The same applies to the coil 1q shown in FIG. 31 in which the two magnetic material plates 511 and 512 are mounted on the coil 50 and the metal plate 55 is provided on the magnetic material plate 512 side. Further, in the coil 1q shown in FIG. 31, when the opposing distance TK becomes zero, a magnetic material plate having a high maximum frequency f2 (Hz) that satisfies Rs> Rn ≧ Rw is selected. In particular, in the coil 1q, the two magnetic material plates 511 and 512 are configured to have different thicknesses and materials, and magnetic material plates having different relative permeability are used so that f2 (Hz) is as high as possible. preferable. Since the magnetic material plates 511 and 512 are actually sheets obtained by solidifying magnetic material powder with a binder, it is difficult to actually measure the relative permeability. As long as the manufacturer's materials are referred to, f2 (Hz) decreases when the relative permeability of the magnetic material plate 511 near the coil 50 is lowered and the relative permeability of the magnetic material plate 512 away from the coil 50 is increased. The rate seems to be small. However, the magnetic material plates 511 and 512 have no established standards such as the relative dielectric constant of the insulator 52, and are only for reference. The configuration rules described in the embodiments of the present invention should be prioritized.

  Normally, a coil of a power transmission device configured by winding a conducting wire is not used when the opposing distance TK of both coils is zero, and requires a predetermined distance, for example, an interval of 3 mm as described above. . As described above, the coupling coefficient of the coil 1f shown in FIG. 38 to the coil 1t shown in FIG. 34 does not decrease even when the facing distance TK is provided. Therefore, it is possible to realize a power transmission coil that can maintain a high power factor and has good power transmission performance. By using this power transmission coil, a power transmission device with good power transmission performance can be realized. As described above, the coil 1f shown in FIG. 38 to the coil 1t shown in FIG. 34 can avoid characteristic fluctuation due to the proximity of the metal body. Moreover, unnecessary radiation can be reduced. Furthermore, even if the opposing distance TK is provided between the two coils, there is an extremely excellent effect that the coupling coefficient does not decrease.

  Note that the coil 1Ga, which is an example of the coil 1k shown in FIG. 27, has a maximum frequency f1 satisfying Rs> Rw of 10 MHz or more and a maximum frequency f2 satisfying Rs> Rn ≧ Rw of 2 MHz or more. Since there is no influence of the magnetic plate and the effects other than the prevention of the change of the coil characteristics due to the proximity of the metal body are the same as those of the coils 1A to 1G, the description on the opposing distance TK of the opposing coils is omitted.

  Needless to say, the coil 50 from the coil 1f to the coil 1t has an effective series resistance of one coil alone in the air-core state as Rw (Ω), and the other coil facing the one coil in the air-core state. Assuming that the effective series resistance of one coil when short-circuited is Rs (Ω), one coil that satisfies Rs> Rw at 100 kHz is used. For example, as the coil 50, the coils 1B to 1G which are the embodiments of the present invention described above are used.

  Further, when the coil 1f to the coil 1t equipped with the magnetic material plate 51 or 511, 512 are used as one coil, it is preferable that the coil 1f to the coil 1t satisfy Rs> Rw at 100 kHz.

  Assuming that the highest frequency satisfying Rs> Rw from the coil 1f to the coil 1t is f1 (Hz), the power transmission device 100 equipped with the coil 1t to the coil 1t transmits power at a frequency less than f1 (Hz). .

When the coil 1f to the coil 1t are the power transmission coil 1 of the power transmission device 100, the coil 1f to the coil 1t are fd (H) having a frequency less than f1 (Hz) by the AC power supply 30b.
z).

  Furthermore, when the coil 1m and the coil 1q of the above-described embodiment are one coil, the same coil is the other coil, the inductance of one coil is Lwa (H), and both coils are inductively coupled. If the inductance of one coil when the other coil is short-circuited is Lsa (H), the coil 1m and the coil 1q satisfy Lwa> Lsa at 100 kHz, and f1 ( It is preferable that Lwa> Lsa is satisfied in a frequency region less than (Hz) and a frequency used for power transmission.

  The maximum frequency f2a satisfying Rna (Ω) and Rsa> Rna ≧ Rwa as the effective series resistance of each coil when the coil 1t to the coil 1t described above are used as one coil and the other opposing coil is opened. Assuming (Hz), one coil is preferably used in a frequency region below f2a (Hz).

  When the coil 1m and the coil 1q described above are set as one coil and the other coil facing each other is opened, the inductance of one coil is Ln, and at least the relationship of Ln> Lw> Ls is 100 kHz. In the frequency region below f1 (Hz), the coil 1m and the coil 1q satisfy the relationship Ln> Lw> Ls at the frequency fa (Hz) used for power transmission. It is preferable.

  More preferably, assuming that the maximum frequency f2 (Hz) satisfying Rs> Rn ≧ Rw from the coil 1f to the coil 1t, the coil 1f to the coil 1t are used in a frequency region less than f2 (Hz), and f2 ( In the frequency region below (Hz), the relationship of Ln> Lw> Ls is satisfied.

  The above-mentioned definition of the thermal condition can be satisfied by obtaining the thermal resistance θi by the same method as that described above.

  When the power transmission unit 30 of the power transmission device 100 includes the coil 1f to the coil 1t, the power transmission unit 30 including the coil 1f to the coil 1t is a power transmission device of the power transmission device of the present invention.

  When the power reception unit 40 of the power transmission device 100 includes the coil 1f to the coil 1t, the power reception unit 40 including the coil 1f to the coil 1t is a power reception device of the power transmission device of the present invention.

(Explanation when connecting the metal plate to the coil center wire)
FIG. 63 is a diagram in which a metal plate mounted on a coil is used as a wire to be taken out from the inner circumference of the coil 1k shown in FIG. 27 to the coil 1t shown in FIG.

  In the coil 50 having the configuration of the coil 1a, the conducting wire taken out from the center of the coil 50 is thicker by the thickness of the conducting wire. In FIG. 63, a conductive wire through hole is provided at the center of the insulating plate 54 of the coil 1k shown in FIG. 27, and the conductive wire 501 at the inner periphery of the coil is connected to the metal plate 55. The end portion 552 of the outer peripheral portion of the coil 50 is used as one terminal, and the end portion 551 of the metal plate 55 is used as the other terminal. Various methods such as soldering and welding can be used as a method of connecting the conductive wire 501 and the metal plate 55. This configuration is applicable not only to an insulating plate but also to a magnetic material plate.

(Example in which a coupling wire is provided in a coil)
In FIG. 64, a litz wire is used for the conductive wires used in the coils having the configurations shown in FIGS. 23 to 34, one end of the litz wire is connected to all the strands, and the other end of the litz wire is connected to the litz wire. FIG. 5 is an equivalent circuit diagram when at least one of the constituent wires is taken out as a coupling line 10a.

  The coupling wire 10a shown in FIG. 64 can be used to detect the operating state of the coil. Alternatively, it can be used for signal transmission. In addition, self-excited oscillation can be performed by applying positive feedback using an inverting amplifier.

  The coupling line 10a is in a substantially tightly coupled state with the power transmission coil 1 or the power reception coil 2. The winding ratio is 1: 1. Therefore, an AC voltage having the same amplitude and phase as that of the power transmission coil 1 or the power reception coil 2 appears in the coupling line 10a. The coupling line 10a can be used for a signal transmission function between the power transmission unit 30 and the power reception unit 40, detection when a metal body is close to the power transmission coil 1, and determination when a power reception coil to which a load is connected is close.

  When the embodiment of FIG. 63 described above is applied to a litz wire, all the wires of the inner periphery are connected to the metal plate 55 and at least one of the wires taken out from the outer periphery is removed. The connecting line 10a is used. The coupling wire 10a may be taken out as a strand without being connected to the common line, and may be a four-terminal coil. In this case, since the power transmission coil and the coupling line are insulated, the degree of freedom in circuit configuration increases when used for signal detection, self-excited oscillation, and the like.

(Example of coil for power transmission)
A coil 1g shown in FIG. 24 to a coil 1t shown in FIG. 34 are also examples of the power transmission coil used in the power transmission device described above.

(Power transmission coil drive conditions)
Whether it is an air-core coil or a magnetic material plate, the above-described f1 (Hz) and f2 (Hz) are increased so that the power transmission coil is less than f1 (Hz) or f2 (Hz). A power transmission device with good performance cannot be realized unless the driving conditions of the coil to be driven are defined.

(Example of a single coil)
Furthermore, the coil used in the power transmission device according to the embodiment of the present invention in which the characteristic is defined as described above is also an invention of a coil having a configuration capable of inductively coupling between the same two coils. .

(Explanation regarding metals used in the present invention)
As described above, when a metal body is close to the air-core coil, the performance of the coil is deteriorated. Further, even when a magnetic material is brought close to the air-core coil, the magnetic material sometimes deteriorates the power transmission performance of the coil. For example, in the embodiment of FIG. 31A, there is a case where a magnetic material having a low magnetic permeability is provided in a bobbin-shaped inner diameter cavity. As described above, even if the coil is equipped with a magnetic material, the performance of the coil in the present invention cannot be improved regardless of the configuration regulations and characteristic regulations other than the embodiments of the present invention described above.

  Even when the coil is equipped with a magnetic material, the air core coil of the present invention must first be selected. In the embodiment described above, the coil 1G, which is the air core coil having the best performance, is selected and the magnetic material plate is provided. However, as described above, it has been explained that it is not easy to eliminate the proximity effect of the metal body while maintaining the power transmission performance of the coil 1G. In the coil shown in the example of Patent Document 2, the influence of proximity to the metal body cannot be eliminated, and operation at a high frequency is difficult. Due to unnecessary radiation, it is difficult to use a frequency of 250 kHz or higher. However, at a minimum, operation at 100 kHz must be realized. For this purpose, it is necessary to select an air-core coil with the best possible performance based on the above-mentioned regulations.

In the embodiment of the present invention, the material of the conductor forming the conducting wire is not particularly limited, but all the coils described in this embodiment use copper as the conductor. Although copper having a small specific resistance is preferably used as the conductor, other metals or alloys having a small specific resistance can also be used as the conductor.

  In addition to diamagnetism, paramagnetism, and ferromagnetism, the magnetic properties of metals include antiferromagnetism. However, attention is focused on the magnetic property that increases the effective series resistance Rw (Ω) of the coil. The metal plate mounted on the coil may be anything other than a metal or alloy that is simply attracted to the permanent magnet.

  The inventor of the present application made measurements using various metals such as titanium, brass, and stainless steel. The 0.1 mm thick titanium plate showed the same characteristic variation as the 0.1 mm thick aluminum plate. A brass plate with a thickness of 0.5 mm showed the same characteristic variation as a copper plate with a thickness of 0.5 mm. The stainless steel plate adsorbed on the 0.5 mm thick permanent magnet caused characteristic deterioration more than the 0.5 mm thick iron plate. Thus, the metal plate may be selected depending on whether it is attracted to the permanent magnet or not.

(Measurement device used for measuring coil characteristics)
The effective series resistance and inductance of each coil described above were measured up to 1 MHz using an Agilent LCR meter, 4284A, and 1-10 MHz using a Hewlett Packard LCR meter, 4275A. . In addition, since measurement of 1 to 10 MHz can be performed only at each point of 1, 2, 4, and 10 MHz, for example, when Rs> Rw is satisfied at 4 MHz and Rs> Rw is not satisfied at 10 MHz. Estimates the maximum frequency f1 (Hz) satisfying Rs> Rw by interpolation.

(Power transmission characteristics of power transmission equipment)
FIG. 65 shows the relationship between the load current of the secondary winding and the voltage at both ends of the secondary winding in a normal transformer (transformer) configured in a tightly coupled state, and the power transmission device 100 according to the embodiment of the present invention. It is a characteristic view which shows the relationship between the load current of the receiving coil 2 of this, and the both-ends voltage of a receiving coil. In FIG. 65, the characteristic of a general transformer is indicated by a solid line, and the characteristic of the power transmission device in the embodiment of the present invention is indicated by a broken line.

  In a normal transformer configured in a tightly coupled state, if the load current value of the secondary winding is equal to or less than the rated value Im (A) of the transformer, the voltage across the secondary winding is substantially constant. General electric devices and electronic devices operate at a constant voltage. Therefore, it is used in a constant voltage region below the rated value Im (A) of the transformer shown in FIG. However, unlike a transformer used for a commercial power source, a transformer used for a general electric device or electronic device has a slightly higher rated value Im (A) than the maximum current consumed by the device. Things are used. This is because if an excessive margin is provided, the volume of the transformer increases and the cost also increases.

  On the other hand, when looking at the relationship between the load current of the receiving coil of the power transmission device and the voltage across the secondary winding in the embodiment of the present invention, the voltage across the secondary winding drops as the load current increases. I can see that This characteristic diagram is calculated by reading numerical values from FIG. 8 of Patent Document 2 and performing normalization after slight correction. In the follow-up test by the inventor of the present application, the voltage drop rate across the secondary winding due to the increase in the load current is larger than that of Patent Document 2.

  In the power transmission device according to the embodiment of the present invention having such characteristics, when the load is set to the maximum current required by the device, the power supply voltage decreases. When the current consumed by the device decreases, the power supply voltage increases. Furthermore, as shown in FIGS. 36 to 39 described above, the dimensions and shape of the power transmission coil and the power reception coil may be different, or the facing distance TK between the coils may be different depending on the use situation. In such a case, it is assumed that adjustment is performed so as to ensure the load current of the power receiving unit at the maximum facing distance TK between the coils in the state of FIG. Needless to say with the above-mentioned equation (4), when the relative position of the coil is as shown in FIG. 36 or the facing distance TK between the coils is shortened, it is easy for the voltage on the power receiving side to rise. Can be guessed. That is, in the power transmission device of the present invention, the receiving coil output does not have a constant voltage characteristic. When the voltage on the power receiving side increases, the device main body including the power receiving unit 40 may be damaged. In addition, an excessive current may flow through the power receiving unit. Although the load resistance values are different, it can be seen from FIG. 65 that when the load current is reduced from 2 A to 1 A, the load voltage becomes about 1.6 times. At least these countermeasures against excessive voltage and excessive current must be taken into consideration.

  Further, as described above, the power transmission device, the power transmission device of the power transmission device, and the power reception device of the power transmission device in the embodiment of the present invention have extremely good power transmission performance. Although depending on the current flowing through the power transmission coil 1 and the power reception coil 2, by using the coil 1G having a diameter of 5 cm described above for both the power transmission coil 1 and the power reception coil 2, it is possible to transmit power of up to about 40 W. . When such power transmission performance is achieved, the power transmission coil 1 has a metal heating action, similar to the overheating coil of the induction heater. Therefore, as described in Patent Document 5, when a metal body such as a clip approaches the power transmission coil, heat generation of the metal body must be prevented. Alternatively, it is necessary to determine a proper power receiving unit. Therefore, specific examples for solving these problems will be described below.

(Description of the relationship between the inductance of the power transmission coil 1 and the load resistance value)
Next, the inductance of the power transmission coil 1 in the case where the power factor is improved using the two-terminal circuit shown in FIG. 116 using a capacitor will be described.

  FIG. 66 is an equivalent circuit diagram in the case where the power transmitting coil 1 and the power receiving coil 2 are inductively coupled, and both coils constitute a transformer. The load resistance RL can be changed.

  FIG. 67 is a diagram showing a change in inductance on the primary side (power transmission side) when an air-core coil is used for the power transmission coil 1 and the power reception coil 2 in the equivalent circuit shown in FIG. The resistance value (relative value) and the Y axis represent inductance (relative value).

  In FIG. 67, Lw represents the inductance of the power transmission coil 1 alone at 100 kHz. That is, Lw is the inductance of the power transmission coil 1 when the power transmission coil 1 and the power reception coil 2 are separated. Ls represents the inductance of the power transmission coil 1 at 100 kHz when the power reception coil 2 whose both ends are short-circuited is inductively coupled to the power transmission coil 1. Ln represents the inductance of the power transmission coil 1 at 100 kHz when the power receiving coil 2 having both ends opened is inductively coupled to the power transmission coil 1. Lx represents the inductance of the power transmission coil 1 corresponding to the value of the load resistance value connected to the power reception coil 2 at 100 kHz. Hereinafter, Lw, Ls, Ln, and Lx are defined similarly in the characteristic diagrams of other coils.

  FIG. 68 is a diagram showing an example of a change in inductance of the power transmission coil 1 when a coil equipped with a magnetic material is used for at least one of the power transmission coil 1 or the power reception coil 2 in the equivalent circuit shown in FIG. The X axis represents the load resistance value (relative value), and the Y axis represents the inductance (relative value).

  FIG. 69 is a diagram showing another example of a change in inductance of the power transmission coil 1 when a coil equipped with a magnetic material is used in at least one of the power transmission coil 1 or the power reception coil 2 in the equivalent circuit shown in FIG. Yes, the X axis represents the load resistance value (relative value), and the Y axis represents the inductance (relative value).

  In FIG. 67, the relationship between the inductances Lw, Ls, Ln, and Lx is Ln = Lw> Ls. In FIG. 68, Ln> Lw> Ls. In FIG. 69, Ln> Ls> Lw.

  66 to 69 show the initial values obtained by actually measuring the inductances Lw, Ls, and Ln at 100 kHz, and the inductance Lx is obtained by changing the numerical value of the load resistance RL by computer simulation. The inductance Lx is also a function of the coupling coefficient k between the power transmission coil 1 and the power reception coil 2. That is, the inductance Lx is a function having at least two independent variables represented by Lx = f (RL, k). Therefore, first, a partial differential equation is established based on Lx = f (RL, k), and it is confirmed that a solution exists in the equation. Thereafter, inductances Lw, Ls, and Ln are actually measured, and an initial condition (integral constant) is obtained and simulated, and complicated calculations are performed. Therefore, description of the details of the simulation related to FIGS. 66 to 69 is omitted. However, the inductances Ls and Ln can be measured, and Ls is an inductance value of the power transmission coil 1 when RL≈0Ω and Ln is infinite (the power receiving coil 2 is in an open state). Therefore, it is obvious that the inductance Lx exists only between Ls and Ln.

  As described above, in a power transmission device in which a power transmission unit and a power reception unit are separable, a resonance circuit is formed by the power transmission unit alone and the power reception unit alone by providing a power factor improving capacitor. In Patent Document 4 to Patent Document 6 and many other prior documents, the relationship between the resonance frequency fh (Hz) of the power transmission unit alone and the resonance frequency fk (Hz) of the power reception unit alone is defined. However, the power transmission device in which the power transmission unit and the power reception unit can be separated must cancel the residual reactance Le (H) of the power transmission coil 1 when transmitting power to improve the power factor.

  66 to 69, in FIG. 66, it can be seen that the inductance of the power transmission coil 1 changes depending on the value of the load resistance RL. In Patent Document 4 to Patent Document 6 and other prior documents, there are some that define the relationship between the resonance frequencies fh and fk as fh> fk, and some that specify fh <fk. It is guessed that the relationship between FIG. 66 and FIG. 69 was not known.

  In practical use of the power transmission device using the power factor improving capacitor, a problem arises in the circuit configuration of FIG. 66 when Lx = Lw in FIG. If the power factor improving capacitor is used in such a case, the frequency fh (Hz) at which the reactance is zero in the power transmission coil 1 alone and the frequency fd (Hz) during power transmission are equal. Therefore, when the power reception unit 40 is not attached to the power transmission unit 30, the power transmission coil 1 is a simple series resonance circuit. When the output voltage of the AC power supply 30b is Vt (V) and the sum of the effective series resistances of the power transmission coil 1 and the power factor improving capacitor C1 is Rm (Ω), an excessive current of Vt / Rm (A) is transmitted. The current flows through the coil 1 and the power factor improving capacitor C1, and the power transmission coil 1 generates heat and is damaged. Such a state can be avoided by changing the relative position of the power transmission coil 1 and the power reception coil 2 and changing the coupling coefficient between the two coils. However, when the value of the load RL fluctuates and there is a region where Lx = Lw, the transmission power must be reduced by the power transmission unit 30 or the output frequency fa (Hz) of the AC power supply 30b must be changed. . When using a power factor improving capacitor, it is necessary to pay attention to the above relationship.

  70 is a diagram illustrating an example of the AC power supply 30b included in the power transmission control circuit 11 illustrated in FIG. In FIG. 70, AC power supply 30b includes a control circuit 30c and switching elements Q1 and Q2. The control circuit 30c is supplied with a DC voltage Vd (V) from the DC power supply 12, and the control circuit 30c alternately applies control signals to the gates of the switching elements Q1 and Q2. A DC voltage Vd (V) is supplied to the drain of the switching element Q1, and the source of the switching element Q1 and the drain of the switching element Q2 are commonly connected to one electrode of the capacitor C1. The source of the switching element Q2 is connected to GND (ground).

  The control circuit 30c alternately turns on the switching elements Q1 and Q2, and supplies a square wave that is a non-sinusoidal wave to the power transmission coil 1 via the capacitor C1. This square wave is a waveform having a duty of 50% and an amplitude Vd (V), the level of which changes between Vd (V) and GND. Since the capacitor C1 and the power transmission coil 1 constitute an LC series resonance circuit such as the equivalent circuit of FIG. 116 including the power reception unit not shown in FIG. 1, the capacitor C1 and the power transmission coil 1 resonate based on the square wave signal. Then, a sine wave alternating current with an amplitude of VL (V) is transmitted from the power transmission coil 1 to the power reception coil 2 shown in FIG. 1 and supplied to the load RL. As will be described later, the amplitude VL (V) is boosted several to several tens of times higher than the amplitude Vd (V).

(Description of LC series resonance circuit)
FIG. 71 is a circuit diagram for measuring the characteristics of the LC series resonance circuit. In FIG. 71, a reference coil Ls, a measurement capacitor Cx, and a resistor R2 are connected between the AC power output terminal of the AC power supply 30b and GND. An oscilloscope 80 is connected to both ends of the measurement capacitor Cx, and the voltage across the capacitor Cx is measured. R2 is a resistance of about 0.1Ω and measures the AC current flowing through the LC series resonance circuit by measuring the voltage across R2, and the AC power supply 30b has a reactance Xc of the capacitor Cx to be evaluated and a reference coil The output frequency is set so that reactance Xi of Ls alone is equal.

  72 is an equivalent circuit diagram including the pure resistance component (effective series resistance) of each element constituting the circuit for measuring the characteristics of the LC series resonance circuit of FIG. 72, Rc (Ω) is the effective series resistance of the capacitor C, Ri is the effective series resistance of the power transmission coil 1, and Rm (Ω) is the effective series resistance Ri of the coil 1 and the effective series resistance Rc of the capacitor C. The added value (Ri + Rc) (Ω), and the output impedance Zs (Ω) indicates the output impedance of the AC power supply 30b.

(Description of capacitor)
First, the inventor of the present application uses the same AC power source 30b, which is a component other than the capacitor, and the power transmission coil 1, the power reception coil 2, and the load RL in the circuit of FIG. I tried the test. As a result, it has been found that even when capacitors having the same capacitance are used, the power transmission performance differs depending on the capacitor dielectric and configuration. Moreover, even if it was the capacitor comprised by the same dielectric material, it discovered that electric power transmission performance differed with the structure of a capacitor. Furthermore, even if the power transmission coil 1 is changed and the configuration is exactly the same as the dielectric of the capacitor, it has been found that the power transmission performance varies depending on the capacitance.

  Therefore, the present inventor prepared 17 types of capacitors C1a to C1s of 0.01 μF, and first measured the characteristics of the capacitors with an LCR meter. C1a and C1b are polystyrene (PS), C1c and C1g are polypropylene (PP), C1d is polycarbonate (PC), C1e and C1f are polyphenylene sulfide (PPS), C1j and C1q are polyethylene (PE), C1h, C1m, C1r, C1s is made of ceramic (CE), C1i, C1k, C1n, and C1p are made of polyethylene terephthalate (PET) as dielectrics.

  Table 2 shows the characteristics of these capacitors rearranged in descending order of the effective series resistance Rc at 100 kHz. In addition, about each capacitor shown in Table 2, about 5 each is measured, an average value is calculated | required, and the characteristic value of the capacitor close | similar to an average value is described. However, capacitors that do not serve as basic data, such as capacitors whose characteristics cannot be measured, capacitors whose measurement values are not reproducible, and capacitors whose power transmission performance is extremely poor, are excluded from Table 2. Further, the above-described polystyrene capacitors that cannot be used due to heat generation and polypropylene capacitors that have extremely poor power transmission performance are also excluded from Table 2.

  In Table 2, C represents the capacitance of the capacitor, and its unit is μF. Rc represents the effective series resistance of the capacitor at each frequency, Xc represents the reactance of the capacitor at each frequency, and the unit is Ω. Q represents the Q of the capacitor at each frequency and is unitless. tan δ represents the dielectric loss tangent of the capacitor at each frequency and is unitless.

  In addition, although not described in Table 2, the power transmission performance is also measured for a capacitor using polyethylene naphthalate (PEN) as a dielectric. The power transmission performance of the polyethylene naphthalate capacitor will be described later.

  In paragraph No. 0013 of Patent Document 1, it is described that 50 kHz to 500 kHz is preferable as the frequency for power transmission. However, the characteristics of the capacitor in the high frequency region vary depending on the capacitor. Furthermore, even if the capacitor has exactly the same configuration using the same dielectric, the power transmission performance varies depending on the capacitance.

  Regarding the frequency characteristics of these capacitors, usable frequency regions will be described later with examples. Using Table 2 as basic data, the characteristics of the capacitor will be discussed. First, the characteristics of the capacitor related to the embodiment of the present invention defined in JIS are listed in Table 3.

(Principle of capacitor characteristic measurement)
First, the principle of measuring capacitor characteristics will be described. FIG. 73 is an example of a principle circuit diagram of a measurement circuit that measures the capacitance of a capacitor, and is a principle diagram showing an example of a capacitance measurement method in an LCR meter.

  The LCR meter having the circuit configuration of FIG. 73 includes an AC constant current source 41 and an AC voltmeter 42. In FIG. 73, Rz is a contact resistance when a probe or the like is brought into electrical contact with the terminal of the capacitor C. The capacitance of the capacitor C is measured by flowing an AC constant current I (A) from the AC constant current source 41 to the capacitor C and measuring the voltage V (V) across the capacitor C with the AC voltmeter 42. A so-called four-terminal measurement method is used so that the voltage V (V) across the capacitor C can be accurately measured. This is to eliminate the influence of the contact resistance Rz (Ω) shown in FIG. 73 when measuring a minute current or minute voltage.

  Although not shown, the LCR meter shown in FIG. 73 includes means for detecting a phase difference θ between the phase of the AC constant current source 41 and the voltage phase measured by the AC voltmeter 42. Therefore, the LCR meter shown in FIG. 73 can measure complex impedance.

Since the reactance Xc (Ω) of the capacitor is impedance Z (Ω),
Xc (Ω) = Z (Ω) = V (V) / I (A). When the angular frequency for measuring the capacitance C (F) of the capacitor C is ω, the capacitance C (F) is obtained as C (F) = Xc (Ω) / ω. Actually, Z is a complex impedance, and the phase of the current I (A) is advanced by θ (0 ° ≦ θ ≦ 90 °) from the phase of the voltage V (V). Therefore, the reactance Xc (Ω) of the capacitor is obtained as Xc (Ω) = Z · sin θ (Ω). The effective series resistance Rc (Ω) of the capacitor is obtained as Rc (Ω) = Z · cos θ (Ω). Therefore, C (F) = Xc (Ω) / ω = Z (Ω) · sin θ / ω. As apparent from the above equation, the Q of the capacitor is Q = Xc (Ω) / Rc (Ω), so that Q = (Z · sin θ / Z · cos θ) = tan θ. Q is a function of only the phase difference, θ, between the current I and the voltage V, regardless of the numerical values of the voltage V and the current I. The phase difference θ is obtained as θ = tan−1 (Q).

(Explanation of time change of capacitance)
FIG. 74 shows the fluctuation of the capacitance for 5 minutes from the measurement start time when the capacitance of the polyethylene terephthalate capacitor C1n and the ceramic capacitor C1r having poor power transmission performance is measured with the LCR meter shown in FIG. It is a graph to show.

  When the inventor of the present application measured the capacitance of a capacitor with poor power transmission performance with the LCR meter of FIG. 73, the capacitance display was not stable, and it took several minutes for the display to be constant. . Also in the LC series circuits of FIGS. 70 and 71, an alternating current Ic (A) flows through the capacitor. Accordingly, it takes a considerable time from the moment when Ic flows until the capacitance of the capacitor is stabilized. This indicates that in the circuits of FIGS. 70 and 71, the frequency at which the reactance becomes zero varies with time. That is, when the polyethylene terephthalate capacitor C1n and the ceramic capacitor C1r shown in Table 2 are used in the circuits of FIGS. 70 and 71, the circuit cannot be stably operated when the current Ic (A) fluctuates. This can be confirmed by measuring the capacitance with a general LCR meter having several digits of resolution and seeing the time until the capacitance value is stabilized, without actually confirming with the circuits of FIGS. It is easy to understand.

  From the actual measurement results, a capacitor that is not stable in about 5 seconds or less has poor power transmission performance. That is, it is necessary that the measured numerical value after 5 seconds from the start of measurement does not vary by ± 0.1% or more. Alternatively, it is necessary that the variation rate of capacitance is 1% or less in a measurement time of 5 minutes, and the variation rate of capacitance is 0.2% or less in a measurement time of 1 minute from the start of measurement.

  In the LCR meter having the circuit configuration of FIG. 73, as described above, a so-called four-terminal measurement method is used. Even if the capacitance is measured using such a minute current or minute voltage, the capacitance cannot be measured stably. Therefore, as shown in FIG. 74, a capacitor whose measured capacitance is not stable is not stable even in the circuits of FIGS. 70 and 71, power transmission performance is poor, and power transmission performance varies. To do.

  It is generally known that the temperature characteristics of a ferroelectric substance, for example, some ceramic capacitors, are very poor, and if the temperature is about 70 ° C., the capacitance may be about half. Such a capacitor having the characteristic that the capacitance greatly varies depending on the temperature cannot be used in the present invention. However, even a capacitor using ceramic as a dielectric has a good capacitance temperature characteristic. As a guide, the capacitance at 25 ° C. is used as a reference, and the temperature change of the capacitance between 0 ° C. and 85 ° C. is ± 5% or less. .

(Explanation of boost ratio)
Next, the operation of the basic circuit constituting the series resonant circuit shown in FIG. 72 will be described. In FIG. 72, the current I (A) flowing through the circuit becomes the maximum value Ir (A) at the frequency ω where ωL (Ω) = (1 / ωC) (Ω). This is a state where the capacitor C1 and the residual inductance Le are both short-circuited in FIG.

  Henceforth, when expressing with a resonance frequency or a resonance point, it shall refer to the frequency used as (omega) L (ohm) = (1 / omegaC) (ohm). That is, in the series resonance circuit, the reactance Xi (Ω) of the coil and the reactance Xc (Ω) of the capacitor are frequencies fr where Xi (Ω) = Xc (Ω). The inventor of the present application used the same AC power source 30b and the power transmission coil 1, prepared various capacitors, and measured the maximum value Ir (A). In FIG. 72, a power transmission coil 1 configured in a plane air-core shape is used. The output impedance Zs of the AC power supply 30b at 200 kHz is 0.2Ω. The open output voltage Vo (V) of the AC power supply 30b is fixed at an effective value of 1 V (2 Vp-p). The output impedance Zs (Ω) of the AC power supply 30b is the output voltage of the AC power supply 30b when the peak value of the open output voltage of the AC power supply 30b is Vo (V) and a non-inductive resistance of 2Ω is connected to the output of the AC power supply 30b. The peak value of V is determined as Vt (V), and Zs (Ω) = 2 (Ω) × (Vo−Vt) (V) / Vo (V).

Similarly, the effective series resistance Ri of the power transmission coil 1 alone at 200 kHz is 1.35Ω, the inductance is about 60 μH, and the reactance Xi is 80Ω. Therefore, when the effective series resistance of the capacitor is Rc (Ω), the maximum value Ir (A) of the current I (A) flowing through the circuit is
Ir (A) = Vo (V) / (Zs (Ω) + Rw (Ω) + Rc (Ω)). And the both-ends voltage Vi (V) of the power transmission coil 1 becomes Vi (V) = Xi (Ω) × Ir (A). That is, if the output impedance Zs (Ω) of the AC power supply 30b is sufficiently low and the effective series resistance Rc (Ω) of the capacitor satisfies Rc (Ω) << Ri (Ω), the power transmission coil 1 If the single Q is Qi, both ends of the power transmission coil 1 are
A voltage of Vi (V) = Vo (V) × Qi is generated. This Qi is called a boost ratio H. Actually, considering the Q and Qc of the capacitor, the Q and Qr of the series resonant circuit are
1 / Qr = 1 / Qc + 1 / Qi. When Qc >> Qi is not established, the step-up ratio H is H = Qr. As described above, at the frequency where ωL1 (Ω) = 1 / ωC1 (Ω), the reactance Xi (Ω) of the coil and the reactance Xc (Ω) of the capacitor C are equal. Therefore, the voltage Vc (V) across the capacitor becomes equal to Vi (V). The inventor of the present application uses various capacitors having a nominal value of 0.01 μF shown in Table 1, and calculates the voltage Vc (V) across the capacitor and the step-up ratio H, H = Vc (V) / Vo (V) described above. I measured it with an oscilloscope 70.

  FIG. 75 is a graph with the effective series resistance Rc of each capacitor at 200 kHz shown in Table 2 as the X axis and the step-up ratio H and the drive circuit current Id (mA) as the Y axis. The black dots on the characteristics shown in FIG. 75 indicate the actual measurement value H of the boost ratio and the drive circuit current ID in each capacitor. In FIG. 75, Ht is a theoretical value of the step-up ratio calculated from Qr. In addition, a black point is a measured value of each capacitor, and an intermediate point is interpolated by a graph.

  FIG. 76 is a circuit diagram in which a resistor R3 is added in series to the circuit of FIG. The action of R3 shown in FIG. 76 will be described later. The output impedance Zs (Ω) of the AC power supply 30b is as extremely low as 0.2Ω. However, in this embodiment, the effective series resistance Ri (Ω) of the coil and the effective series resistance Rc (Ω) of the capacitor are 2Ω or less, and Zs cannot be ignored. Therefore, the output voltage of the AC power source connected to the LC series circuit is set to Vt (V) so that Zs need not be considered. Vt (V) is smaller than Vo (V), but if Vt (V) is measured, the current flowing through the LC series circuit can be calculated regardless of Zs (Ω). Hereinafter, Vt (V) and Vo (V) will be properly used according to the embodiment.

  From FIG. 75, it can be seen that the boost ratio H differs depending on the capacitor. At the series resonance point, the reactance component becomes zero and only the effective series resistance Rm (Ω) which is a pure resistance component. Therefore, a current Ir (A) of Ir (A) = Vt (V) / Rm (Ω) flows through the circuit.

Therefore, the voltage Vi (V) generated at both ends of the coil is
Vi (V) = Ir (A) × Xi (Ω).

The voltage Vc (V) generated across the capacitor is
Vc (V) = Ir (A) × Xc (Ω).

  Since Xi (Ω) = Xc (Ω) at the series resonance point, Vi (V) = Vc (V). Since Vi (V) and Vc (V) are both 180 degrees out of phase and have the same amplitude, the voltage between point A and point B in FIG. 76 is theoretically zero. That is, the point A and the point B are in the same state as a short circuit. Actually, since there is an effective series resistance Rc (Ω) of the capacitor and an effective series resistance Ri (Ω) of the coil, a sine wave residual voltage is generated between the points A and B.

Current Ir (A) = Vo (V) / (Zs (Ω) + Ri (Ω) + Rc (Ω)),
Substituting Zs = 0.2Ω and Ri = 1.35Ω,
Zs + Ri + Rc = Rc + 1.55Ω, and if Rc≈0Ω, the circuit current Ir (A) at the resonance frequency is theoretically Ir = 1V / 1.55Ω = 666 mA.

  In the circuit of FIG. 71, R2 is a resistance for AC current measurement of 0.1Ω, and a polystyrene capacitor C1a having an effective series resistance Rc (Ω) of Table 2 of 0.01Ω is used, and an oscilloscope 80 is used as shown in FIG. The maximum value of the voltage across R2 was 83 mV. Therefore, the peak value, 83 mV / 0.1Ω = 830 mA, flows through the circuit of FIG. That is, an effective current of 830 / √2 = 586 mA is flowing, and a result almost as expected is obtained. The difference between the theoretical value 666 mA and the actually measured value 586 mA is presumed to be because the AC current measuring resistor R2 is added in series to the circuit.

The effective series resistance Rm (Ω) in the circuit of FIG. 72 is Rm = Ri + Rc = 1.55Ω. At the resonance point, the power Pr consumed by the effective series resistance Rm is
Pr = 0.586 (A) ² × 1.55 (Ω) = 0.52 W. The output voltage of the DC power supply 12 is 2 V, the output current is 0.26 A, and the output power is 2 V × 0.26 A = 0.52 W, which is a result that almost agrees with the theory.

However, some capacitors in Table 2
Ir (A) = Vo (V) / (Zs (Ω) + Ri (Ω) + Rc (Ω)), and
The relationship Ir (A) = Vt (V) / (Ri (Ω) + Rc (Ω)) = Vt (V) / (Rm (Ω)) is not satisfied. Further, the boost ratio H does not satisfy the theoretical value of the boost ratio H obtained from the relationship 1 / Qr = 1 / Qc + 1 / Qi, where Q is Q of the series resonance circuit. That is, H ≠ Qr.

For example, the ceramic capacitor C1r in Table 2 has an effective series resistance Rc of 1.57Ω at 200 kHz. Therefore, Zs + Rw + Rc is
Zs + Rw + Rc = 0.2 + 1.35 + 1.57 = 3.12Ω.

In FIG. 72, the circuit current It at the theoretical resonance point is an effective value,
It = 1V / 3.12Ω = 320 mA.

Therefore, the theoretical power consumption Pt, Pt = I2R (W) in FIG.
Pt = I ² R = 0.32A × 0.32A × 3.12Ω = 0.32W.

On the other hand, the actual measurement current of the drive circuit of FIG. 70 is 63 mA, and the power input to the circuit of FIG.
2V × 0.063A = 0.126W. That is, in theory, the circuit of FIG. 71 that should consume 0.32 W consumes only 0.126 W in actual measurement.

Further, when the circuit current I in FIG.
I = √ (P / R) = √ (0.126 W / 3.12Ω) = 200 mA.

  The theoretical value It is 320 mA, but the actual measurement value I is 200 mA.

The voltage Vc across the capacitor calculated from the theoretical value is an effective value,
Vc = 72.4Ω × 0.32A = 23.17V.

  The theoretical boost ratio Ht is Ht = 23.17 / 1 = 23.17, but the actually measured boost ratio H is only about 10.8. The step-up ratios Ht and H represent the capacitor voltage.

Summarizing the above results, the ratio between theoretical and measured values is
In current, 200 mA / 320 mA = 0.625
In voltage, 10.8 / 23.17 = 0.466
In power, 126mW / 320mW = 0.394
It becomes. Both values are smaller than the theoretical values. However, as will be described later, it may be larger than the theoretical value. This is presumably because the capacitance of the capacitor is not 0.01 μF, and from Table 1, there is a maximum deviation of about ± 10%, and the resonance frequency is shifted from 200 kHz. Using the series resonant circuit of FIG. 71, the performance of the capacitor can be determined in this way.

  In FIG. 75, theoretical boost ratios Ht and Ht = Qr are plotted. In FIG. 75, the ceramic capacitor C1h deviates from the theoretical value calculated from the effective series resistance Rc. This is a capacitor that does not match the theoretical value described above with reference to the ceramic capacitor C1r, and does not satisfy the following rules. This will be described later.

From the above experimental results, the theoretical step-up ratio Ht and the ratio of the actually measured step-up ratio H, H / Ht,
If the condition of H / Ht> 0.9 is satisfied, predetermined power transmission performance described later can be ensured.

Referring to FIG. 75, there is a capacitor where H / Ht> 1. As described above, the capacitance of these capacitors is smaller than 0.01 μF, and the resonance frequency is increased. Since the coils are the same, the reactance Xi (Ω) of the coils is increased. As a result, the reactance Xi (Ω) of the capacitor also increases. The drive circuit current IDr (A) at the resonance point is determined only by the effective series resistance Ri (Ω) of the coil and the effective series resistance Rc (Ω) of the capacitor. Therefore, since the drive circuit current IDr (A) is the same, the voltage across the capacitor,
It is assumed that Vc (V) = Xc (Ω) × IDr (A) increases.

Here, if It (A) = Vt (V) / (Ri (Ω) + Rc (Ω)), the ratio of the theoretical circuit current It to the actually measured circuit current Ir, Ir / It is
If the condition of Ir (A) / It (A)> 0.9 is satisfied, predetermined power transmission performance described later can be ensured. This is because, at the series resonance point, when the output voltage of the AC power supply 30b is Vt (V) and the voltage across the capacitor is Vc (V), the condition of Vc (V) / Vt (V)> 0.9 is satisfied. It is equivalent to being satisfied. As described above, H = Vc (V) / Vt (V). Therefore, the condition of Vc (V) / Vt (V)> 0.9 is equivalent to H> 0.9 × Qr, where H is the actually measured boost ratio and Q and Qr of the LC series circuit. . Depending on the difference in resonance frequency, H> Ht may be satisfied. However, since normally Ht> H, it is not necessary to specify the upper limit of H in particular. From the actual measurement result of FIG. 75, when H> Ht, H was about 1.2 times Ht. Note that Ht is defined because the symbol of the boost ratio H is defined, but Ht = Qr. This is the same as the theory described above.

The ratio of theoretical circuit power Pt to measured circuit power P, P / Pt,
If the condition of 0.8 <P / Pt is satisfied, predetermined power transmission performance described later can be ensured.

  The above-described correlation between H / Ht, I / It, P / Pt, and power transmission performance will be described later with examples. Next, 1 / Qr = 1 / Qc + 1 / Qi is calculated.

From Table 2, Xc = 72.4Ω at 200 kHz of the ceramic capacitor C1r, and Xc (Ω) = Xi (Ω) at the resonance point.
Qi = Xi / Ri = 72.4Ω / 1.35Ω = 53.62
Qc = Xc / Rc = 72.4Ω / 1.57Ω = 46.11
1 / Qi + 1 / Qc = 1 / 53.62 + 1 / 46.11 = 0.04
Qr = 1 / 0.041 = 24.8.
Although there is a slight difference from the theoretical boost ratio Ht = 23.17 obtained above, the theoretical boost ratio Ht obtained above takes into account the output impedance Zs of the AC power supply, and the output impedance Zs. , Ri + Rc = 1.35Ω + 1.57Ω = 2.92Ω, Ir = Vt (V) / (Ri (Ω) + Rc (Ω)) = 1V / 2.92Ω = 0.342A,
Vc = 72.4Ω × 0.342A = 24.8V, and Ht = Qr. As described above, the actual boost ratio H is 10.8, and the capacitor C1r has a theoretical correlation,
The relationship 1 / Qr = 1 / Qc + 1 / Qi is not satisfied.

  As described above, regarding the characteristics of the capacitor used in the circuit configuration of FIGS. 1 and 70, in which an alternating current flows in the capacitor itself, the difference between the theoretical value and the experimental value has been completely studied. There is no prior literature.

  From the calculation in FIG. 75, it was found that there is a capacitor whose boost ratio H matches the theoretical value, and that depending on the capacitor, the boost ratio H is different from the theoretical value Ht. Therefore, the inventor of the present application again measured the voltage across the capacitor with an oscilloscope in FIG.

(Explanation of Examples of Capacitor Performance Evaluation Method and Performance Evaluation Device)
77 is an AC voltage waveform across the polypropylene capacitor C1c of Table 2 at the resonance point of FIG. 78 is an AC voltage waveform across the ceramic capacitor C1r shown in Table 2. FIG. FIG. 79 is an AC voltage waveform across the ceramic capacitor C1r when R3 is about 30Ω in the circuit of FIG.

  77 and 78 are obtained by measuring the voltage across each capacitor with reference to GND in FIG. 71, with R2 in FIG. 71 removed. Note that the GND of the oscilloscope 80 must be connected to the GND of the AC power supply 30b as shown in FIG. This is because when a positive peak value Vp and a negative peak value Vn with respect to GND, which will be described later, are measured and a ratio of Vp and Vn, Vp / Vn, is obtained, Vp / Vn> 1. It is. When reverse connection is established, Vp / Vn <1, which is not within the numerical value specification. Also, if the GND of the oscilloscope 80 is not connected to a low impedance point, the observation waveform becomes unstable.

  As can be seen from a comparison between FIG. 77 and FIG. 78, in FIG. 77, the voltage waveform across the polypropylene capacitor C1c is almost symmetrical with respect to the zero point (GND in FIG. 71). On the other hand, in FIG. 79, the voltage waveform at both ends of the ceramic capacitor C1r is shifted in the positive direction with respect to the zero point, and has a positive and negative asymmetric waveform. That is, a positive DC component is included in the voltage waveform across the ceramic capacitor C1r. Such a phenomenon becomes more prominent when a non-inductive resistor R3 of about 30Ω is inserted in series in the circuit shown in FIG. FIG. 79 shows a waveform that is largely shifted in the positive direction with respect to the zero point. It has been experimentally confirmed that when the voltage across the polypropylene capacitor C1c is measured under the same conditions as in FIG. 79, there is very little shift in the positive direction.

  FIG. 80 is a graph with the effective series resistance Rc at 200 kHz as the X axis and the shift ratio S from the sine wave as the Y axis for each capacitor shown in Table 2. As shown in FIG. 80, there is no correlation between the effective series resistance Rc of each capacitor indicated by a black dot and the shift ratio S from the sine wave.

  FIG. 81 is a graph in which the shift ratio from the zero point of each capacitor shown in Table 2 is taken as the X axis and the Y axis is taken as the power transmission performance. 82 is a graph in which the effective series resistance Rc at 200 kHz of each capacitor shown in Table 2 is taken as the X axis and the Y axis as the power transmission performance. FIG. 83 is a graph of each capacitor shown in Table 2 with the dielectric loss tangent tan δ as the X axis and the Y axis as the power transmission performance.

  81 to 83, the power transmission performance is indicated by secondary side power P2 (W) and transmission efficiency η. The secondary-side power P2 has a constant primary-side AC voltage, the same relative positions and load resistance values of the power transmission coil 1, the power reception coil 2, and both coils, and the maximum power P2 (W ). The transmission efficiency η is an effective value of the AC voltage obtained by monitoring both ends of the DC power Pd (W) input to the power transmission side and the load resistance with an oscilloscope 80, and calculating from the pp voltage Vp (V) at both ends of the load resistance. Ve (V), Ve = Vp / 2√2 (V), the ratio η of the load power Ps (W) calculated from the above, η = PsW / PdW. The secondary power P2 is 3.25 W or more and the power transmission efficiency η is 80% or more as a reference, and the X-axis condition that satisfies this criterion is defined. In the following description, power transmission performance means secondary power P2 and transmission efficiency η.

  The above criterion is because the power loss of the power transmission device occurs due to Zs, Rc, Ri in FIG. 72, and the power loss of 1 W is the upper limit for practical use in power transmission around 4 W. When calculated from this condition, the power transmission efficiency is 75%. Moreover, although the magnitude | sizes of a coil differ, the electric-thermal energy conversion efficiency of an induction heater (electromagnetic cooker) is about 85%. Therefore, 80% transmission efficiency is defined as an intermediate value between 75% and 85%. The maximum power that can be transmitted to the secondary side is about 4.1 W. Therefore, from the 80% transmission efficiency, the secondary power lower limit is defined as 4.1 × 0.8≈3.25 W.

  When the time average value of the output voltage of the AC power supply 30b is not zero and a DC component is included, such as a square wave of zero potential and plus potential, in the circuits of FIGS. 71 and 76, both ends of the capacitor The voltage Vc (V) is a sine wave shifted to the plus side with respect to the zero potential. Further, in the circuit of FIG. 76, when R3 is set to a value larger than Xc = 80Ω, for example, about 100Ω, the Q of the series resonance circuit decreases, the waveform approaches a triangular wave, and the minimum value of the voltage waveform at both ends of the capacitor Becomes higher than zero potential.

  In practice, in FIG. 71, it is difficult to make the effective series resistance Ri (Ω) of the coil 1 and the output impedance Zs (Ω) of the AC power supply zero. Therefore, the reactance Xc (Ω) of the capacitor or the reactance Xi (Ω) of the coil 1, the effective series resistance Ri (Ω) of the coil 1, and the output impedance Zs (Ω) of the AC power source are measured in advance. Observe the waveform.

  Compared with the voltage Vc (V) applied to the capacitor, the current Ic (A) flowing through the capacitor is advanced by 90 degrees. Multiplying the instantaneous values of Vc (V) and Ic (A) and integrating for one period, the product of Vc and Ic becomes zero. That is, the capacitor which is a reactive element does not consume AC power. It is assumed that a DC voltage Vj as shown in FIG. 79 is superimposed on the current applied to the capacitor. In this case, multiply the instantaneous value of Vc and Ic and integrate for one period.

Vc = Vj + Vm · sinφ (V), Ic = cosφ (A),
Vc × Ic = (Vj + Vm · sinφ) × cosφ (W)
= Vj · cos φ + Vm · sin φ · cos φ (W)
= Vj · cosφ + Vm · (1/2) sin2φ (W)
Mathematically, if φ is an independent variable and Vj · cos φ and Vm · sin 2φ are definitely integrated from 0 to 2π, both become zero. That is, even if a DC component is superimposed on an AC voltage applied to the capacitor, the capacitor does not consume power. This is the effect that the capacitor blocks the direct current and allows only the alternating current to pass.

  Therefore, in FIG. 76, the both-ends voltage of the resistor R3 through which the direct current flows and the both-ends voltage of the power transmission coil 1 are substantially symmetrical with respect to the zero point (GND in FIG. 71). On the other hand, the voltage across the capacitor C is shifted with respect to the zero point as shown in FIGS.

Assuming the above circuit theory, the ratio S at which the AC voltage waveform Vc across the capacitor shifts with respect to the zero point is obtained as follows. First, the sum of all effective series resistances in the series circuit is Rr, the reactance of the capacitor at the resonance frequency is Xcr (Ω), the phase angle is θ (degrees), and the pp value of the voltage across the capacitor is Vp (V). Assuming that the shift value from the zero point is Vm (V), as described above,
tan ⁻¹ (Xcr / Rr) = θ (degrees),
Vm = Vcp × (½) cos θ (V).

In the above equation, reactance at 200 kHz, Xc = Xi = 80Ω, effective series resistance, Ri = 1.35Ω, output impedance of AC power supply, Zs = 0.2Ω, sum of all effective series resistances in the series circuit, Substituting Rr = 1.55Ω,
tan ⁻¹ (Xcr / Rr) = tan ⁻¹ (80 / 1.55) = 88.89 degrees Vm = Vcp × (½) cos 88.89 degrees = (Vcp / 2) · 0.0193
Vm = Vcp × 0.00968.

Therefore, when an ideal capacitor is used, assuming that the absolute value of the positive peak value of the capacitor is Vp and the absolute value of the negative peak value is Vn,
Vp = (Vp−Vn) × 1.000968 (V)
Vn = (Vp−Vn) × 0.9903 (V)
Is the theoretical value. The ratio of Vp and Vn, Vp / Vn is
Vp / Vn = 1.00968 / 0.9903 = 1.0195.

  As a precaution, the above shift value is calculated using an effective value instead of pp.

Vp = ((Vp−Vn) / 2√2) × 1.000968 (V)
Vn = ((Vp−Vn) / 2√2) × 0.9903 (V)
((Vp−Vn) / 2√2) is a constant. Therefore, the ratio of Vp (V) to Vn (V), Vp / Vn (no unit), remains the same as Vp / Vn = 1.0195.

  When the effective series resistance Rc (Ω) of the capacitor satisfies Rc << Rr, the ratio of Vp to Vn, Vp / Vn, is Vp / Vn <1, taking into account significant figures and measurement errors. .02 should be satisfied. In Table 1 described above, only some capacitors such as C1c and C1f satisfy Vp / Vn <1.02.

On the other hand, when Rc> 0.1Ω, the effect of Rc comes out, so from the data in Table 2, looking at a slight margin as the average value of Rc, Rc = 1.5Ω, Rr = 3Ω Recalculating with the above formula,
tan ⁻¹ (80/3) = 87.85 degrees Vcp × (1/2) cos 87.85 = Vcp · 0.0187
Vp = (Vp− (Vn)) × 1.0187 (V)
Vn = (Vp− (Vn)) × 0.9812 (V)
The ratio of Vp and Vn, Vp / Vn is
Vp / Vn = 1.0187 / 0.9812 = 1.0381.

  In this way, in consideration of the effective resistance of the capacitor and taking into account significant figures and measurement errors in Table 2 described above, 1 <Vp / Vn <1.04. However, a capacitor that satisfies this condition is also C1e. , C1a and C1b only. That is, it cannot be considered that this asymmetry with respect to the zero potential is caused only by the effective series resistance Rc of the capacitor. It must be considered that the waveform asymmetry with respect to zero potential is caused by factors other than Rc. For example, Lc shown in FIG. 118 can be considered. Therefore, in FIG. 71, a capacitor that satisfies the power transmission performance requirement that P2> 3.25 W and η> 80% from the condition that the shift ratio S is S <1.06 is selected. I do not get.

  When the integration circuit Gi is connected to both ends of the capacitor Cx shown in FIG. 71, the output voltage Vg of the integration circuit Gi is Vg = Vp−Vn and is proportional to the shift ratio S. In order to measure the shift ratio S with the oscilloscope 80, there is a limit to the measurement accuracy. However, the amplitude values Vp (V) and Vn (V) of the voltage waveform of the capacitor can be measured almost accurately with an oscilloscope. If the shift ratio S is S = Vg / Vn + 1 = (Vp−Vn + Vn) / Vn = Vp / Vn, a more accurate shift ratio S is obtained. Alternatively, positive and negative peak hold circuits may be provided at both ends of the capacitor to obtain Vp and Vn.

  The oscilloscope 80 shown in FIG. 71 constitutes means for measuring the positive peak value Vp and the negative peak value Vn. That is, each measuring means described above describes the evaluation apparatus for a capacitor for a high-frequency power circuit according to the present invention.

  It goes without saying that the higher the secondary power, the better the transmission efficiency, regardless of the above rules. When the polypropylene capacitor C1c is used, the primary power in FIG. 72 is 4.5 W, and the secondary AC power is 4.1 W. The sum of the output impedance Zs and the effective series resistance Ri of the coil is 1.55Ω, and the primary side voltage is an effective value of 1V. Therefore, the Joule loss of P = 1V2 / 1.55Ω = 0.65W is generated by the output impedance Zs and the effective series resistance Ri of the coil. That is, the calculated transmission efficiency is 100% or more. The surplus of 0.25 W appears because the waveform on the secondary side is not a perfect sine wave, and the pp voltage value divided by 2√2 does not give an accurate effective value. Is done. That is, when the polypropylene capacitor C1c is used, the transmission efficiency of the power transmission system (between the power transmitting and receiving coil 2) is extremely 100% excluding the power loss due to the output impedance Zs of the AC power supply 30b and the effective series resistance Rc of the capacitor. It seems close.

  As is apparent from FIG. 81, as the capacitor having asymmetrical positive and negative values, that is, Vp / Vn, as shown in FIG. On the other hand, a capacitor with Vp / Vn close to 1 has good power transmission performance.

  In FIG. 81, it can be seen that as Vp / Vn becomes larger than 1, the curve of the power transmission characteristic monotonously decreases. Referring to FIG. 81, when the circuit of FIG. 71 is driven by a square wave of zero potential and positive potential, the sine wave voltage waveform of the capacitor has a shift ratio S of 1.06 or less from the zero point. It will be necessary. However, the specified value, 1.06, was measured at 200 kHz using a 0.01 μF capacitor as described above. Under this condition, the value of Vp / Vn when using the capacitor C1c and the reference coil is 1.02. However, when the same capacitance as the capacitor C1c having a capacitance of 0.47 μF and another reference coil were used and measured at 50 kHz, Vp = 22.8V, Vn = 20.7V, Vp / Vn = 1. 1

  Although it depends on the reference coil and frequency, the inventor of the present application conducted various experiments. Assuming that the capacitance is C (μF) and the coefficient γ is γ = log (C / 0.001), It was found that the specified value can be approximated as 1 + 0.06 × γ. As an example, at 0.47 μF, γ = log (0.47 / 0.001) = log (470) = 2.67. The specified value is 1 + 0.06 × γ = 1 + 0.162 = 1.162.

  When the same polyethylene terephthalate capacitor as the 0.1 μF capacitor C1i is used at 100 kHz, Vp = 16.5V, Vn = 15.1V, and Vp / Vn = 1.09. γ becomes γ = log (0.1 / 0.001) = log (100) = 2. Therefore, the specified value is 1 + 0.06 × α = 1 + 0.12 = 1.12. When the capacitance is 0.01 μF, γ = log (0.01 / 0.001) = log (10) = 1. Therefore, the specified value is 1 + 0.06 × γ = 1 + 0.06 = 1.06. When the capacitance is 0.01 μF or less, the specified value is 1.06.

  Therefore, when the predetermined coefficient is B and the capacitance of the capacitor is C (μF), when C (μF) <0.01, B = 1.06. When C (μF)> 0.01, the predetermined coefficient B is B = 1 + 0.06 × log (C / 0.001).

  The value of Vp / Vn described above also varies depending on the reference coil and the frequency of power transmission. Therefore, the predetermined coefficient B is obtained from the capacitance of the capacitor by the above-described calculation formula. Then, using the power transmission coil 1 that is actually used, Vp and Vn of the voltage across the capacitor are measured at the frequency used for power transmission. Under this condition, when the predetermined count is B, the capacitor only needs to satisfy the condition of Vp / Vn ≦ B.

On the other hand, in FIG. 82, as described above, as the effective series resistance Rc (Ω) of the capacitor increases, the power transmission characteristic curve decreases, and then the effective series resistance Rc increases and decreases around 1.4Ω. Such monotonic singularities are seen. Also in FIG. 83, as the dielectric loss tangent tan δ of the capacitor increases, the curve of the power transmission characteristic does not monotonously decrease and a singular point is seen.
However, in both FIG. 82 and FIG. 83, it is possible to define the characteristics from the correlation between the characteristics as a capacitor and the power transmission performance. In FIG. 82, when the effective series resistance Rc of the capacitor is 1.55Ω or less, the above-described power transmission performance standard is satisfied. The above 1.55Ω is the sum of the effective series resistance Ri = 1.35Ω of the power transmission coil 1 alone and the output impedance Zs = 0.2Ω of the AC power supply 30b.

  However, FIG. 82 shows data at a capacitance of 0.01 μF and a frequency of 200 kHz. Therefore, it is preferable that the reactance considering the frequency and the capacitance is defined by the dielectric loss tangent tan δ in the equation. As long as FIG. 83 is referred to, the relationship between the power transmission performance and the dielectric loss tangent is not monotonously decreased as the X value increases as shown in the graph of FIG. At a predetermined minimum frequency, for example, 100 kHz, 200 kHz, etc., the dielectric loss tangent tan δ of the capacitor is 2% or less, which is a minimum satisfactory condition. However, there is a singular point that does not satisfy the above-described reference values, such as a secondary power of 3.25 W or more and a transmission efficiency of 80% or more in a region where the dielectric loss tangent tan δ of the capacitor is 2% or less. Therefore, the dielectric loss tangent tan δ of the capacitor is more preferably 1% or 0.5% or less.

  FIG. 84 is a graph showing the frequency characteristics of the effective series resistance Rc of a capacitor of the same type as the ceramic capacitor C1r having poor power transmission performance but having a different capacitance.

  FIG. 85 is a graph showing the frequency characteristic of the dielectric loss tangent tan δ of a capacitor of the same type as the ceramic capacitor C1r having poor power transmission performance but having a different capacitance.

  FIG. 86 is a graph showing the frequency characteristics of the effective series resistance Rc of a capacitor of the same type as the polypropylene capacitor C1c having good power transmission performance but having a different capacitance.

  FIG. 87 is a graph showing the frequency characteristic of the dielectric loss tangent tan δ of a capacitor of the same type as the polypropylene capacitor C1c having good power transmission performance and having a different capacitance.

  As long as FIG. 85 and FIG. 87 are referred to, it can be seen that the frequency characteristics of the dielectric loss tangent tan δ differ depending on the capacitor dielectric and capacitance. As shown in FIG. 85, the ceramic capacitor C1r using a dielectric having a high relative dielectric constant has a small dielectric loss tangent tan δ of a capacitor having a large capacitance in the low frequency region. However, in the high frequency region, the dielectric loss tangent tan δ of the capacitor having a large capacitance is large. That is, in a capacitor using a high dielectric, the increase rate of the dielectric loss tangent tan δ accompanying the increase in frequency is smaller as the capacitance is smaller. On the other hand, referring to FIG. 87, the polypropylene capacitor C1c using a dielectric having a low relative dielectric constant has a large dielectric loss tangent tan δ of a capacitor having a large capacitance in almost all frequency regions. 85 and 87 agree with the above-described actual measurement results. That is, it is necessary to select the type of capacitor according to the capacitance and the operating frequency of the power transmission device.

  Referring to FIG. 85, in the ceramic capacitor C1r made of a dielectric material having a high relative dielectric constant, only a capacitor having a capacitance of 0.47 μF at 200 kHz has a dielectric loss tangent tan δ of 1% or less. On the other hand, referring to FIG. 87, the polypropylene capacitor C1c composed of a dielectric material having a low relative dielectric constant has a capacitance of 0.01 μF to 0.47 μF and a dielectric loss tangent tan δ of 1% or less at 200 kHz. It has become.

As mentioned at the outset, to improve the power factor is to cancel the positive reactance. The LC series circuit has a function of returning a non-sinusoidal waveform such as a square wave, a triangular wave, and a sawtooth wave to a sine wave. For that purpose, the Q of the LC circuit must be high, and as a rule, the capacitor needs a Q of at least 10 or more. As shown in Table 2, it is necessary to select a capacitor satisfying the characteristics as described above for a capacitor having a Q of several tens, regardless of a capacitor having a Q exceeding 1000 or 10,000 at 100 kHz.

  In FIG. 82 in which the effective series resistance Rc of the capacitor is the X axis and the power transmission performance is the Y axis, and in FIG. 83 where the dielectric loss tangent tan δ of the capacitor is the X axis and the power transmission performance is the Y axis, the capacitor C1h Inside. On the other hand, in the graph of FIG. 81, all of the capacitors C1h are outside the specified region. Thus, in the present invention, it is possible to exclude capacitors having poor power transmission performance that cannot be defined only by the effective series resistance Rc of the capacitor and the dielectric loss tangent tan δ of the capacitor. According to such a rule, a capacitor having good power transmission performance is selected to configure the power transmission device. As a result, there is an extremely excellent effect that a power transmission device with good power transmission performance can be realized.

  From FIG. 86, when the frequency is in a low frequency region of about 20 kHz, the effective series resistance Rc of the capacitor is several Ω to several tens Ω, and the power loss due to the effective series resistance Rc increases. Therefore, it is preferable that the lower limit of the frequency at which a capacitor having a capacitance of 0.01 μF can be used for power transmission is about 20 kHz. However, when the capacitor has a capacitance of 0.1 μF or more, the effective series resistance Rc is reduced to a value that causes almost no power loss even at 10 kHz. However, the presence of the effective series resistance Rc (Ω) deteriorates the power transmission performance. Therefore, if the effective series resistance of the power transmission coil 1 is Ri (Ω), power transmission is performed at a minimum frequency that satisfies at least Ri (Ω)> Rc (Ω) as the effective series resistance Rc (Ω) of the capacitor. Is preferred. The effective series resistance Rc (Ω) of the capacitor described above relates to the definition of the lowest frequency at which the capacitor can be used. The maximum frequency at which the capacitor can be used will be described later by showing the frequency characteristics of the capacitor impedance.

(Explanation of dielectric absorption)
Next, as one of the factors having a correlation with the power transmission performance, the dielectric absorption of the capacitor can be considered. Dielectric absorption is a direct current characteristic of the capacitor. There are various theories about the cause of the dielectric absorption. As one theory, it is considered that the dielectric polarization occurs while the DC voltage Vw (V) is applied to both terminals of the capacitor for a long time. A DC voltage is no longer applied to the capacitor, and both terminals of the capacitor are short-circuited to discharge the charge accumulated in the capacitor. When the capacitor is opened after discharging, an open circuit voltage Vb (V) is generated across the capacitor. The ratio of Vw (V) to the capacitor Vb (V) is the dielectric absorption K, and K = Vb (V) / Vw (V). The smaller the K, the better the capacitor characteristics. The dielectric absorption K is always a positive value smaller than 1 and is a unitless numerical value.

(Explanation when one capacitor is installed at each end of the coil)
In paragraph 0018 of Patent Document 4, it is described that a capacitor having twice the capacitance necessary for the resonance frequency is provided at each end of the coil. As described above, other patent documents have similar descriptions. In the previous invention, it has been described that the performance of a capacitor can be judged by the ratio S at which Vp and Vn shift from the zero point when the voltage Vp and Vn across the capacitor are measured using a reference coil. The inventor of the present application pays attention to this point, and Vp, Vn, S = Vp / Vn, both when the reference coil is equipped with one capacitor and when two capacitors are provided at both ends of the reference coil. And I measured the power transmission performance.

  In FIG. 71, the source (Vd-terminal) of the switching element Q2 is connected to the GND of the oscilloscope 80, and the connection point between the reference coil Ls and the capacitor Cx is connected to the signal input terminal of the oscilloscope 70. As described above, when measuring Vp and Vn, R2 is short-circuited. By connecting in this way, Vp> Vn and S = Vp / Vn> 1.

  FIG. 88 is a connection diagram when the voltage across the capacitor is measured with an oscilloscope when the capacitor is connected to the VOUTH side. FIG. 89 is a connection diagram for measuring the voltage across each capacitor with an oscilloscope when one capacitor is connected to each end of the power transmission coil 1.

  The output of the AC power supply 30b shown in FIGS. 88 and 89 is indicated by terminals VOUTH and VOUTL having low impedance. A series circuit of the capacitor Cx and the power transmission coil 1 is connected to the terminals VOUTH and VOUTL of the AC power supply 30b shown in FIG. The GND of the oscilloscope 80 is connected to VOUTH or VOUTL of the AC power supply 30b. A voltage input IN of the oscilloscope 80 is connected to a connection point between the capacitor Cx and the power transmission coil 1.

  In FIG. 89, a series circuit of a capacitor Cx1, a power transmission coil 1, and a capacitor Cx2 is connected to terminals VOUTH and VOUTL of the AC power supply 30b. When the capacitor voltages of the two capacitors Cx1 and Cx2 are simultaneously measured with the same oscilloscopes 80a and 80b, the outputs OUTH and OUTL of the AC power supply 30b are short-circuited via the GND of the oscilloscopes 80a and 80b. It is necessary to measure Vp and Vn separately.

  In order to measure the voltage across the capacitor Cx, it is necessary to connect the GND of the oscilloscopes 80a and 80b to the reference voltage point. In FIG. 71, the capacitor Cx and the coil Ls are interchanged, and one terminal of the capacitor Cx is connected to the drain of the switching element Q1 (the + terminal of Vd). One terminal of the reference coil Ls is connected to the source of the switching element Q2 (the negative terminal of Vd). An LC series resonance circuit is configured by connecting the other terminal of the capacitor Cx and the other terminal of the coil Ls. The drain of the switching element Q1 (+ terminal of Vd) is connected to the GND of the oscilloscope 80, and the connection point between the reference coil Ls and the capacitor Cx is connected to the signal input terminal of the oscilloscope 80. In this way, when the oscilloscope 80 measures the voltage across Cx of the LC series resonance circuit in which the capacitor Cx and the coil Ls are interchanged in FIG. 71, Vn> VP. Therefore, the shift ratio is S1, and S1 = Vn / Vp> 1.

  In order to measure Vp and Vn when two capacitors Cx1 and Cx2 are provided at both ends of the reference coil Ls, the connection method of the oscilloscope 80 may be changed as described above. When two capacitors Cx1 and Cx2 are provided at both ends of the reference coil Ls and the connection method of the oscilloscope 80 is changed as described above, Vp <Vn, so that Sn = Vn / Vp> 1 and Sn Define When two capacitors are provided at both ends of the reference coil and the connection method of the oscilloscope 80 is the same as in FIG. 71, Sp = Vp / Vn. S = Vp / Vn and S1 = Vn / Vp are the shift ratios when one capacitor is installed in the reference coil.

  The inventor of the present application connected two capacitors shown in Table 2 in parallel to create two composite capacitors having a capacitance of 0.02 μF. Vp, Vn, when using one synthetic capacitor having a capacitance of 0.01 μF with this synthetic capacitor connected in series and when connecting a synthetic capacitor with a capacitance of 0.02 μF to both ends of the reference coil S, S1, Sp, and Sn were measured. The effective series resistance Rc (Ω) of each capacitor described in Table 2 is half of Rc (Ω) in a synthetic capacitor having a capacitance of 0.01 μF. However, Rc (Ω) is equal to Rc (Ω) described in Table 2 in the case of a synthetic capacitor having a capacitance of 0.01 μF in which two synthetic capacitors having a capacitance of 0.02 μF are connected in series. When Vp, Vn, S, Sp, and Sn are measured under such conditions, the conditions are the same as when one single capacitor is used.

As a result of measurement, Sp and Sn were almost equal in any capacitor. Also, S and S1 were almost equal. However, Sp was often smaller than S. As a general experimental result, if the value of (Sp-1) / (S-1) is 0.75 or less, the power transmission performance of the capacitor is one at each end of the coil than when one capacitor is installed in the coil. It was better to equip. For example, if S = 1.08 and Sp = 1.05,
(Sp-1) / (S-1) = 0.05 / 0.08 = 0.625 <0.75, which satisfies the conditions.

If S = 1.1 and Sp = 1.08,
(Sp-1) / (S-1) = 0.08 / 0.1 = 0.8> 0.75, which does not satisfy the condition. If (Sp-1) / (S-1) = Fa,
Fb = (Sp-1) / (S1-1)
Fc = (Sn-1) / (S-1)
Fd = (Sn-1) / (S1-1)
The equivalent result can be obtained. Any of Fa to Fd may be 0.75 or less. This sorting method is a sorting method that does not depend on the reference coil, capacitance, frequency, or the like as compared with the sorting method in which the value of S = Vp / Vn is simply measured and is equal to or less than the predetermined numerical value B.

  It is presumed that the connection of one capacitor to each end of the coil is due to the effect that the value of Vp / Vn is close to 1. As a cause of this, in the equivalent circuit of FIG. 116, there is a possibility that the effective series resistance Re (Ω) varies depending on the value of the alternating current flowing through the capacitor C1. Or as mentioned above, the possibility of the influence of the equivalent series inductance Lc of a capacitor is also considered. However, the correlation that the power transmission performance is better as the shift ratio S from the reference voltage of the voltage across the capacitor is closer to 1 is clear from FIG. By connecting one capacitor to each end of the power transmission coil 1, the shift ratio S is reduced, and from the above experimental results approaching 1, by connecting one capacitor to each end of the power transmission coil 1, It can be said that the power transmission performance can be improved. In the conventional example, these effects are not clarified at all.

  In addition, the case where two capacitors of 0.02 μF are used and the two capacitors are connected to both ends of the coil is compared with the case where one capacitor of 0.01 μF is used. When two capacitors were used, the value of Vp / Vn approached 1 and the power transmission performance was improved. Since both the coil and the capacitor have effective series resistance and there is a correlation between the symmetry of Vp and Vn and the power transmission performance, the connection method as shown in FIG. 89 improves the “symmetry” of Vp and Vn. There seems to be a possibility.

  FIG. 90 is a circuit diagram for measuring dielectric absorption of a capacitor.

  FIG. 90 shows an impedance conversion circuit using an operational amplifier. The inventor of the present application created an impedance conversion circuit having an input impedance of 1010Ω or more as shown in FIG. 90, and made it possible to measure dielectric absorption with a commonly used digital tester. In FIG. 90, an inverting input terminal and an output terminal of an operational amplifier 61 operating as an impedance conversion circuit are connected, and a DC voltmeter 62 is connected between the output terminal of the operational amplifier 61 and GND. The operational amplifier 61 and the DC voltmeter 62 operate as an electronic DC voltmeter having an input impedance of 1012Ω or more. Generally used digital testers and the like have an input impedance of several MΩ and cannot accurately measure dielectric absorption.

The switch 63 is provided to apply a DC voltage Vw = V1 (V) for accumulating electric charge to the measurement capacitor C. The switch 64 is connected to the capacitor C via a resistor R4 having an accuracy of 5Ω ± 10%. The switch 65 is provided for measuring the dielectric absorption K by applying the charge accumulated in the capacitor C to the non-inverting input terminal of the operational amplifier 61. The operational amplifier 61 has a high impedance with a non-inverting input terminal bias current of about 1 pA and an input impedance of about 10 ¹² Ω.

  The procedure for measuring dielectric absorption with the measuring circuit shown in FIG. 90 is defined in JIS, but will be described below. First, the switch 63 is closed and a voltage of ± 10% of the rated voltage of the capacitor C, Vw = V1 (V), is applied to the capacitor C for 1 hour. Thereafter, the switch 63 is opened, the switch 64 is closed, and the resistor R4 is connected to both ends of the capacitor C for 10 seconds to discharge the charge stored in the capacitor C. After discharging for 10 seconds, the switch 64 is opened to open the capacitor C, the switch 65 is closed to apply the voltage Vb (V) across the capacitor C to the operational amplifier 61, and the voltmeter 62 measures it for 15 minutes. The maximum value of the voltage across the capacitor in the measurement time of 15 minutes is defined as Vb (V). The dielectric absorption K is obtained as K = Vb / Vw.

  The inventor of the present application measured a dielectric absorption K by applying a positive voltage to one terminal of a nonpolar capacitor. After that, when a negative voltage was applied to one terminal and the dielectric absorption K was measured, a result different from the case where the dielectric absorption K was measured by applying a positive voltage to one terminal was obtained. As will be described later, the asymmetry of dielectric absorption correlates with the power transmission performance. Hereinafter, dielectric absorption measured by applying a positive voltage to one terminal, for example, the terminal A of the capacitor C shown in FIG. 90, is denoted as Kp. The dielectric absorption measured by applying a negative voltage to one terminal is expressed as Kn. For example, dielectric absorption measured by applying a positive voltage to the B terminal of the capacitor C shown in FIG. 90 is expressed as Kn.

  As shown in Table 3, in JIS, dielectric absorption is specified only for nonpolar DC capacitors, and the specified value is 0.1%. However, in the embodiment of the present invention, an AC voltage is applied to a nonpolar capacitor to pass an AC current. Dielectric absorption is a direct current characteristic, and the impedance of a capacitor is theoretically infinite at direct current. A bias current flows through the input terminal of the operational amplifier 61 shown in FIG. According to JIS regulations, the insulation resistance of the capacitor is 1 GΩ to 30 GΩ. Even if an operational amplifier having a bias current value of about 1 pA is used, the capacitor is charged by the bias current, and the absolute value of the output terminal voltage of the operational amplifier increases with time. Therefore, a high resistance Rh of 200 MΩ is attached between GND at the non-inverting input terminal of the operational amplifier 61 in FIG.

Further, when the rated voltage of the capacitor is about 100V, actual measurement becomes dangerous. Therefore, a voltage of 20 V was applied to the capacitor C regardless of the rated voltage of the capacitor C. When the voltage applied to the capacitor C is high, the electric field strength in the dielectric increases and dielectric polarization is likely to occur. Therefore, the actual measurement value is within 0.25%, but it is necessary to allow a margin of about 5 times. Therefore, as the capacitor C of the power transmission device, the dielectric absorption K must be at least 1% or less. Actually, when the inventor of the present application forms a circuit with aerial wiring, the insulation resistance of the entire circuit is set to 10 ¹² Ω or more, the rated voltage of 100 V is applied to the capacitor C1m, and the dielectric absorption is measured. 93%. This corresponds to five times the actual measurement value shown in FIG.

  As can be seen from FIG. 91, a capacitor having a small dielectric absorption Kp has good power transmission, and a capacitor having a large dielectric absorption Kp has poor power transmission. Furthermore, as described above, the inventor of the present application interchanges the terminal A of the capacitor C to which a positive voltage is applied and the terminal B of the capacitor C to which a negative voltage is applied in the circuit of FIG. saw.

FIG. 92 is a graph in which the dielectric absorption Kn is measured by switching the terminal A and the terminal B of the capacitor in FIG. 90, Kn is the X axis, and the power transmission performance is the Y axis. FIG. 93 is a graph with the ratio Kr of the dielectric absorption Kp and Kn of each capacitor shown in Table 2 as the X axis and the power transmission performance as the Y axis. The ratio Kr is defined such that when Kp> Kn, Kr = Kp / Kn, and when Kn> Kp, Kr = Kn / Kp, and Kr> 1.

  According to FIG. 93, the asymmetry of dielectric absorption is clearly seen. A capacitor with good power transmission performance has low dielectric absorption asymmetry. In other words, the value of the ratio Kr is close to 1. A capacitor having poor power transmission performance has a large asymmetry of dielectric absorption. In other words, the value of the ratio Kr is greater than 1. In this way, a capacitor with good power transmission performance can be selected.

  That is, when one terminal A and the other terminal B of the capacitor shown in FIG. 90 are interchanged, a result is obtained in which the dielectric absorption Kp and Kn have different values. This is remarkable in the ceramic capacitor C1r having the smallest step-up ratio. As is clear from FIG. 93, it can be seen that a capacitor with a ratio Kr close to 1 has good power transmission, and a capacitor with a ratio Kr larger than 1 has poor power transmission. From the correlation between Kr and power transmission performance in FIG. 93, the ratio of Kp and Kn, Kr, Kr = Kp / Kn, is in the range of 1 <Kp / Kn <1.5. I understand that it is necessary.

(Explanation of the embodiment regarding the temperature rise of the capacitor)
FIG. 94 is a circuit configuration diagram for determining the performance of the capacitor C. The AC power source 72 of FIG. 94 is configured such that the frequency can be varied, the output current Ia can be measured by the AC ammeter 73, and the output voltage Vt (V) can be measured by the AC voltmeter 74. When an AC voltage Vt (V) is applied to the capacitor C, a current Ia (A) due to reactive power flows through the capacitor. Ia (A) = Vt (V) / Xc (Ω). FIG. 95 is a circuit diagram for causing a current to flow through the capacitor Cx by configuring the LCR series resonant circuit shown in FIG. 71 described above.

  FIG. 96 is a graph in which, in the circuits of FIGS. 94 and 95, when an alternating current having a frequency of 200 kHz and an effective value of 0.5 A is passed through the capacitor, the rising temperature of the capacitor is taken as the X axis and the power transmission performance is taken as the Y axis. It is. In order to measure an accurate temperature, the temperature of the capacitor is measured with an infrared non-contact type thermometer.

  As is apparent from FIG. 96, when the temperature of the capacitor C increases, the power transmission performance deteriorates. It can be seen that a predetermined power transmission performance can be obtained if the temperature rise of the capacitor is 10 ° C. or less. It is more preferable if the temperature rise of the capacitor is 5 ° C. or less.

(Description of performance evaluation method and performance evaluation device)
97A is a block diagram showing an example of measuring the performance of a capacitor, and FIG. 97B is an integration included in the double integration type A / D converter 75 shown in FIG. 97A. FIG.

  The A / D converter 75 that operates as the pulse number measuring means and the AD converting means includes an integrating circuit shown in FIG. An integrating capacitor Cx is connected between the inverting input terminal b and the output terminal a of the operational amplifier 78. The integrating capacitor Cx is connected to one end of the resistor Rt and the inverting input terminal of the operational amplifier 78, and the other end of the resistor Rt is input. It is connected to the end VIN. The non-inverting input terminal of the operational amplifier 78 is connected to GND.

The reference voltage Vref is connected to the reference voltage inputs REF_HI and REF_LO of the A / D converter 75. A display 76 is connected to the A / D converter 75. A switch 77 is connected to the measurement voltage input terminals IN_HI and IN_LO. The switch 77 switches whether the reference voltage Vref and the inverted reference voltage −Vref are applied to the measurement voltage input terminals IN_HI and IN_LO, or the short circuit state. When switched to the short circuit state, the switch 77 operates as an initialization means for setting the voltage across the integrating capacitor to zero.

  In this embodiment, the A / D converter 75 causes the constant current Ip in the positive direction to flow through the integration capacitor Cx for a predetermined time T after the integration capacitor Cx is initialized, and integrates after the predetermined time T has elapsed. When a predetermined constant current In in the negative direction is passed through the capacitor Cx and the time until the voltage across the integrating capacitor Cx becomes zero is Tn, the number of pulses at the predetermined time T when Ip = In is counted. When the pulse count for the predetermined time T is N and the pulse count for the time Tn until the voltage across the integration capacitor Cx becomes zero is Nn, the absolute value of the difference between Nn and N is 0. Select a capacitor of .004 × N count or less.

  A double integration A / D converter is used as the A / D converter 75, and the reference voltage and the input voltage of the A / D converter 75 are the same. When the output, for example, the display of the A / D converter 75 shows a deviation from the theoretical value 1, the performance of the power factor capacitor used for the power transmission measure can be determined. The A / D converter 75 has a resolution of 1000 counts or more. This resolution 1000 is the resolution for the theoretical value 1. That is, if the count number is 999, the deviation from the theoretical value 1 is 0.001. Therefore, if the count number is N and the deviation (digit) is 0.004 × N count or less, the power factor improvement performance of the capacitor is good. If the count number N is, for example, 10,000, a more accurate determination can be made. The theoretical value 1 may be 2 n bits, for example, 2047 if it is 11 bits.

  Further, a constant current In in the negative direction is passed through the capacitor for a predetermined time T, and after a predetermined time T has elapsed, a predetermined constant current Ip in the positive direction is passed through the capacitor until the voltage across the capacitor becomes zero. If the time is Tp, and | Ip | = | In |, where the pulse count number of T is N and the pulse count number of Tp is Np, the absolute value of the difference between Np and N is Select a capacitor of 0.004 x N counts or less.

  In this example, when the reference voltage of the A / D converter is the same as the input voltage, the polarity of the input voltage is inverted. As described above in terms of dielectric absorption, even a nonpolar capacitor has a different display when the same positive voltage and the same negative voltage are applied to the A / D converter. By the above method, the power factor improvement performance of the capacitor can be accurately determined.

  Furthermore, if the absolute value of the difference between Np and Nn is 0.003 × N counts or less, the power factor improvement performance of the capacitor is good.

  In this example, the power factor improvement performance of the capacitor can be determined more accurately by comparing the absolute values of the differences between Np and Nn.

  FIG. 98 is a waveform showing the integration circuit output Vo of the A / D converter 75.

In the A / D converter 75 shown in FIG. 97 (B), when the voltage between the measurement voltage input terminals IN_HI and IN_LO, which are differential inputs, is Vm, the integrated current Ii (A) is
Ii (A) = Vm (V) / Rt (Ω). The output of the A / D converter 75 is displayed on the display 76. The input signal integration time is a fixed value of 1000 counts. Therefore, the peak voltage Vpeak (V) shown in FIG. 98 increases in proportion to Vm (V). The relationship between the integration capacitor Cx and Vpeak is Vpeak (V) = Ii (A) / Cx (F). This is because the input signal integration time is constant. After 1000 counts of the input signal integration, the A / D converter 75 is switched to inverse integration. The reverse integration is performed from Vpeak with a constant negative slope by the currents Iref (A) and Iref (A) = Vref (V) / Rt (Ω) generated by the reference voltage Vref operating as a current source. The counter integration time is counted, and the count value when the output of the integration circuit becomes zero is displayed as a 4-digit digital value.

  Assuming that an ideal capacitor Cx shown in FIGS. 97A and 97B is used, and assuming that the voltage input to the measurement voltage input terminals IN_NI and IN_LO of the A / D converter 75 is Vin, A / The display D displayed on the display 76 of the D converter 75 output is D = 1000 × (Vin / Vref). In this case, since Vin = Vref, the display D is always 1000. That is, in an ideal capacitor, the input voltage integrated waveform shown in FIG. 98 and the reference voltage inverse integrated waveform should be symmetric with respect to the peak voltage Vpeak. However, in a capacitor with poor power transmission performance, the input voltage integration waveform and the reference voltage inverse integration waveform are not symmetrical with respect to the peak voltage Vpeak, and a deviation occurs in the count number of integration time. If it is determined whether or not the deviation is within a predetermined value, it is possible to distinguish whether the power transmission performance of the capacitor is good or bad. This will be specifically described below.

  As shown in FIG. 97A, when the switch 77 is switched to the A side, the reference voltage Vref is input to the input Vin. At this time, the display Dp of the display 77 is 1000. When the switch 77 is switched to the B side, the input of the A / D converter 75 is short-circuited. At this time, the display Dz of the display 76 becomes ± 0. When the switch 77 is switched to the C side, the reference voltage −Vref is input to the input Vin. At this time, the display Dn of the display 76 is −1000. An example of the double integral A / D converter 75 having such a function is ICL7106 manufactured by Intersil Corporation. In this embodiment, ICL7136, which is an improved version of ICL7106, is used. In order to perform more precise measurement, there are ICL7135 having a resolution of ± 19999, ICL7109 having a resolution of ± 12 bits in order to take data into a computer instead of a display and measure it. Both are double integration type A / D converters, and A / D converters other than the double integration type cannot be used in the present invention. It should be noted that, other than the double integration type A / D converter described above, those having the same operation principle can be used in the present invention.

  By configuring the circuit as shown in FIG. 97A, the above-described theoretical value is always displayed even if the reference power supply Vref varies. Therefore, the performance of the capacitor can be determined by seeing the deviation from this theoretical value. Although not shown in FIGS. 97A and 97B, the double integration type A / D converter 75 is provided with a circuit (auto zero circuit) for canceling the offset voltage of the operational amplifier 78 constituting the integration circuit. ing. This auto-zero circuit usually automatically adjusts the offset voltage of the integrating circuit to 50 μV or less. Therefore, if Vref is 100 mV or more, the display is always zero when the switch 77 is at the point B. In addition, Vref was set to 200 mV and measured. However, double-integration A / D converters are easily affected by noise, so the display will not be stable and accurate measurement will be difficult unless you pay sufficient attention to mounting, such as shielding by wiring and metal cases. So be careful.

  In FIG. 99, each capacitor shown in Table 2 is connected to the A / D converter 75, and when the switch 77 is point B, the absolute value (unit digit) of deviation from the zero point of each capacitor is taken as the X axis. It is the graph which made electric power transmission performance Y-axis. In FIG. 99, when 0 digit and 1 digit display are repeated, 0 digit is set, and when 1 digit is displayed, 1 digit is set. When measuring Dp and Dn, which will be described later, the deviation from the zero point is corrected. When comparing the power transmission performance of each capacitor shown in Table 2 plotted in FIG. 99 and the digit value, there is no particular relationship between the displayed value and the power transmission performance when the input is shorted due to the difference in the capacitor. It is.

  First, it is assumed that the switch 77 shown in FIG. 97A is switched to the B side, and the input voltage of the A / D converter 75 is set to zero. The display at this time is Dz as a correction value. When 0 display and 1 display are repeated equally, Dz = + 0.5. When −0 display and −1 display are repeated equally, Dz = −0.5. This Dz is corrected by subtracting when measuring Dp and Dn. In the measurement of Dp and Dn, 999.5 is used as the measurement value when 0999 display and 1000 display are repeated equally.

  In FIG. 100, it is assumed that the switch 77 shown in FIG. 97 has been switched to the A side, and the first deviation (digit) of the theoretical display 1000 and the actual display Dp when the input voltage = the reference voltage Vref is set. ) On the X axis, and the power transmission performance on the Y axis. FIG. 101 shows the absolute value of the second deviation (digit) of the theoretical display −1000 and the actual display Dn when the switch 77 is set to the position C in FIG. 97 and the input voltage = −reference voltage. Is a graph with X-axis and power transmission performance as Y-axis. FIG. 102 is a graph in which the absolute value of the difference (digit) between the absolute values of the actual display Dp and Dn is the X axis, and the power transmission performance is the Y axis.

  The capacitors shown in Table 2 are connected to the A / D converter 75. If the absolute value of the display of the A / D converter 75 is between 0996 and 1002, both positive and negative, it can be determined that the capacitor has good performance. In FIGS. 99 and 100, the reason why the maximum width of 4 digits is provided is that the A / D converter 75 normally has a basic error of ± 1 digit. It is also a value (digit) that satisfies the requirements of secondary power 3.25 W and transmission efficiency η 80% or more.

If both the actual displays Dp and Dn can be measured, as shown in FIG. 100, if the absolute value of the difference between the actual displays Dp and Dn is within 3 digits in the display of the A / D converter 75, the predetermined power transmission You can see the performance. For example, if the display of Dp is actually 1001, the display value of Dn only needs to satisfy 0998 <Dn <1004. Alternatively, a precision power supply different from the reference power supply Vref may be used, and the input voltage may be set so that the display of the A / D converter 75 becomes 1990. In this case, if the difference between the actual displays Dp and Dn is within 6 digits, a predetermined power transmission performance can be obtained. These 6 digits are
It is specified as 3 × (1990/1000) ≈6 digits.

  The above formula describes the case where an A / D converter having a maximum count value and 19999 counts is used. For example, in the case where an A / D converter having a maximum count value and 19999 counts is used, FIG. In the case of this circuit configuration, 3 × (10000/1000) ≈30 digits is the specified value. This is the same for Dp and Dn described above. When an A / D converter with 19999 counts is used, the specified value is 40 digits or less.

  Compared with the case where dielectric absorption is measured, the performance of the power factor improving capacitor of the power transmission device can be judged in a short time by using a double integration type A / D converter. As described above, the performance of the power factor improving capacitor of the power transmission device can also be determined from the DC characteristics of the capacitor. In addition to the double integration type A / D converter, V / F converter (voltage frequency converter), F / V converter (frequency voltage converter), VCO (voltage controlled oscillator), sample hold amplifier (instantaneous voltage) Value holding circuit), RMS / DC converter (AC RMS-DC converter), analog circuits such as timers, analog ICs, etc., and circuits that use capacitors, judge the performance of power factor improvement capacitors for power transmission devices Is possible.

  In the V / F converter, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input voltage and the output frequency. An example of such an IC is AD654 from Analog Devices.

  In the F / V converter, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input frequency and the output voltage. An example of such an IC is LM2907 from National Semiconductor.

  Even in a VCO, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by looking at the linearity of the input voltage and the output frequency. An example of such an IC is C-MOS 4046 (manufactured by each company).

  In the sample and hold amplifier, for example, the performance of the power factor improving capacitor of the power transmission device can be judged by inputting a constant DC voltage and looking at the difference between the output voltage and the input voltage. An example of such an IC is Analog Devices' AD585.

  In the RMS / DC converter, for example, the AC power is thermally converted, the accurate AC power is measured, and compared with the output of the RMS / DC converter, thereby determining the performance of the power factor improving capacitor of the power transmission device. Can do. An example of such an IC is AD637 from Analog Devices.

  In the timer, for example, it is possible to judge the performance of the power factor improving capacitor of the power transmission device by installing various capacitors and measuring and comparing the output with a frequency counter. An example of such an IC is a timer IC 555 (manufactured by each company).

  In the analog IC as described above, any capacitor affects the accuracy and function of each IC. Therefore, if the accuracy and function of a plurality of capacitors are compared in the IC as described above, the performance of the power factor improving capacitor of the power transmission device can be judged in the same manner as the double integration type A / D converter. The analog IC described above is an example, and there are many other analog ICs that can determine the performance of the capacitor.

  Thus, dielectric loss tangent at a predetermined frequency, symmetry with respect to zero potential, step-up ratio H, stability of capacitance when an alternating current flows through the capacitor, dielectric absorption, symmetry of dielectric absorption, double integral formula A By defining the actual display Dp of D / D converter 75, the deviation from the theoretical value of Dn, and the display difference between Dp and Dn, an optimum capacitor can be selected for improving the power factor of the power transmission device. Become. By using such a capacitor to improve the power factor of the power transmission device, it is possible to realize a power transmission device with good power transmission performance that was difficult to realize with the conventional technology.

  Next, the difference in power transmission performance due to the difference in capacitance will be examined for capacitors having exactly the same configuration using the same dielectric.

  FIG. 103 is a diagram illustrating the frequency characteristics of the absolute value | Z | of the complex impedance of each capacitor in a capacitor having the same configuration and different capacitance from the polypropylene capacitor C1c having the best power transmission characteristics. FIG. 104 is a diagram illustrating the frequency characteristics of the absolute value | Z | of the complex impedance of each capacitor in a capacitor having the same configuration and different capacitance from the ceramic capacitor C1r having the worst power transmission characteristics.

If the complex impedance Z of the capacitor is Z = Rc + jωC = Rc + Xc,
| Z | is represented by | Z | = √ (Rc2 + Xc2) (Ω).

  Referring to FIGS. 103 and 104, it can be seen that | Z | decreases as the frequency increases. In a capacitor having a large capacitance, | Z | is a minimum value Zb (Ω) in the vicinity of 1 MHz to 2 MHz. It can be seen that | Z | increases at higher frequencies. 103 and 104, the phase angle θ of the 0.47 μF capacitor is plotted. At a frequency fb (Hz) or more at which | Z | is the minimum value Zb (Ω), the phase angle θ is shifted by 180 degrees, and it can be seen that the capacitor operates as an inductor. Therefore, the frequency fb (Hz) at which | Z | becomes the minimum value is the highest frequency at which the capacitor can be used. The frequency fb is a series resonance point determined by the capacitance C of the capacitor and the equivalent series inductance Le in the equivalent circuit of FIG. As will be described later, when the capacitors are connected in parallel, both the effective series resistance Rc and the equivalent series inductance Le decrease. Therefore, a large current can flow through the capacitor and heat generation can be suppressed.

  103 and 104, the polypropylene capacitor C1c and the ceramic capacitor C1r having different dielectrics, structures, and characteristics have substantially the same frequency at which | Z | (Ω) becomes the minimum value if they have the same capacitance. You can see that Thus, by looking at the frequency characteristic of the impedance | Z | (Ω) of the capacitor, the maximum frequency at which the capacitor can be used is known. Alternatively, by looking at the frequency characteristic of the phase difference θ between the current flowing through the capacitor and the voltage across the capacitor, the maximum frequency fb (Hz) at which the capacitor can be used can be found.

  The frequency used for power transmission may be any frequency as long as it is equal to or lower than the maximum frequency fb (Hz). However, capacitor characteristics cannot be defined at any frequency. Therefore, the maximum frequency fb (Hz) is obtained with a capacitor having a predetermined capacitance Ca (F). The characteristic of the capacitor described above may be measured at a frequency equal to or lower than fb / 2 (Hz) with the lowest value of the highest frequency fb (Hz) as a reference, and it may be confirmed whether or not the above-mentioned regulation is satisfied. Alternatively, the same frequency, such as 200 kHz or 500 kHz, that is equal to or lower than the maximum frequency fb may be used as a measurement frequency, and performance comparison of different capacitors may be performed. In this way, even when the components other than the capacitor and the power transmission frequency are changed, the power transmission performance can be ensured. From the above measurement results, the upper limit is about 0.47 μF that can be used as a single capacitor. As will be described later, when a capacitance of 0.47 μF or more is necessary, it is preferable to connect capacitors having a capacitance of 0.47 μF or less in parallel.

  In general, the minimum value of the effective series resistance Rc is regarded as the minimum value of impedance in FIGS. 103 and 104. However, as long as it is compared with FIGS. 84 and 86, the coincidence with FIGS. 103 and 104 is not seen. However, these are only differences between the definition and the measurement method. The present invention realizes a power transmission device with good power transmission performance by selecting a capacitor suitable for power factor improvement of a power transmission device that cannot be defined only by the effective series resistance of the capacitor or the dielectric loss tangent tan δ. As described above, the measurement of the capacitor characteristics using the shift ratio S from the zero point of the sine wave voltage across the capacitor, the specific value of the dielectric tangent, the dielectric absorption, the double integral type A / D converter, etc. Capacitors with good performance can be selected regardless of After that, it is preferable to measure the frequency characteristics of FIGS. 84, 85, 86, 87, 103, and 104 and use the capacitor as a guideline of the usable frequency.

(Description of capacitor configuration)
105 and 106 are diagrams showing specific capacitor configurations according to embodiments of the present invention. As shown in FIGS. 105A, 105B, 106A, and 106B, the capacitor C1 in FIG. 1, which is a circuit configuration diagram of the power transmission device, includes a dielectric material 86 and a metal foil 85. Constructed by being wound, or by alternately laminating dielectric material 86 and metal foil 85, and having a capacitance of 1000 pF or less per 1 cm ^{2 of} single layer made of dielectric material and metal foil Is preferably used. In FIG. 105, instead of the metal foil, the electrode may be formed by depositing metal on a dielectric film. A capacitor having such a configuration is called a metallized capacitor, and even if a pinhole is generated in the dielectric film, the metal vapor deposition layer around the pinhole is evaporated to return to a normal characteristic. This function is called self-healing action or self-healing function.

In the embodiment of the present invention, a capacitor in which a dielectric having a high insulation resistance and a high withstand voltage is installed between conductive foils is used, and a capacitance per 1 cm ² constituted by ^two conductive foils and a dielectric material is used. , 1000 pF or less. A capacitor using a dielectric has different characteristics depending on the structure of the capacitor depending on the capacitance value. The voltage at which dielectric breakdown occurs is as low as about 1000 V / mm. Therefore, a capacitor using a dielectric having a high voltage that causes dielectric breakdown must be used. For example, the dielectric breakdown voltage of a plastic film used for a film capacitor is about 1000 V / μm, which is about 1000 times that of air.

In addition, when the inventors of the present invention conducted experiments using the AC power supply and coil of the embodiment of the present invention and various capacitors, the dielectric was a plastic film, and the structure shown in FIGS. 105 and 106 was used. The capacitor had good power transmission performance. However, a capacitor C1z having an operable voltage of several hundred to 1,000 V or more, for example, using barium titanate as a dielectric, may have better power transmission performance than a capacitor using a plastic film as a dielectric. The C1z is a disk having a diameter of several centimeters or more, and the dielectric material is thick and the shape is large. However, when calculated backward from the capacitance value and the disk area, the capacitance per 1 cm ^{2 of} the single layer is about 200 pF or less. Thus, the above-mentioned condition of the electrostatic capacity of 1000 pF or less per 1 cm ^{2 of} the single layer was satisfied. Therefore, it is important to define the capacitance per unit area as a capacitor used in the power transmission device. As will be described later, the electrostatic capacity per unit area is defined as a condition for determining the other when one of the electrode plate interval and the relative dielectric constant of the dielectric is determined.

The inventor of the present application actually wound the foil-like conductor and film of FIG. 105 (B), and has the same dielectric capacity as that of the polypropylene capacitor C1c having the best power transmission performance in Table 1. When a 0.47 μF capacitor was disassembled and back-calculated from the surface area and nominal capacity, the capacitance per 1 cm ^{2 of} single layer area was about 1000 pF or less. The actual measurement results are shown below.

  When a polypropylene capacitor having the same configuration as the capacitor C1c of 0.47 μF was disassembled and the electrode plate dimensions were measured, it was about 2 cm × 250 cm.

The electrode plate area Sc is Sc = 2 cm × 250 cm = 500 cm ² .

Since the capacitance is 0.47 μF = 470000 pF, the capacitance per 1 cm ² is
470000 pF / 500 cm ² = 940 pF / cm ² .

The capacitance C is expressed as C = εo · εs · S / d. Here, εo is a dielectric constant in vacuum, and is a physical constant of εo = 8.85 × 10 ⁻¹² (F / m). εs is a relative dielectric constant (no unit), S is an area (m2) of the electrode plates, and d is a distance (m) between the electrode plates. Transform the above formula,
Substituting the above constants and measured values as d = εo · εs · S / C,
d = 8.85 × 10 ⁻¹² (F / m) · εs · 10 ⁻⁴ m ² / (940 × 10 ⁻¹² F)
d = (8.85 / 940) · εs · 10 ⁻⁴ m = εs · 9.41 × 10 ⁻⁷ m
Considering dimensional measurement error etc., if 9.41 × 10 ⁻⁷ m is 1 × 10 ⁻⁶ m,
According to the data, d = εs · 1 × 10 ⁻⁶ m = εs · μm, the relative dielectric constant εs of the same dielectric film as that of the capacitor C1c is 1.5 to 4, and between the electrode plates The distance d is required to be at least 1.5 μm. As far as the inventors of the present application have experimented, the polypropylene capacitor C1c having the best power transmission performance falls into a category with a large volume for the same capacitance. That is, the power transmission performance of the capacitor tends to improve in proportion to the volume. Therefore, it is preferable that the upper limit of the relative dielectric constant capable of reducing the electrode plate area S is 4.

  The thickness of the dielectric film measured with a micrometer was about 3 μm. Therefore, the dielectric constant εs of the dielectric film is estimated to be about 3 from the above-described induction formula, d = εs · μm, which is consistent with the numerical value of the material.

(Description of capacitor dielectric)
The capacitor dielectric is a polymer of polyimide, polyethylene terephthalate, polycarbonate, polysulfone, polyphenylene sulfide, polyethylene naphthalate, an olefin monomer represented by a general formula CR ₂ CQ ₂ , and CR ₂ CQ ₂ , R and Q are composed of a polymer which is an addition polymer of a monomer having a functional group containing H, or a mixture of at least two of the above-described polymers.

Here, R and Q are functional groups containing H (hydrogen), for example, a single atom such as Cl (chlorine), a functional group such as CH3 (methyl group), or C ₆ H ₅ (phenyl group). , And the like. For example, when all of R ₂ and Q in CR ₂ CQ ₂ are H, the monomer is ethylene and the polymer is polyethylene. When all of R and Q in CR ₂ CQ ₂ are F, the monomer is tetrafluoroethylene, and the polymer is polytetrafluoroethylene (Teflon (registered trademark)). When one of R in CR ₂ CQ ₂ is a phenyl group and the other R and Q are all H, the monomer is styrene and the polymer is polystyrene. When all of R in CR ₂ CQ ₂ are H and all of Q are F, the monomer is vinylidene fluoride and the polymer is polyvinylidene fluoride. In CR ₂ CQ ₂ , when one of R and Q in CH ₂ is CH ₃ and the remaining Q and R are all H, the monomer is propylene and the polymer is polypropylene. By selecting such a dielectric material and appropriately configuring the capacitor, a capacitor having characteristics suitable for improving the power factor of the power transmission device described above can be obtained. For example, a capacitor having good dielectric absorption characteristics and good symmetry of dielectric absorption characteristics can be obtained.

  The above notation is not an official compound nomenclature established by IUPAC (International Union of Pure and Applied Chemistry). The above notation is a commonly used compound name.

  As far as the inventors of the present application have experimented, if the capacitance is 0.01 μF and the transmission power is about 4 W under the above-mentioned power transmission circuit conditions, polypropylene (PP), polyphenylene sulfide (PPS), polystyrene (PS), polycarbonate ( The power transmission performance was good in the order of the capacitor using the PC) film as a dielectric. However, a capacitor having a different configuration using polypropylene (PP) as a dielectric has a lower power transmission performance than a capacitor using a polyethylene naphthalate (PEN) film as a dielectric. Next, a capacitor using polyethylene naphthalate (PEN) or polyethylene (PE) as a dielectric had good power transmission performance. A capacitor using polyethylene terephthalate (PET) as a dielectric is inferior in performance to a capacitor using polyethylene (PE) as a dielectric.

  However, as described above, the polystyrene capacitor may generate heat depending on the configuration and may not be used in the present invention. In such a polystyrene capacitor, the characteristics of the effective series resistance Rc and the dielectric loss tangent tan δ are close to the maximum performance. However, the shift values S and Vp (V) / Vn (V) from the zero point described above were larger than the polyethylene capacitor. Therefore, the power transmission performance by the dielectric of the capacitor is only an experimental result, and it is important to select the capacitor in accordance with the above-mentioned characteristic definition.

  Note that, as described above, a ferroelectric capacitor as a dielectric, for example, a ceramic capacitor, has a dielectric loss tangent, capacitance stability, capacitance temperature characteristics, dielectric absorption, etc. The performance as a capacitor that affects the power transmission performance is poor, and it is not suitable for a power factor improving capacitor of a power transmission device. However, as described above, when the capacitance increases, power transmission performance that is not significantly different from a capacitor using a dielectric having a small relative dielectric constant can be obtained. Even if the capacitors have exactly the same structure using the same dielectric, the power transmission performance differs depending on the capacitance. Further, as described above, there is an upper limit of the usable frequency due to the capacitance of the capacitor. Although it depends on the frequency, as a guideline, when the capacitance is about 0.1 μF or more, it seems that no significant difference is seen in the power transmission performance between the film capacitor and the ceramic capacitor.

  Referring to Table 2, the effective series resistance Rc at 200 kHz is 0.01Ω for the polystyrene capacitor C1a and 0.03Ω for the polypropylene capacitor C1c. That is, as long as the comparison is made with the value of the effective series resistance Rc, the performance of the polystyrene capacitor C1a is better. Similarly, as long as the values of the dielectric loss tangent tan δ, Q are compared, the performance of the polystyrene capacitor C1a is better. However, the value of Vp / Vn, the dielectric absorption K, the dielectric absorption Kp, the symmetry Kr of Kp, the actual power transmission performance, etc. are better for the polypropylene capacitor C1c. Therefore, it is not possible to judge the performance of the power factor improving capacitor of the power transmission device only by the dielectric loss tangent tan δ or Q. It is only a necessary condition that the dielectric loss tangent tan δ of the capacitor is equal to or less than a predetermined value at the predetermined frequency. 81 to 83, 91 to 93, 100 to 103, etc., it is necessary to select a capacitor with a large numerical value of the capacitor characteristic shown on the Y axis (located on the upper side of the graph).

  In the power transmission device having the circuit configuration shown in FIG. 1, even if a capacitor having good power transmission performance such as a polypropylene capacitor C1c is used, the power transmission control circuit 11, the power transmission coil 1, the power reception coil 2, the relative position of both coils, the load Unless components such as a resistance value are appropriately selected, power transmission performance as shown in FIG. 81 or the like cannot be obtained. Actually, in FIG. 81, the components other than the capacitor are exactly the same, but the power transmission performance differs depending on the capacitor. Therefore, in order to maintain the power transmission performance when the components other than the capacitor are changed, it is necessary to equip the power transmission device with the power factor improving capacitor of the present invention. Components other than the capacitor, particularly the coil and the load resistance value on the secondary side, have a great influence on the power transmission performance. However, in the embodiment of the present invention, if the capacitance is determined, a capacitor with good power transmission performance can be uniquely selected based on the above-mentioned rules. In particular, the capacitor selected by the shift value S from the zero point of the sine wave, the dielectric absorption K, and the symmetry Kr of dielectric absorption is always superior to other capacitors even if the components other than the capacitor change. Transmission performance can be maintained. By installing such a capacitor, a power transmission device with good power transmission performance can be realized.

  In addition, the upper limit which can use a capacitor can be prescribed | regulated by the measurement shown to FIG. 102, FIG. 103 mentioned above. Then, capacitors having good characteristics shown on the Y axis (located on the upper side of the graph) such as those shown in FIGS. 81 to 83, 91 to 93, 96, and 99 to 102 are selected. As described above, the capacitors using the same dielectric have different power transmission performance. Naturally, the power transmission performance varies depending on the dielectric and configuration of the capacitor. However, it is not always possible to use the highest performance capacitor for mounting. Surface mount capacitors require heat resistance during mounting. When a dielectric is exposed to a temperature that exceeds the melting point of the dielectric (exactly the glass transition point, sometimes expressed as the softening point, it is expressed as the melting point), deformation occurs and electrostatic Capacity will fluctuate.

  For such applications, a polyphenylene sulfide capacitor having a high melting point is selected although the power transmission performance is slightly inferior. Alternatively, the mounting may be difficult due to the large physical dimensions of the capacitor. In such a case, a polyethylene naphthalate capacitor having a small physical dimension is selected instead of a polystyrene capacitor or a polycarbonate capacitor having a large physical dimension. As will be described later, the capacitor generates heat when an alternating current flows. In such a case, the capacitors are connected in parallel to ensure a current that can be passed through the capacitor. Since the capacitor connected in parallel has better characteristics than the single capacitor, the characteristics of the capacitor described above are measured by a composite capacitor connected in parallel. This is because at least the effective series resistance Rc can be reduced by connecting capacitors in parallel.

  Moreover, the ceramic capacitor has good heat resistance, and has a power transmission performance that is not so different from that of a film capacitor when the capacitance is 0.1 μF or more. Therefore, by connecting ceramic capacitors having an electrostatic capacity of 0.1 μF or more in series, parallel, and series-parallel, both heat resistance and size reduction can be satisfied. Furthermore, the operable voltage and the passable current can be increased. However, the ceramic capacitor needs to satisfy the above-mentioned regulations. However, the capacitance may be 0.1 μF or less as long as the above-described regulations are satisfied. In general, a chip-shaped (for surface mounting) ceramic capacitor has better characteristics as described above than a lead-shaped ceramic capacitor. The performance of the chip-shaped capacitor is better for the film capacitor.

  By adopting the method of using the capacitor as described above, it is possible to realize a power transmission device with good power transmission performance according to various specifications. A capacitor is selected according to the current flowing through the capacitor. The above-described capacitor having good power transmission performance is generally expensive. By selecting a capacitor by the above-described method, an inexpensive capacitor can be used according to the performance required by the power transmission measure. The embodiment of the present invention described above cannot be defined even in Patent Document 4 which simply describes a dielectric loss tangent or a dielectric.

(Example of circuit configuration)
FIG. 107 is a diagram showing another embodiment of the power transmission device of the present invention. A DC power source 13 and a switching element Q3 are connected in series to the power transmission coil 1, and the switching element Q3 is connected to the control circuit 201. When a unidirectional pulse current is passed through the power transmission coil 1 with the drive waveform VG as shown in FIG. 108 (A) under control, the voltage waveform VL (V) shown in FIG. appear.

  In FIG. 107, it is preferable to connect a capacitor C3 in parallel to the power transmission coil 1 to reduce the circuit current when there is no load and to bring the voltage and current waveform of the power transmission coil 1 closer to a sine wave. The switching element Q3 is driven with the drive waveform VG. The period (frequency) of the drive waveform is constant, and only the time for turning on the switching element Q3 can be changed. By setting it as such a structure, it is prevented that the electric current which flows into the power transmission coil 1 is saturated, and the power loss by the power transmission coil 1 is prevented.

  In addition, by installing the capacitor C3 in parallel with the power transmission coil 1, the voltage waveform and current waveform of the power transmission coil 1 can be brought close to a sine wave, and power loss due to the power transmission coil 1 can be prevented. The capacitor C3 provided in parallel with the power transmission coil 1 is not limited to the capacitor having the characteristics specified in the embodiment as described above, but it is more preferable to use a capacitor having the characteristics as described above. Alternatively, in order to prevent the negative spike voltage due to the counter electromotive force shown in the voltage waveform VL shown in FIG. 108B when the switching element Q3 shown in FIG. It is preferable to connect a capacitor C3 in parallel.

  In the embodiment shown in FIG. 107, the power receiving coil 2 must be equipped with the capacitor C2. However, if the power receiving coil 2 is equipped with the capacitor C2, the power transmission control circuit 11 (drive circuit) of the power transmission coil 1 is not limited to that shown in FIG. Various things such as the embodiment shown in FIG. 70 and a sine wave output can be applied. When the AC power supply 30b of the embodiment shown in FIG. 70 is used, it is preferable to use a power supply whose duty is not fixed at 50% and whose duty is variable. Further, when the power transmission coil 1 is driven by a sine wave, the power transmission coil 1 does not consume any effective power when the power reception coil 2 is not inductively coupled to the power transmission coil 1. This is because all the power supplied to the power transmission coil 1 becomes reactive power. Therefore, there is an advantage that the power transmission coil 1 does not generate any heat even when AC power is supplied to the power transmission coil 1 alone even when power is not transmitted.

  FIG. 109 is a diagram showing another embodiment of the present invention. The circuit configuration is the same as that in FIG. 1, and a capacitor C2 is provided in series between the power receiving coil 2 and the load RL. With the circuit configuration as shown in FIG. 109, the impedance variation on the power transmission side due to the variation in the load resistance value RL can be reduced as compared with the circuit configuration in FIG. The capacitors in the embodiment of the present invention are used for C1 and C2 in FIG. However, C2 in FIG. 109 is mainly installed in a place with a small mounting space such as a power receiving device. In the circuit configuration of FIG. 109, since the frequency range in which predetermined power can be transmitted is wide as described above, the capacitor described in the above embodiment is not necessarily used as the capacitor C2 as long as the passable current is satisfied. You don't have to.

(Explanation of two-terminal equivalent circuit depending on where the capacitor is equipped)
FIG. 110 is an equivalent circuit diagram in the case where the power transmission coil 1 is equipped with a capacitor in series and a simplified two-terminal equivalent circuit diagram.

  FIG. 111 is an equivalent circuit when the receiving coil 2 is equipped with a capacitor in series and a simplified two-terminal equivalent circuit diagram.

  FIG. 112 is an equivalent circuit and a simplified two-terminal equivalent circuit diagram when capacitors are provided in series in both the power transmission coil 1 and the power reception coil 2.

(Explanation when a capacitor is installed on the power transmission side)
An equivalent circuit in which the capacitor C1 is connected in series to the power transmission coil 1 is shown in FIG. 110 (A), and a simplified two-terminal equivalent circuit is shown in FIG. 110 (B), with the capacitor C1, the residual inductance Le, and , A series circuit of resistors Rx. The inductance of the power transmission coil 1 is indicated by L1, and the inductance of the power reception coil 2 is indicated by L2. This is the same as FIG.

  As shown in FIG. 110 (B), I1 has a maximum value at a frequency at which the reactance of the series circuit by the current path I1 becomes zero, and the theoretical power factor is 1. The value of the residual inductance Le is determined by the value of the load resistance RL, the coupling coefficient between both coils, and the coil configuration (air core or core). When the power transmitting coil 1 or the power receiving coil 2 is equipped with a magnetic material, Le> L1 or Le = L1 may be satisfied, which causes a practical problem. That is, the resonance frequency fh (Hz) of the power transmission coil 1 alone becomes equal to the drive frequency fd (Hz) of the power transmission coil 1, and an excessive current flows through the power transmission coil 1 when power transmission is not performed. . This problem and the relationship between the value of the residual inductance Le and the value of L1 will be described later.

(Explanation when a capacitor is installed on the power receiving side)
An equivalent circuit in which the capacitor C2 is connected in series to the power receiving coil 2 is shown in FIG. 111 (A), and a simplified two-terminal equivalent circuit is shown in FIG. 111 (B) as a residual inductance Le, an inductance L2, and A parallel circuit of the capacitor C2 and a series circuit of the resistor Rx are used. In FIG. 111, I1 has a maximum value at a frequency fr1 (Hz) at which the reactance of the series circuit by the current path I1 becomes zero, and the theoretical power factor is 1. At a frequency fr2 (Hz) at which the reactance of the series circuit by the current path I2 becomes zero, a parallel resonance point determined by L2 and C2 is obtained, and I1 becomes zero. If the coupling coefficient between the power transmission coil 1 and the power reception coil 2 is k (k> 0), it is known that Le = L2 (1-k ² ),
fr1 = 1 / √ (L2 (1-k ² ) C2)
fr2 = 1 / √ (L2 · C2)
Therefore, in FIG. 111 (B), fr1> fr2. Therefore, in the circuit configuration in FIG. 111B, there is no problem due to Le> L1, unlike the circuit configuration in FIG. 110B. The above only shows that the problem due to Le> L1 does not occur in FIG. 111B, so that it is omitted to introduce that Le = L2 (1-k ² ). deep. Even if there is a problem, if the power transmission coil 1 is driven by a sine wave, the power transmission coil 1 alone does not consume power as described above.

(Explanation when capacitors are installed on both the power transmission side and the power reception side)
An equivalent circuit in which the capacitor C1 is connected in series to the power transmission coil 1 and the capacitor C1 is connected in series to the power reception coil 2 is shown in FIG. As shown in FIG. 112B, a simplified two-terminal equivalent circuit is a series circuit of a capacitor C1, a residual inductance Le, a parallel circuit of an inductance L2 and a capacitor C2, and a resistor Rx. In the current path I1 on the power receiving side alone, I1 becomes a maximum value at a frequency fr1 (Hz) at which the reactance of the series circuit becomes zero, and the theoretical power factor becomes 1. In the current path I2 of the parallel circuit, at the frequency fr2 (Hz) at which the reactance of the series circuit becomes zero, the parallel resonance point determined by L2 and C2 is obtained, and I1 becomes zero. Further, in the current path I3 of the regular series circuit as viewed from the power transmission side, I3 becomes a maximum value at a frequency fr3 (Hz) at which the reactance of the series circuit becomes zero, and the theoretical power factor becomes 1. When the capacitance of a capacitor in which C1 and C2 are connected in series is C12 (F), in FIG.
fr1 = 1 / √ ((Le + L2) C2)
fr3 = 1 / √ (Le · C12)
Since Le + L2> Le and C2> C12, fr3> fr1. further,
fr2 = 1 / √ (L2 · C2)
It is known that the parallel resonance point fr2 is between the series resonance points fr1 (Hz) and fr3 (Hz). Therefore, the relationship is fr1 <fr2 <fr3.

  Here, in order not to satisfy Le · C12 ≠ L2 · C2, that is, fr2 ≠ fr3, the resonance frequency fh (Hz) of the single power transmission side and the resonance frequency fk (Hz) of the single power reception side are set equal. Just keep it. In this case, fr3> fr2 is always satisfied.

Let's show the above. In the equivalent circuit of FIG. 112 (B), the parallel resonance frequency fr2 (Hz) is
fr2 = 1 / 2π√ (L2 · C2)
The series resonance frequency fr3 (Hz) seen from the power transmission coil 1 side is
fr3 = 1 / 2π√ (Le · C12).

Since the series resonance frequency fh of the single power transmission side is equal to the series resonance frequency fk of the single power reception side,
Ω ² = 1 / (L1 · C1), which is the series resonance condition on the power transmission side,
From the two equations ω ² = 1 / (L2 · C2), which is the parallel resonance condition on the power receiving side,
1 / (L1 · C1) = 1 / (L2 · C2) is established. Transforming the previous equation,
C1 = C2 (L2 / L1).

Substituting C1 = C2 (L2 / L1) into C1 · C2 / (C1 + C2),
C2 ² (L2 / L1) / (C2 (L2 / L1) + C2) = C2 / L2 / (L1 + L2). When Lb is Lb = L1 · L2 / (L1 + L2), the series resonance frequency fr3 (Hz) viewed from the power transmission coil 1 side is
fr3 = 1 / 2π√ (C2 · Lb).

L2−L1 · L2 / (L1 + L2) = L2 ² / (L1 + L2)> 0,
L2> L1 · L2 / (L1 + L2). Therefore, since Lb <L2,
In fr3 = 1 / 2π√ (Lb · C2), Lb · C2 <L2 · C2, and
Since fr2 = 1 / 2π√ (L2 · C2), fr3> fr2. That is, the series resonance frequency fr3 (Hz) viewed from the power transmission coil 1 side is necessarily higher than the parallel resonance frequency fr2 (Hz) viewed from the power transmission coil 1 side. The conventional example does not show the characteristics shown in FIGS. 66 to 68, and simply defines the magnitude relationship between the resonance frequency fh (Hz) of the single power transmission side and the resonance frequency fk (Hz) of the single power reception side. However, as shown in the above calculation formula, fh = fk is set, and the drive frequency fd (Hz) of the power transmission coil may be set as fd> fh.

(Example of connecting multiple capacitors)
As described above, in the series circuit shown in FIG. 116, a boosting effect that a voltage higher than the power supply voltage Vt occurs at both ends of the capacitor occurs. In the circuit configurations of FIGS. 1, 107, and 109, which are the same as the equivalent circuit of FIG. 116, when the operable voltage of the capacitor described in the embodiment of the present invention is low, the same type is used to ensure the operable voltage of the capacitor. Alternatively, a plurality of different types of capacitors may be connected in series. In addition, when the passable current of the capacitor described in the embodiment of the present invention is low, a plurality of same or different types of capacitors may be connected in parallel in order to secure the passable current of the capacitor. Alternatively, in order to secure both the operable voltage and the passable current of the capacitor, a plurality of capacitors of the same type or different types may be connected in series and parallel. As described above, the capacitors are connected in parallel in order to secure a current that can be passed through the capacitor and to suppress the heat generation of the capacitor to 5 ° C. to 10 ° C. or less.

  Preferably, when capacitors are connected in series and parallel, the number of capacitors connected in series is the same as the number of capacitors connected in parallel. FIG. 113 is a connection diagram when capacitors are connected in series and parallel. 113 (A), 113 (B), and 113 (C) are connection examples when the number of capacitors connected in series is three and the number of capacitors connected in parallel is two.

  The reason why the value of the capacitor Cd connected in series and in parallel is within ± 20% is to make the voltage applied to each capacitor Cd and the current flowing through each capacitor Cd as close as possible. There are capacitors with an accuracy of 1% or less, but these are expensive. Since the capacitance described in a general capacitor has a deviation of about 5 to 10%, the nominal value of the capacitor Cd may be selected within about ± 20%. For example, on the basis of 0.01 μF, the upper limit is 0.012 μF, and the lower limit is 0.0082 μF. More preferably, capacitors having the same nominal value are used for all Cd. When connecting capacitors of the same type and nominal value in series and parallel, an excessive voltage is applied to the capacitor Cd1 when the connection is made as shown in FIG. 113D, so that such a connection method is not used. When capacitors are connected in series and parallel, there is provided a rule that the number of capacitors connected in series is the same as the number of capacitors connected in parallel.

  If more precise adjustment is required, the capacitor Cc is used as shown in FIG. In order to finely adjust the capacitance of the capacitor, the single capacitor Cd, the composite capacitor Cp, or the capacitor Cc connected in parallel to the Cd constituting the composite capacitor Cp must always satisfy the characteristic definition of the embodiment of the present invention. No, but preferably satisfied. However, the capacitor Cc connected in parallel is also connected to the capacitor Cc connected in parallel when the output frequency of the AC power supply is set to a point where the reactance is zero determined by Lp and C. Cc has an operable voltage performance higher than the voltage Vc (V), and a capacitor Cc connected in parallel with a passable current performance higher than the AC current Ia (A) flowing in the capacitor Cc connected in parallel. It is a condition that it is doing. Note that the composite capacitor Cp is composed of terminals on both the left and right ends shown in each drawing of FIG.

  At the point where the reactance of the series circuit of the coil and the capacitor becomes zero, the current Ia (A) flows through the capacitor due to the resonance action, and the voltage Vr (V) = Vs (V) × Qr is applied. It is important that the effective series resistance Rc of the capacitor is selected to be sufficiently low, and the frequency satisfying the thermal condition is selected for power transmission.

  The voltage Vc (V) across the capacitor may be measured when the maximum power is transmitted in the circuit of FIG. After the actual measurement, when the voltage Vc (V) at both ends exceeds the operable voltage of the capacitor, the capacitors are connected in series as described above. Similarly, the current flowing in the capacitor is measured when the maximum power is transmitted in the circuit of FIG. The alternating current that can be passed through the capacitor may be defined as a ripple current. However, if there is no regulation, the temperature rise of the capacitor is measured as described above. When the temperature rise exceeds 10 ° C., capacitors are connected in parallel. When both the operable voltage and the passable current of the capacitor are exceeded, the capacitors are connected in series and parallel. When the capacitors are connected in series, parallel, and series-parallel, the value of Cd constituting the composite capacitor is selected so that the capacitance of the composite capacitor becomes a predetermined value. And the above-mentioned characteristic definition is measured by the synthetic capacitor.

  Further, an LCR meter 4275A manufactured by Hewlett-Packard Company was used for measuring the effective resistance and inductance of each capacitor described above. In addition, since measurement can be performed only at each point of 1, 2, 4, and 10, intermediate points are interpolated by a graph. A Kenwood oscilloscope, CS-5370, was used for AC waveform measurement. In the AC wave diagram, the peak value is measured with the cursor and entered as a numerical value on the graph. CS-5370 is also used for measuring power transmission performance. For example, the voltage across the non-inductive load resistor connected to the power receiving coil 2 is measured, and the effective power transmitted to the non-inductive load resistor is obtained.

  As mentioned above, although embodiment of this invention was described with reference to drawings, this invention is not limited to the thing of embodiment shown in figure. Various modifications and variations can be made to the illustrated embodiment within the same range or equivalent range as the present invention.

  The power transmission device, the power transmission device of the power transmission device, and the power reception device of the present invention can be used to transmit power from the power transmission unit to the power reception unit without using electric wires or mechanical contacts.

1 is a block diagram of a power transmission device according to an embodiment of the present invention. It is a figure which shows the coil used as the power transmission coil 1 or the receiving coil 2 of the electric power transmission apparatus shown in FIG. It is a figure which shows the modification of the external shape of the coil shown in FIG. It is an equivalent circuit for obtaining the input impedance of the transformer. It is a figure which shows the equivalent circuit of the coil single-piece | unit in the coil of the electric power transmission apparatus in one Embodiment of this invention. It is a figure showing the equivalent circuit of the transformer part of the electric power transmission apparatus comprised like FIG. 114 demonstrated in the prior art example. It is a figure showing the equivalent circuit of a transformer when a secondary side coil is short-circuited. It is a figure showing the equivalent circuit of a transformer when load resistance RL is connected to the secondary side coil. The figure which shows the relationship between effective electric power transmission efficiency (eta) and frequency when it is set as Rw, Rn, Rs, and load resistance value RL = 10 (ohm) of the coil 1A which wound the winding of 1 turn of single conductor of wire diameter 1mm by 25 turns. It is. It is a figure which shows the relationship between Rw, Rn, Rs, kr, ki, and a frequency of the coil 1B which closely wound 40 turns of the single conductor wire with a wire diameter of 0.6 mm with an outer diameter of 70 mm. It is a figure which shows the relationship between the phase angle of the coil 1C which wound the single conducting wire of wire diameter 0.3mm closely by 70 turns with a diameter of 70 mm, and the phase angle and frequency of the coil 1C single-piece | unit. It is a figure which shows the relationship between Rw, Rn, Rs, and frequency of the coil 1D which wound the single conducting wire of wire diameter 0.3mm closely by 31 turns with a diameter of 30 mm. It is a figure which shows the relationship between Rw, Rn, Rs, kr, and the frequency of the coil 1E which wound the 14-turn coil which provided the space | gap with the outer diameter of 70 mm with the wire diameter of 1 mm. It is a figure which shows the relationship between Rw, Rn, Rs, kr, ki, and the frequency of the coil 1F which wound the Litz wire which bundled 75 formal single conductor wires with a copper wire diameter of 0.05 mm closely with an outer diameter of 70 mm for 30 turns. . It is a figure showing the relationship between Rw, Rn, Rs, kr, ki and the frequency of the coil 1G which closely wound 20 turns of the Litz wire which bundled 75 formal single conductor wires with a copper wire diameter of 0.05 mm with an outer diameter of 50 mm. It is a figure which shows the relationship between the frequency of the coil which wound the formal single conducting wire of 0.2 mm, 0.4 mm, 0.8 mm, and 1 mm 25 times in the shape of a plate, and the effective resistance Rw of each coil. FIG. 10 is a diagram illustrating a relationship between effective power transmission efficiency η and frequency when Rw, Rn, Rs, and load resistance value RL = 10Ω when the coil 1F is opposed to the coil 1A illustrated in FIG. 9. It is an equivalent circuit diagram of a litz wire. FIG. 14 is a diagram showing a comparison of states in which the coil effective series resistance Rw of the closely wound coil 1A shown in FIG. 9 and the loosely wound coil 1E shown in FIG. 13 increases. It is a figure which shows the relationship between Rw and the frequency of each coil which provided the gap | interval width of 0, 0.2 mm, and 0.4 mm, and wound 25 turns for the formal wire of wire diameter 0.4mm. It is an actual measurement figure which shows the frequency characteristic of each resistance value and power factor when coil 1A is used for the power transmission coil 1 and the receiving coil 2, and load resistance value RL is changed. It is an actual measurement figure which shows frequency characteristics of each resistance value and power factor when coil 1A is used for power transmission coil 1, coil 1F is used for power reception coil 2, and load resistance value RL is changed. It is a figure which shows the structure of the coil 1f which prevents radiation required. It is a figure which shows the coil 1g which provided the insulating board between the coil and the magnetic material board. It is a figure which shows the coil 1h which provided the magnetic material board of 2 layers of the magnetic material board and the magnetic material board in the one surface side of the coil. It is a figure which shows the coil 1j which provided the insulating board with thickness I between the two magnetic material boards which comprise the coil 1h shown in FIG. It is a figure which shows the structure of the coil 1k which prevents the characteristic fluctuation | variation of a coil when a metal body adjoins to a coil. It is a figure which shows the structure of the coil 1m which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 1n which prevents both the proximity | contact effect of a metal body and unnecessary radiation. It is a figure which shows the structure of the coil 1p which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 1q which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 1r which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 1s which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a figure which shows the structure of the coil 1t which prevents both the proximity | contact effect of a metal body, and unnecessary radiation. It is a three-dimensional view of the elements that constitute both opposing coils. It is a figure which shows the state which each center corresponds and opposes, when the outer diameters of two coils differ. It is a figure which shows the state which each center has shifted | deviated and opposed when the outer diameters of two coils differ. It is a figure which shows the relative positional relationship of a power transmission coil and a receiving coil in case a power transmission coil and a receiving coil are both elliptical. It is a figure which shows an example of the power transmission coil 1 and the receiving coil 2 which oppose when a coil opposing surface is other than circular. It is a figure which shows the relationship between the effective series resistance Rw of each coil shown in FIGS. 23-26, and a frequency. It is a figure which shows the relationship between Q of each coil shown in FIGS. 23-26, and a frequency. It is a characteristic view which shows the effective series resistance Rw of the coil 1G single-piece | unit, and the inductance Lw of the coil 1G single-piece | unit. This is a characteristic of the coil 1Ga in which various metal plates are opposed to the coil 1G via a 10 mm insulator. FIG. 6 is a characteristic diagram showing an effective series resistance Rw and an inductance Lw when various metal plates are brought close to a coil 1Gb in which one magnetic material plate is provided on the coil 1G. FIG. 6 is a characteristic diagram showing an effective series resistance Rw and an inductance Lw when various metal plates are brought close to a coil 1Gc in which two magnetic material plates are provided on the coil 1G. It is a characteristic view which shows the structure of each coil, and the relationship of Q. FIG. 28 is a characteristic diagram in which the effective series resistance Rw and the inductance Lw are measured with the coil 1D shown in FIG. 12 having the same configuration as that of the coil 1k shown in FIG. FIG. 28 is a characteristic diagram obtained by measuring an effective series resistance Rw and an inductance Lw of the coil 1E shown in FIG. 13 in a configuration equivalent to the coil 1k shown in FIG. It is a figure which shows an example of a structure of the coil which can be inductively coupled between the same two coils equipped with the core. It is a figure which shows the relationship between Rw, Rs, Rn of a coil single-piece | unit of the coil 1H which has the structure of FIG. 49, and a frequency. It is a figure which shows the relationship between Lw, Ls, Ln, and a frequency of the coil 1H and the two coils 1Ha and 1Hb. It is a figure which shows the relationship between Lw, Ls, Ln, the coupling coefficient ki calculated | required approximately from Lw and Ls, and frequency measured when the coil 1G is an air-core state. It is a figure which shows the structure of the inseparable transformer which wound the primary coil and the secondary coil around the toroidal core. In coil 1Gb, it is a figure which shows the relationship between Lw, Ls, Ln, ki, ki2, and a frequency when two coils 1Gb are used. It is a characteristic view which shows the effective series resistance Rw and the inductance Lw in 100 kHz when various metal plates are made to adjoin to the coil 1H. It is a figure which shows the relationship between Rw, Rs, Rn and frequency when two coils 1Gd are used in the coil 1Gd provided with a magnetic material plate in the coil 1G. The effective series resistance Rw (Ω) of a single coil of coil 1H and two coils 1H are inductively coupled, and when the other coil is short-circuited, the effective series resistance Rs (Ω) of one coil is opened and the other coil is opened. It is a figure which shows the relationship between Rw, Rs, Rn and frequency when the effective series resistance of one coil is Rn (Ω). The inductance Lw (μH) of the single coil 1H, the inductance Ls (μH) of one coil when the two coils 1H are inductively coupled and the other coil is short-circuited, and the inductance of one coil when the other coil is opened It is a figure which shows the relationship between Lw, Ls, and Ln and frequency when letting be Ln (μH). It is a characteristic view which shows the effective series resistance Rw and the inductance Lw in 100 kHz when various metal plates are made to adjoin to the coil 1H. It is a figure which shows the relationship between Rw, Rs, Rn, and a frequency when two coils 1Gb are used and the opposing distance Z is 3 mm. It is a figure which shows the relationship between Rw, Rs, Rn and frequency when two coils 1Gd equipped with 0.5 mm-thick aluminum metal plate are used for the coil 1Gb and the opposing distance is zero. It is a figure which shows the relationship between Rw, Rs, Rn and a frequency when two coils 1Gd are used and the facing distance is 3 mm. It is a figure which uses the metal plate with which the coil was equipped as a wire taken out from a coil inner peripheral part in the coil 1t shown in FIG. 34 from the coil 1k shown in FIG. The lead wire used for the coil of each structure shown in FIGS. 23 to 34 is a litz wire, one end of the litz wire is connected to all the strands, and the other end of the litz wire is a strand constituting the litz wire. It is an equivalent circuit diagram when at least one is taken out as a common line. Relationship between load current of secondary winding and voltage at both ends of secondary winding in normal transformer (transformer) configured in tight coupling state, load of receiving coil of power transmission device in embodiment of the present invention It is a characteristic view which shows the relationship between an electric current and the both-ends voltage of a secondary side winding. It is an equivalent circuit diagram in case the power transmission coil 1 and the power receiving coil 2 are inductively coupled, and both coils constitute a transformer. In the equivalent circuit shown in FIG. 66, it is a figure which shows the change of the inductance of the primary side (power transmission side) at the time of using an air core coil for the power transmission coil 1 and the receiving coil 2. FIG. In the equivalent circuit shown in FIG. 66, it is a figure which shows the other example of the change of the inductance of the power transmission coil 1 at the time of using the coil equipped with the magnetic material for at least one of the power transmission coil 1 or the power receiving coil 2. FIG. In the equivalent circuit shown in FIG. 66, it is a figure which shows the other example of the change of the inductance of the power transmission coil 1 at the time of using the coil equipped with the magnetic material for at least one of the power transmission coil 1 or the power receiving coil 2. FIG. FIG. 2 is a circuit diagram of a DC-AC conversion circuit included in the power transmission control circuit of FIG. 1. It is a circuit diagram which comprises the power transmission part of FIG. FIG. 72 is an equivalent circuit diagram including a resistance component existing in the circuit of FIG. 71. It is a circuit diagram which shows the principle which measures an impedance. It is a graph which shows the time change of the electrostatic capacitance of a capacitor. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, and the step-up ratio H. FIG. 72 is a circuit diagram in which a resistor R3 is provided in series with the circuit of FIG. 71. 71 is a voltage waveform of the polypropylene capacitor C1c when the polypropylene capacitor C1c is installed in the circuit of FIG. 71 is a voltage waveform of the ceramic capacitor C1r when the ceramic capacitor C1r is installed in the circuit of FIG. 76 is a voltage waveform across a capacitor in the circuit of FIG. 76. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, and the shift ratio S from the zero point of the sine wave voltage waveform of both ends of a capacitor. It is a graph which shows the relationship between the shift ratio S from the zero point of the sine wave voltage waveform of a capacitor both ends, and electric power transmission performance. It is a graph which shows the relationship between the effective series resistance Rc of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between the dielectric loss tangent tan-delta of a capacitor, and electric power transmission performance. It is a graph which shows the frequency characteristic of the effective series resistance Rc of the ceramic capacitor C1r. It is a graph which shows the frequency characteristic of the effective series resistance Rc of the ceramic capacitor C1r. It is a graph which shows the frequency characteristic of the dielectric loss tangent tan-delta of the polypropylene capacitor C1c. It is a graph which shows the frequency characteristic of the dielectric loss tangent tan-delta of the polypropylene capacitor C1c. It is a connection diagram in the case of measuring the voltage across the capacitor with an oscilloscope when the capacitor is connected to the VOUTH side. It is a connection diagram in the case of measuring the voltage across each capacitor with an oscilloscope when one capacitor is connected to each end of the power transmission coil. It is a circuit diagram which measures the dielectric absorption characteristic of a capacitor. It is a graph which shows the relationship between the dielectric absorption Kp of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between the dielectric absorption Kn of a capacitor, and electric power transmission performance. It is a graph which shows the relationship between Kr which is the ratio of dielectric absorption Kp and Kn of a capacitor, and electric power transmission performance. It is an example of the circuit which measures the heat_generation | fever by the effective series resistance Rc of a capacitor with the electric current which flows into a capacitor. It is another example of the circuit which measures the heat_generation | fever of a capacitor by the effective series resistance Rc of a capacitor with the electric current which flows into a capacitor. It is a graph which shows the temperature rise of a capacitor, and the relationship between electric power transmission performance. It is a block diagram showing other embodiment which measures the performance of a capacitor. FIG. 98 is a diagram illustrating a voltage waveform of an output of the integrating circuit in FIG. 97. It is a graph which shows the relationship between the display Dz of A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the display Dp of an A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the display Dn of an A / D converter, and electric power transmission performance. It is a graph which shows the relationship between the difference Dd of the display Dp and Dn of an A / D converter, and electric power transmission performance. It is a graph which shows the frequency characteristic of the impedance of polypropylene capacitor C1c. It is a graph which shows the frequency characteristic of the impedance of ceramic capacitor C1r. It is an example which shows the structure of the capacitor in embodiment of this invention. It is another example which shows the structure of the capacitor in embodiment of this invention. It is a circuit diagram which shows other embodiment of the power transmission apparatus of this invention. It is a figure which shows the both-ends waveform of the coil in FIG. It is a block diagram which shows other embodiment of the power transmission apparatus of this invention. It is the equivalent circuit at the time of equip | installing the capacitor | condenser with the power transmission coil 1 in series, and the simplified equivalent circuit diagram of 2 terminals. It is the equivalent circuit at the time of equip | installing the capacitor | condenser with the receiving coil 2 in series, and the simplified equivalent circuit diagram of 2 terminals. It is an equivalent circuit when a capacitor is provided in series in both the power transmission coil 1 and the power reception coil 2, and a simplified equivalent circuit diagram of two terminals. It is a circuit diagram which connects a some capacitor. It is a prior art example which shows the structure of an electric power transmission apparatus. It is the top view and sectional drawing of a power transmission coil or a receiving coil. 114 is a simplified equivalent circuit when the power transmission coil 1 is viewed from the power transmission unit. 114 is an equivalent circuit of the power transmission device in which the primary side coil and the secondary side coil shown in FIG. 114 are separable. It is an equivalent circuit diagram of a capacitor.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 Power transmission coil, 1a coil, 2 Power reception coil, 12 DC power supply, 13, 72 AC power supply, 30c, Control circuit, 30 Power transmission part, 30a, 201 Power transmission control circuit, 40a Power reception control circuit, 41 AC constant current source, 42 AC Voltmeter, 61 operational amplifier, 62 DC voltmeter, 63, 64, 65 switch, 70 transformer, 73 toroidal core, 71 primary coil, 72 secondary coil, 75 A / D converter, 76 display, 77 switch, 78 Operational amplifier, 80, 80a, 80b Oscilloscope, 81, 82, 83, 84, Capacitor, 85 Foil-shaped metal, 86 Foil-shaped dielectric material, 100 Power transmission device, Q1, Q2, Q3 Switching element, C, C1-C3, Cc , Cd, Cp capacitor, C1c polypropylene capacitor, C1h, C1r ceramic Yapashita, Le residual inductance, Ls reference inductance, Rc, Ri effective series resistance, R1 to R4, Rh resistance, the internal impedance of Zs AC power source.

Claims (18)

An AC power source which is a power conversion means for converting DC power into AC power, and a power transmission unit including at least a power transmission coil,
The power receiving unit including at least the load and the power receiving coil is configured to be separable,
In the power transmission device that transmits the power from the power transmission unit to the power reception unit by facing the power transmission coil and the power reception coil,
Between the AC power source and the power transmission coil,
Between the load and the power receiving coil,
A series circuit in which at least one nonpolar capacitor is connected in series to cancel the residual reactance component of the power transmission coil and improve the power factor,
Of the opposing coils, the effective series resistance of one of the single coils is Rw (Ω),
The effective series resistance of the one coil when both ends of the other coil facing the one coil are short-circuited is Rs (Ω),
When the maximum frequency satisfying Rs> Rw of the one coil is f1 (Hz),
The one coil and the other coil are selected so that the f1 (Hz) is 100 kHz or more,
The power transmission device, wherein a frequency for driving one of the coils is set to a frequency less than the f1 (Hz).   The power transmission device according to claim 1, wherein a dielectric loss tangent tan δ at a predetermined frequency of the capacitor is 1% or less.   The power transmission device according to claim 2, wherein the dielectric loss tangent tan δ has a value at 200 kHz of 1% or less.   4. The power transmission device according to claim 1, wherein a temperature change of the capacitance of the capacitor is ± 5% or less between 0 ° C. and 80 ° C. 5. When an AC constant current is passed through the capacitor,
The fluctuation of the voltage across the capacitor is
± 0.1% after 5 seconds from the start of current flow,
The fluctuation rate of the capacitance is 1% or less in a measurement time of 5 minutes after starting to flow current.
The capacitance variation rate is 0.2% or less in a measurement time of 1 minute after starting to flow current,
5. The power transmission device according to claim 1, wherein at least one condition is satisfied. The output voltage of the AC power supply is a square wave with a duty of 50%, the level of which changes between a zero potential and a predetermined potential.
If the frequency at which the reactance Xc of the capacitor is equal to the reactance Xi of the power transmission coil is fr,
The output frequency of the AC power supply is set to fr,
At the output frequency fr,
The positive peak value of the voltage across the capacitor with respect to the zero potential is Vp,
The negative peak value of the voltage across the capacitor with respect to the zero potential is Vn,
The ratio of Vp and Vn is Vp / Vn,
When the predetermined coefficient is B,
The power transmission device according to claim 1, wherein the capacitor satisfies at least Vp / Vn ≦ B. In a series resonant circuit in which the capacitor is connected in series to an AC power source and a reference coil,
The output frequency of the AC power supply is set to a resonance frequency determined by the inductance of the reference coil and the capacitance of the capacitor,
Q of the series resonant circuit is Qr,
When the resonance current flows through the series resonance circuit, the output voltage of the AC power supply is Vt,
The voltage across the capacitor is Vc,
If the boost ratio for boosting Vt to Vc is H, and H = Vc / Vt,
The power transmission device according to claim 1, wherein H> 0.9 × Qr is satisfied. Dielectric absorption when a positive voltage is applied to one of the two terminals of the capacitor is represented by Kp,
Dielectric absorption when a negative voltage is applied to the one terminal is Kn,
When Kp> Kn,
The ratio Kr between Kp and Kn is Kr = Kp / Kn,
When Kp <Kn,
The Kr is Kr = Kn / Kp,
The power transmission device according to claim 1, wherein the capacitor satisfies 1 <Kr <1.5. An integration circuit is constituted by the capacitor,
The integration circuit includes:
A current source that allows a constant current of the same value of a predetermined value to flow in both forward and reverse directions;
Initialization means for zeroing the voltage across the capacitor;
Pulse number measuring means for measuring at least 1000 counts of a reference pulse for measuring the integration time,
When the initial voltage of the capacitor is set to zero by the initialization unit, the pulse measurement value of the pulse number measurement unit is initialized to zero,
The constant current Ip in the positive direction is passed through the capacitor for a predetermined time T,
After the elapse of the predetermined time T, a predetermined constant current In in the negative direction is passed through the capacitor,
If the time until the voltage across the capacitor becomes zero is Tn,
| Ip | = | In |
The pulse count number of the predetermined time T is N,
When the pulse count number of the time Tn until the voltage across the capacitor becomes zero is Nn,
The power transmission device according to claim 1, wherein a capacitor having an absolute value of a difference between Nn and N of 0.004 × N counts or less is selected. The constant current In in the negative direction is passed through the capacitor for a predetermined time T,
When a predetermined constant current Ip in the positive direction is passed through the capacitor after the predetermined time T has elapsed, and the time until the voltage across the capacitor becomes zero is Tp,
| Ip | = | In |
The pulse count number of the predetermined time T is N,
When the pulse count of time Tp until the voltage across the capacitor becomes zero is Np,
The power transmission device according to claim 9, wherein a capacitor having an absolute value of a difference between the pulse count numbers Np and N of 0.004 × N counts or less is selected. The constant current In in the negative direction is passed through the capacitor for a predetermined time T,
When a predetermined constant current Ip in the positive direction is passed through the capacitor after the predetermined time T has elapsed, and the time until the voltage across the capacitor becomes zero is Tp,
| Ip | = | In |
The pulse count number of the predetermined time T is N,
When the pulse count number of time Tp until the voltage across the capacitor becomes zero is Nn,
The power transmission device according to claim 9, wherein a capacitor having an absolute value of a difference between Np and Nn of 0.003 × N counts or less is selected. When the effective series resistance of the power transmission coil is Ri and the effective series resistance of the capacitor is Rc,
The power transmission device according to any one of claims 1 to 11, wherein the power transmission unit transmits the power at a minimum frequency that satisfies Ri> Rc.   The power transmission device according to any one of claims 1 to 12, wherein the one coil is used for at least one of the power transmission coil or the power reception coil, and the two coils cannot be separated. Furthermore, it includes power conversion means for converting DC power into AC power,
When the output frequency of the power conversion means is fa (Hz),
The power transmission device according to claim 1, wherein the fa is set to a frequency less than the f1. Furthermore, the effective series resistance of the one coil when both ends of the other coil facing the one coil are opened is Rn (Ω),
When the highest frequency satisfying Rs> Rn ≧ Rw is f2 (Hz),
The power transmission device according to claim 2, wherein the fa is set to a frequency lower than the f2. Furthermore, the thermal resistance of the one coil is θi (° C./W),
The allowable operating temperature of the one coil is Tw (° C.),
The ambient temperature of the place where the one coil is installed is Ta (° C.),
When the alternating current flowing through the one coil when transmitting electric power is Ia (A),
In the fa,
Rw ≦ (Tw−Ta) / (Ia ² × θi),
The power transmission device according to claim 2, wherein power is transmitted from the power transmission unit to the power reception unit so that the one coil satisfies the following relationship. The power transmission device comprising the power transmission unit of the power transmission device according to claim 1, wherein the capacitor is connected to at least the power transmission coil of the power transmission unit,
The power transmission unit is a power transmission device of a power transmission device that transmits power to the power reception unit. The power receiving device comprising a power receiving unit of the power transmission device according to claim 1, wherein the capacitor is connected to at least the power transmission coil of the power transmission unit,
The power reception unit is a power reception device of a power transmission device that receives power from the power transmission unit.

Images