JP2004165612A - Variable inductor, oscillator including same inductor, radio terminal with the oscillator, amplifier including the inductor, and radio terminal provided with the amplifier - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a variable inductor, wherein its electrical characteristics are proper, and the reductions in its size and cost are facilitated, and is adapted to an integrated circuit. <P>SOLUTION: The inputted signal from a signal input terminal 11 is distributed among a plurality of signal paths 13a-13n, by a distributor 12, constituted by using active elements, and the distributed signals are fed to a plurality of inductors 14a-14n so disposed that at least one selected from among the inductors is mutually coupled to at least one of the other inductors. The distribution ratio of the distributor 12 is so varied that there is varied the inductance of at least one inductor 14a of the inductors 14a-14n, which is observed from variable-inductor terminals 15, 16 present at both its ends. <P>COPYRIGHT: (C)2004,JPO

Description

[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to a variable inductor, and more particularly to a variable inductor that changes inductance by using a plurality of inductors mutually coupled to a circuit including active elements.
[0002]
In addition, the present invention relates to an oscillator and a radio terminal using an inductor, and also relates to an improvement in a circuit design technique of the amplifier and the radio terminal and a method of varying the gain of the amplifier.
[0003]
[Prior art]
Generally, in order to change the characteristics of an electronic circuit, the characteristics of active elements or the values of passive elements included in the circuit are changed. Regarding the active element, the characteristics of the active element can be changed by changing the bias voltage applied to the active element. As for passive elements, a variable resistance element can be easily realized by using a passive element, for example, an on-resistance of a MOSFET, and a variable capacitance element can be easily realized by using a PN junction.
[0004]
Regarding inductors as passive elements, it is generally considered difficult to vary inductance while maintaining good characteristics. Non-Patent Document 1 (ELECTRONICS LETTERS 2nd January, Vol. 28, No. 1, pp. 78-80, 1992) discloses a method of configuring an inductor using an active element and making the inductance variable. However, since an active element is used for the inductor, there is a problem that noise and distortion characteristics are poor.
[0005]
Therefore, various techniques have been proposed in which the inductance is made variable without forming the inductor with an active element. However, both proposals have problems in practical use. For example, Patent Literature 1 (Japanese Patent Laid-Open No. 8-162331) discloses a method of inserting a switch in the middle of an inductor and changing the inductance by turning the switch on and off. In this method, there is a problem that the performance of the variable inductor deteriorates due to the on-resistance of the switch.
[0006]
Japanese Patent Laid-Open No. 2000-223317 discloses a method of physically changing the shape of an inductor by a laser beam. This method has a problem in that since it is necessary to physically adjust the inductor after it is manufactured, the manufacturing cost increases, and it is difficult to change the inductance while the circuit is operating.
[0007]
Patent Literature 3 (Japanese Patent Application Laid-Open No. 7-320942) proposes a technique for forming a variable inductor by using mutual coupling of a plurality of inductors. In this method, the shape of the inductor is physically changed in order to change the mutual coupling coefficient. Therefore, there is a problem in that the circuit constituting the variable inductor is reduced in size and cost.
[0008]
Further, DRPehlke et al., In US Pat. No. 5,994,985, use a directional coupler to separate an input signal into two and use a signal flowing through two mutually coupled inductors. We have proposed a technique to control the amplitude and phase of the to change the inductance. However, since the directional coupler is generally not suitable for integration, there is a problem that it is difficult to realize a variable inductor by an integrated circuit.
[0009]
By realizing a variable inductor, the oscillation frequency (f) of a voltage controlled oscillator (VCO: Voltage Controlled Oscillator) including an LC resonance circuit (inductor: inductance L, capacitor: capacitance C) can also be controlled. The oscillation frequency (f) of the voltage controlled oscillator including the LC resonance circuit is generally expressed by f = 1 / [2π (LC) 1/2]. The frequency (f) is controlled. However, in the conventional LC resonance circuit, since it is difficult to realize a variable inductor as described above, generally, the capacitance C is made variable, for example, by changing the reverse bias application voltage to the PN junction diode. The capacitance C is made variable, and the oscillation frequency is changed by changing the capacitance.
[0010]
When such a voltage controlled oscillator is formed on a semiconductor substrate as an integrated circuit, that is, when it is formed into an IC, parasitic capacitances, for example, a parasitic capacitance of an inductor, a drain parasitic capacitance of a MOS transistor, and a gate parasitic of a MOS transistor. It is unavoidable that such a parasitic capacitance occurs due to the occurrence of capacitance and the like, and there is a problem that the parasitic capacitance reduces the fluctuation width of the variable capacitance C of the LC resonance circuit. For example, assuming that the capacitance C variation is ΔC, the design variation rate is assumed to be ΔC / C, but the parasitic capacitance is added to the denominator as a non-variable part. Therefore, in practice, there is a problem that the variation rate is reduced to ΔC / (C + parasitic capacitance). Although it depends on the circuit design, if the capacitance C and the parasitic capacitance are approximately the same, the rate of change will be reduced to about 1/2.
[0011]
Due to the existence of such parasitic capacitance, the ratio of the variable capacitance C to the capacitance value of the LC resonance circuit must be reduced. Therefore, as compared with the change rate of the variable capacitor C, the change in the capacitance value in the LC resonance circuit is reduced, and as a result, the variable range of the oscillation frequency is narrowed.
[0012]
However, today, the frequency bands used in mobile phones, wireless LAN devices, and the like are expanding, and a plurality of frequency bands may be handled by a single device. I have. From such a viewpoint, realization of a variable inductor is desired.
[0013]
Further, the variable inductor is also applicable to an amplifier having an inductor. For example, in an amplifier including a degeneration inductor, when the inductance is reduced, the gain and noise characteristics of the amplifier are improved, but the distortion characteristics are deteriorated. Conversely, increasing the inductance improves the distortion characteristics, but degrades the gain and noise characteristics. Since there is a trade-off relationship in this way, the value of the inductance is determined so that desired characteristics are obtained when designing the amplifier.
[0014]
In order to realize low distortion characteristics while maintaining high gain and low noise characteristics, it is common to increase the amount of current. In an amplifier used in a receiver of a wireless terminal, characteristics required for the amplifier vary depending on the magnitude of a received signal. In general, when the received signal is small, it is important to amplify the signal with low noise. Therefore, the amplifier is required to have good gain and noise characteristics. On the other hand, when the received signal is large, good distortion characteristics are required.
[0015]
In a conventional amplifier having a degeneration inductor, the inductance is fixed, and the amount of supplied current is controlled in order to change the characteristics of the amplifier. That is, when it is desired to improve the distortion characteristics, the current is controlled so as to increase the amount of supplied current. However, increasing the amount of current to change the characteristics of the amplifier has the problem of increasing power consumption.
[0016]
Further, Non-Patent Document 4 discloses a circuit example of a variable gain amplifier. In this variable gain amplifier, the first-stage grounded emitter circuit constituted by the first transistor Q1 is always operated, and the gain is switched by switching the second to fourth transistors Q2 to Q4 constituting the next-stage grounded base circuit. Has been realized. Since the first transistor Q1 operates, there is no large change in the input impedance even when the gain is switched, but a constant current is always consumed, and the distortion characteristics are almost constant.
[0017]
However, the conventional circuit disclosed in Non-Patent Document 4 has a problem that a large current is consumed even when the gain is low, and the distortion characteristics are almost the same as those when the gain is high. Basically, an amplifier stage with high gain and good distortion characteristics requires a certain amount of current consumption, but if the gain is not so high or the signal is attenuated, a large amount of current is consumed. Thus, it is possible to realize an amplification stage having good distortion characteristics. However, when a plurality of different amplification stages are switched and used, there is a problem that the input impedance changes.
[0018]
[Patent Document 1]
JP-A-8-162331
[0019]
[Patent Document 2]
JP 2000-223317 A
[0020]
[Patent Document 3]
JP-A-7-320942
[0021]
[Patent Document 4]
U.S. Pat. No. 5,994,985
[0022]
[Non-patent document 1]
ELECTRONICS LETTERS 2nd January, Vol 28, No.1, pp 78-80, 1992
[0023]
[Non-patent document 2]
MTMurphy, "Applying the Series Feedback Technique to LNA Design," MICROWAVE JOURNAL, Nov., pp.143-152, 1989
[0024]
[Non-Patent Document 3]
JCRudell, et al., "A 1.9GHz Wide-Band IF Double Conversion CMOS Integrated Receiver for Cordless Telephone Applications," ISSCC97, pp.304-305, 1997
[0025]
[Non-patent document 4]
Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications KLFong, “Dual-Band High-Linearity Variable-Gain Low-Noise Amplifier for Wireless Applications,” IEEE ISSCC99, pp224-225, 1999
[0026]
[Problems to be solved by the invention]
As described above, the variable inductor according to the related art has a problem in electrical characteristics, and has a problem that miniaturization, cost reduction, and integration into an integrated circuit are difficult.
[0027]
In wireless terminals such as mobile phones, there is a strong demand for adaptive characteristic changes in amplifier characteristics according to received signal levels, while low power consumption is also strongly demanded. In an amplifier using a fixed inductor for degeneration, the only way to improve the distortion characteristics is to increase the amount of current, resulting in an increase in power consumption.
[0028]
The conventional circuit disclosed in Non-Patent Document 4 has a problem that a large current is consumed even when the gain is low, and the distortion characteristics are almost the same as those when the gain is high. Basically, an amplifier stage with high gain and good distortion characteristics requires a certain amount of current consumption, but if the gain is not so high or the signal is attenuated, a large amount of current is consumed. Thus, it is possible to realize an amplification stage having good distortion characteristics. However, when a plurality of different amplification stages are switched and used, there is a problem that the input impedance changes.
[0029]
SUMMARY OF THE INVENTION The present invention has been made in view of the circumstances described above, and has as its object to provide a variable inductor suitable for an integrated circuit, which has good electric characteristics, is easy to reduce in size and costs, and is easy to reduce. Is to provide.
[0030]
Another object of the present invention is to provide an oscillator that suppresses the influence of the parasitic capacitance on the variable range of the oscillation frequency.
[0031]
Another object of the present invention is to provide an amplifier capable of changing the characteristics of the amplifier without increasing power consumption.
[0032]
Still further, an object of the present invention is to allow a certain amount of current to flow to operate an amplification stage having high gain and good distortion characteristics when the desired gain is high, and to reduce the amplification when the desired gain is low. It is an object of the present invention to provide a variable gain amplifier that can operate an amplification stage having good distortion characteristics with current consumption and reduce a change in input impedance when the amplification stage is switched.
[0033]
[Means for Solving the Problems]
According to the invention,
A signal input terminal to which an input signal is input,
A distributor that includes first and second active elements, and that changes a distribution ratio of first and second currents flowing through the first and second active elements;
An output terminal from which the first and second currents are output;
A first inductor through which the first current flows; and
A second inductor through which the second current flows, the second inductor being interconnected to the first inductor;
There is provided a variable inductor comprising:
[0034]
According to the invention,
A voltage controlled oscillation circuit having a first inductor;
A frequency control circuit having a first inductor and a second inductor that is electromagnetically coupled to each other, and supplies a control current to the second inductor. A frequency control circuit that controls the oscillation frequency of the voltage-controlled oscillation circuit by changing the inductance value of the inductor;
An oscillator comprising:
[0035]
Further, according to the present invention,
An amplifier circuit having a first inductor;
A control unit that includes a second inductor that is electromagnetically coupled to the first inductor and supplies a control current to the second inductor. The control unit changes the control current to change the control current of the first inductor. A control unit that changes an inductance value and changes an amplification characteristic of the amplification circuit;
An amplifier, comprising:
[0036]
BEST MODE FOR CARRYING OUT THE INVENTION
A variable inductor according to an embodiment of the present invention, an oscillator having the variable inductor incorporated in the circuit thereof, a radio terminal having the oscillator, an amplifier having the variable inductor incorporated in the circuit, and the amplifier with reference to the drawings. A wireless terminal provided with is described in detail.
[0037]
In the following description, an example will be described in which an FET is used as a transistor which is an active element used in a distributor of a variable inductor circuit. However, a variable inductor circuit can be realized by using a bipolar transistor instead of an FET. .
[0038]
<Variable inductor>
(1st Embodiment)
First, a variable inductor according to a basic embodiment of the present invention will be described. 1 and 2 show a circuit configuration of a variable inductor according to one embodiment of the present invention. An input signal (Input) input to the signal input terminal 11 is distributed to a plurality of signal paths 13a, 13b,..., 13n by a distributor 12 configured using active elements. , 13n are respectively inserted with inductors 14a, 14b,..., 14n. The inductors 14a, 14b,..., 14n are made of, for example, spiral-shaped conductors, and are arranged close to each other so as to be mutually coupled.
[0039]
Each of the inductors 14a, 14b,..., 14n generates a magnetic flux depending on the magnitude of the input signal, and the magnetic flux also acts on another adjacent inductor to form a mutual coupling with the other inductor. Occurs. Therefore, each of the inductors 14a, 14b,..., 14n has a self-inductance caused by a magnetic flux generated by itself and a mutual inductance determined by a magnetic flux generated by another inductor mutually coupled to each other. For example, the inductance of the inductor 14a, that is, the inductance between the terminals 15 and 16 (referred to as a variable inductor terminal) at both ends of the inductor 14a is equal to the self-inductance Lsa of the inductor 14a and the inductor 14a and the other inductors 14b,. , Man. Here, depending on the direction of the mutually coupled inductors (the direction in which both the current and the winding direction are considered), the mutually coupled inductors also have a relationship in which the magnetic flux generated from the inductor reinforces each other. Can be set to weaken each other. Therefore, the inductance La between the variable inductor terminals 15 and 16 can be made larger or smaller than the self-inductance Lsa.
[0040]
.., 13n are supplied from the distributor 12 to the inductors 14a, 14b,. Here, by adjusting the distribution ratio of the signal levels distributed from the distributor 12 to the inductors 14a, 14b,... 14n, the mutual inductances Mab,..., Man, ie, the mutual coupling of the inductors 14a, 14b,. Can control the amount of magnetic flux. Therefore, the inductance between the variable inductor terminals 15 and 16 can be set to a desired value.
[0041]
The distributor 12 is configured by an active element such as a transistor as described later. Therefore, unlike the conventional variable inductor circuit using the directional coupler, the distributor 12 shown in FIGS. 1 and 2 can be easily integrated. In the circuit shown in FIG. 1, the distributor 12 is connected between the signal input terminal 11 and one end of each of the inductors 14a, 14b,. In the circuit shown in FIG. 2, one end and the other end of inductors 14a, 14b,.
[0042]
Next, some more specific embodiments of the variable inductor shown in FIGS. 1 and 2 will be described.
[0043]
(Second embodiment)
3 and 4 show a variable inductor according to an embodiment in which the distributor 12 includes a transistor circuit whose source is grounded. The circuit according to this embodiment corresponds to a specific circuit example of the basic circuit configuration shown in FIG. An input signal from the signal input terminal 11 is amplified by FETs whose sources are grounded, for example, MOSFETs (hereinafter simply referred to as transistors) 21a and 21b, distributed to two signal paths, and inserted into the signal paths. It is supplied to the inductors 14a and 14b. The gate terminals of the transistors 21a and 21b are connected to the signal input terminal 11, the source terminals are connected to one ends of the inductors 14a and 14b, and the drain terminals are connected to the output terminals 22a and 22b of the variable inductor. In this variable inductor, signal currents amplified by the MOSFETs 21a and 21b are output from output terminals 22a and 22b of the variable inductor.
[0044]
The other ends of the inductors 14a and 14b are connected to one ends of the current sources 23a and 23b and one ends of the capacitors 24a and 24b, and are grounded with respect to an AC component (high-frequency component). The other ends of the current sources 23a and 23b and the other ends of the capacitors 24a and 24b are connected to the ground. Among the currents flowing through the inductors 14a and 14b, a DC component flows through the current sources 23a and 23b, and an AC component (high-frequency component) is bypassed by the capacitors 24a and 24b.
[0045]
In the variable inductors shown in FIGS. 3 and 4, the black dots near the symbols representing the inductors 14a and 14b indicate the starting position of the winding of the inductor. It is assumed that magnetic fluxes generated in the same line direction are in phase.
[0046]
When the inductors 14a and 14b are in the same direction as shown in FIG. 3, the inductors 14a and 14b are generated in directions in which magnetic fluxes reinforce each other. Therefore, assuming that the self-inductance of the inductor 14a alone is Lsa, the self-inductance of the inductor 14b alone is Lsb, and the mutual coupling coefficient between the inductors 14a and 14b is kab, the effective inductance of the inductor 14a considering the mutual coupling, that is, The inductance La between the variable inductor terminals 15 and 16 is represented by the following equation (1).
[0047]
La = Lsa + kab · Lsb (1)
Assuming that the mutual inductance between the inductors 14a and 14b is Mab, Equation (1) is represented by Equation (2).
[0048]
La = Lsa + Mab (2)
On the other hand, when the inductors 14a and 14b are in the opposite directions as shown in FIG. 4, the inductors 14a and 14b are generated in a relationship in which the magnetic flux weakens each other. This inductance La is expressed by equation (3) or (4).
[0049]
La = Lsa-kab · Lsb (3)
Or
La = Lsa-Mab (4)
Here, the mutual coupling coefficient kab, that is, the mutual inductance Mab is determined by the physical arrangement of the inductors 14a and 14b, the sizes of the transistors 21a and 21b, the current values Ia and Ib of the current sources 23a and 23b, and the like. Therefore, without changing the physical arrangement of the inductors 14a and 14b, the value of the inductance La can be made variable by adjusting the current values Ia and Ib and the size of the transistors 21a and 21b.
[0050]
For example, if the current source 23b is a variable current source capable of controlling the current value Ib by an external control signal, and the current value Ib is continuously changed, the mutual inductance Mab is changed accordingly. Thereby, the inductance La between the variable inductor terminals 15 and 16 is continuously changed. When the current source 23b is turned on / off, the inductance La can be binaryly switched between (Lsa + Mab) or (Lsa-Mab) and Lsa.
[0051]
In the embodiment shown in FIGS. 3 and 4, the common source circuit using FETs is used for the distributor 12, but a common emitter circuit using bipolar transistors can be used. In the case of a common emitter circuit, the gate terminal, the drain terminal, and the source terminal of the FET may be replaced with the base terminal, the collector terminal, and the emitter terminal of the bipolar transistor. Further, in the circuits shown in FIGS. 3 and 4, two inductors are used. However, the same configuration as that of the present embodiment can be applied to a circuit using three or more inductors.
[0052]
Further, as a modified example of this embodiment, the distributor 12 is configured by a plurality of transistors each of which has a gate terminal or a base terminal commonly connected to the signal input terminal 11 via at least one of the plurality of inductors. May be. As another modified example of the present embodiment, each gate terminal or base terminal is commonly connected to the signal input terminal 11, and the drain terminal or the collector terminal is distributed by a plurality of transistors connected to one ends of a plurality of inductors. The container 12 may be configured.
[0053]
(Third embodiment)
FIG. 5 shows a variable inductor according to a third embodiment of the present invention using a transistor circuit having a gate grounded to a distributor 12. The signal input terminal 11 is connected to the source terminals of at least one of the first transistors 31a, 31b,..., 31n and the source terminals of the plurality of second transistors 32a, 32b,. The first transistors 31a, 31b,..., 31n have drain terminals connected to one end of the first inductor 14a, and gate terminals connected to control signal input terminals 33a, 33b,. The drain terminals of the second transistors 32a, 32b, ..., 32c are commonly connected to one end of the second inductor 14b, and the gate terminals are connected to the control signal input terminals 34a, 34b, ..., 34n, respectively.
[0054]
The control signals φ33a, φ33b,..., Φ33n are input to the control signal input terminals 33a, 33b,. , 34c receive control signals φ34a, φ34b,..., Φ34n, respectively. When the control signals φ33a, φ33b,..., Φ33n and φ34a, φ34b,..., Φ34n are changed in a binary manner, the distribution ratio of the signal level to the inductors 14a, 14b, that is, the ratio of the current flowing through the inductors 14a, 14b, Be changed. Thereby, the mutual inductance Mab between the inductors 14a and 14b is changed, and as a result, the effective inductance La (the inductance between the terminals 15 and 16) of the inductor 14a can be changed.
[0055]
The distribution ratio of the signal level to the inductor 14a is determined by the number of transistors that are turned on by the control signals φ33a, φ33b,..., Φ33n among the transistors 31a, 31b,. Similarly, the distribution ratio of the signal level to the inductor 14b is determined by the number of transistors that are turned on by the control signals φ34a, φ34b,..., Φ34n among the transistors 32a, 32b,.
[0056]
The control signals φ33a, φ33b,..., Φ33n and φ34a, φ34b,..., Φ34n are analog signals, and the current flowing through the transistors 31a, 31b,. Thus, the inductance La between the terminals 15 and 16 may be changed continuously.
[0057]
In FIG. 4, a plurality of first transistors 31a, 31b,..., 31n are connected to the inductor 14a. However, if the variable range of the inductance may be small, the number of the first transistors may be one. Similarly, a plurality of second transistors 32a, 32b,..., 32n are connected to the inductor 14b. However, if the variable range of the inductance may be small, one second transistor may be used.
[0058]
In the variable inductor according to this embodiment, a common gate circuit using FETs is used for the distributor 12, but a common base circuit using bipolar transistors may be used. In the case of the grounded base circuit, the gate terminal, the drain terminal and the source terminal of the FET may be replaced with the base terminal, the collector terminal and the emitter terminal of the bipolar transistor, respectively.
[0059]
Further, as a modification of the present embodiment, the distributor 12 has a source terminal or an emitter terminal connected to the signal input terminal 11 via at least one first inductor, and a gate terminal or a base terminal connected to the control signal input terminal. At least one first transistor, a source terminal or an emitter terminal connected to the 11 signal input terminal via the first inductor, and a drain terminal or a collector terminal interconnected with the first inductor. And at least one second transistor having a gate terminal or a base terminal connected to the control signal input terminal.
[0060]
(Fourth embodiment)
FIG. 6 shows a variable inductor according to a fourth embodiment of the present invention using a source follower circuit for the distributor 12. The gate terminals of a plurality of (two in the example of FIG. 4) transistors 41a and 41b are connected to the signal input terminal 11. The drain terminals of the transistors 41a and 41b are connected to the power supply Vdd, which is a constant potential point, and the source terminals are connected to one ends of the inductors 14a and 14b, respectively. Current sources 43a and 43b are further connected to the source terminals of the transistors 41a and 41b. Therefore, the transistors 41a and 41b operate as a source follower circuit.
[0061]
Here, similarly to the second embodiment, the current source 43b is a variable current source capable of controlling the current value Ib by an external control signal, and if the current value Ib is continuously changed, the inductor The mutual inductance Mab between the terminals 14a and 14b is changed, and the inductance La between the terminals 15 and 16 changes continuously. When the current source 43b is turned on / off, the inductance La is binary-switched between (Lsa + Mab) or (Lsa-Mab) and Lsa, where Lsa is the self-inductance of the inductor 14a.
[0062]
In this embodiment, an emitter follower circuit using FETs is used for the distributor 12, but it is apparent that an emitter follower circuit using bipolar transistors can be used.
[0063]
Further, as a modification of the present embodiment, each gate terminal or base terminal is connected to the signal input terminal 11 via an inductor, and each drain terminal or collector terminal is distributed by a plurality of transistors connected to a constant potential point. The container 12 may be configured.
[0064]
(Fifth embodiment)
FIG. 7 shows a variable inductor according to a fifth embodiment of the present invention in which a cascode connection circuit is used for the distributor 12. Gate terminals of the first and second transistors 51 and 52 are connected to the signal input terminal 11. Source terminals of the transistors 51 and 52 are connected to current sources 53 and 54, respectively. The drain terminal of the first transistor 51 is connected to the source terminal of the third transistor 55, and the drain terminal of the second transistor 52 has a plurality of (two in the example of FIG. 5) fourth transistors 56a and 56b. Are commonly connected to the source terminal of That is, the transistor 51 and the transistor 55 are cascode-connected, and the transistor 52 is cascode-connected to the transistors 56a and 56b.
[0065]
The drain terminals of the third transistor 55 and the fourth transistors 56a, 56b are connected to one ends of the inductors 14a, 14b, 14c, respectively. The gate terminal of the third transistor 55 is connected to the control signal input terminal 58, and the gate terminals of the fourth transistors 56a and 56b are connected to the control signal input terminals 57a and 57b, respectively. As indicated by the black circles near the symbols representing the inductors 14a, 14b, and 14c, the inductor 14b is oriented in such a manner that the magnetic flux is strengthened with respect to the inductor 14a, and the inductor 14c is oriented in such a manner that the magnetic flux is weakened with each other. Each is arranged.
[0066]
Now, in a state where the control signal φ is input to the control signal input terminal 58 and the transistor 55 is turned on, the control signal φ + is input to the control signal input terminal 57a, and the control signal φ− is input to the control signal input terminal 57b. Accordingly, when the transistor 56a connected to the inductor 14b is turned on and the transistor 56b connected to the inductor 14c is turned off, the inductance La between the both ends 15, 16 of the inductor 14a is larger than the self inductance Lsa of the inductor 14a. Become. Conversely, when the control signal φ− is input to the control signal input terminal 57a and the control signal φ + is input to the control signal input terminal 57b, and the transistor 56a is turned off and the transistor 56b is turned on, both ends of the inductor 14a The inductance La between 15, 15 is smaller than the self-inductance Lsa.
[0067]
When the cascode connection circuit is used for the distributor 12 as described above, a signal is selectively supplied to the inductors 14b and 14c by turning on and off the fourth transistors 56a and 56b cascode-connected to the second transistor 52. As a result, the inductance La between the terminals 15 and 16 can be increased or decreased.
[0068]
In this embodiment, a cascode connection circuit using FETs is used for the distributor 12, but it is apparent that a cascode connection circuit using bipolar transistors can be used.
[0069]
(Sixth embodiment)
FIG. 8 shows a variable inductor according to a fifth embodiment of the present invention. The signal input terminal 11 is connected to a gate terminal of a transistor 61 included in the amplifier 60. A current source 63 and an AC (high frequency) bypass capacitor 64 are connected to the source terminal of the transistor 61. One end of the inductor 14a and the input terminal of the buffer circuit 62 are connected to the drain terminal of the transistor 61, and one end of the other inductor 14b is connected to the output terminal of the buffer circuit 62.
[0070]
The buffer circuit 62 is configured by a circuit that can be formed on an integrated circuit such as a common source circuit or a source follower circuit, and has a variable gain. When the gain of the buffer circuit 62 is changed, the distribution ratio of the signal level to the inductors 14a and 14b is changed as in the previous embodiments, so that the mutual inductance is changed and the inductance between the variable inductor terminals 15 and 16 is changed. be able to.
[0071]
In this embodiment, an FET is used for the transistor 61 for the distributor 12, but a bipolar transistor may be used as in the previous embodiments.
[0072]
(Seventh embodiment)
In the embodiments described so far, a case has been described in which all of the distributors 12 have a single-phase circuit configuration, but the distributor 12 may be configured with a differential circuit. FIG. 9 shows an example in which the distributor 12 has a differential circuit configuration as the variable inductor according to the seventh embodiment of the present invention. In the circuit shown in FIG. 9, a differential amplifier cascode-connected to the distributor 12 is used.
[0073]
The gate terminals of the differential pair transistors 71a and 71b and the gate terminals of the differential pair transistors 72a and 72b are connected to the differential input terminals 11a and 11b, respectively. The common source terminals of the differential pair transistors 71a and 71b are connected to a current source 78, and the drain terminals of the differential pair transistors 72a and 72b are connected to a current source 79, respectively. The drain terminals of the differential pair transistors 71a and 71b are connected to the source terminals of the transistors 73a and 73b, respectively. The drain terminals of the differential pair transistors 72a and 72b are connected to the source terminals of the transistors 74a and 74b and the sources of the transistors 75a and 75b. Connected to each terminal.
[0074]
Gate terminals of the transistors 73a, 73b, 74a, 74b, 75a, 75b are connected to control signal input terminals, respectively. The drain terminals of the transistors 73a and 73b are connected to the differential output terminals 22a and 22b, respectively, and to one ends of the inductors 76a and 76b, respectively. The drain terminals of the transistor 74a and the transistor 75b are commonly connected to one end of the inductor 77a, and the drain terminals of the transistor 74b and the transistor 75a are commonly connected to one end of the inductor 77b. The other ends of the inductors 76a, 76b, 77a, 77b are connected to a power supply Vdd, which is a constant potential point.
[0075]
The differential input terminals 11a and 11b receive signals Input + and Input- having phases opposite to each other. The input of the signals Input + and Input− having opposite phases is equivalent to the fact that the sign of the mutual inductance M12 between the inductors 76a and 77a and the sign of the mutual conductance M13 between the inductors 76b and 77b are inverted. Therefore, the control signals φ + and φ− can control the mutual coupling related to the transconductances M12 and M13 in a direction in which the magnetic fluxes cancel each other and in a direction in which the magnetic fluxes are strengthened, thereby changing the inductor. It is possible.
[0076]
FIG. 10 shows an example in which a distributor is formed using a differential circuit using bipolar transistors. Although there are differences in the transistors used, they are similar in principle to the circuit shown in FIG.
[0077]
As described above, according to the variable inductor according to the embodiment of the present invention, the inductor can be electromagnetically changed using the distributor that can be formed on the integrated circuit. In addition, such a variable inductor has good electrical characteristics, is easily reduced in size and cost, and is suitable for integration into an integrated circuit.
[0078]
<Oscillator with Variable Inductor and Wireless Terminal with This Oscillator>
Next, an oscillator incorporating the above-described variable inductor according to the present invention and a wireless terminal including the oscillator will be described.
[0079]
FIG. 11 is a block diagram schematically showing an oscillator according to an embodiment of the present invention.
[0080]
The oscillator shown in FIG. 11 has a core circuit unit (VCO core) 102 including an LC resonance circuit having an inductor L0, and an inductor L11 electromagnetically coupled to the inductor L0 with a coupling coefficient k1, and supplies the inductor L11. An oscillation frequency control unit 104-1 (Frequency Controller) capable of controlling the current to be supplied is provided.
[0081]
A control signal Control_1 is input to the oscillation frequency control unit 104-1, and at least one or both of the amplitude and the phase of the current flowing through the inductor L11 are changed according to the control signal Control_1. As a result, the inductance of the inductor L0 of the core circuit unit 102 coupled to the inductor L11 is changed, and the oscillation frequency is changed. For example, when a current flows in the inductor L11 in a direction that generates a magnetic field in a direction that reinforces the magnetic field generated in the inductor L0, the inductance value of the inductor L0 increases, and is proportional to the −1/2 power of the inductance value. The oscillation frequency decreases. Further, when a current in a direction weakening the magnetic field generated in the inductor L0 flows through the inductor L11, the value of the inductance of the inductor L0 decreases, and as a result, the oscillation frequency increases. If the current amplitude is controlled by making the current directions flowing through the inductor L0 and the inductor L11 the same, it is possible to control the increase or decrease of the inductor L0.
[0082]
Generally, the control range of the frequency obtained by making the capacitance C variable is about Cmax / Cmin = 2. On the other hand, in the method of changing the inductance, for example, assuming that the current flowing through the inductor L and the current flowing through the inductor L11 are the same and k = about 0.7, Lmin = (1-k) L and Lmax = (1 + k) L , Lmax / Lmin = about 6.
[0083]
Normally, when the oscillation frequency is changed by a variable capacitor, only a change in the range of 5 to 10% of the oscillation frequency can be obtained. However, in an oscillator including a variable inductor as shown in FIG. The oscillation frequency can be changed to a range of 100%.
[0084]
The inductor L11 is not limited to one, and a plurality of inductors L11 to L1n such as the inductor L11 may be provided as shown in FIG. In the circuit shown in FIG. 11, n oscillation frequency control units 104-n are provided, and the inductors L11 to L1n of the respective oscillation frequency control units are electromagnetically coupled with the inductance L of the VCO circuit 102 and the coupling coefficient (k1 to kn). Is bound to
[0085]
When only one inductor L11 is electromagnetically coupled to one oscillation frequency control unit 104-1, that is, the inductor L of the core circuit unit 102, the amplitude or phase of the current flowing through the inductor L11 is changed. Thus, the value of the inductance of the inductor L0 can be changed.
[0086]
In a circuit including a plurality of inductors L11 to L1n, the VCO circuit 102 can be controlled in various modes. For example, even if the same current as the current flowing through the inductor L11 has the same coupling coefficient as the inductor L11 and flows through the inductor L1n, this current is turned on / off and the number of inductors L1n through which the current flows is changed. The activated L1n that is electromagnetically coupled to the inductor L0 may be changed.
[0087]
Further, inductors L11 to L1n having different coupling coefficients k1 to kn are prepared, and these inductors L11 to L1n can be switched and used. Further, the current flowing through each of the inductors L11 to L1n is changed, so that the core circuit unit 102 can be more finely controlled.
[0088]
In FIG. 12, an oscillation signal (voltage signal) is input to an oscillation frequency control circuit 104 from a differential LC resonance type voltage controlled oscillation circuit 106 corresponding to the VCO core circuit 102, and the oscillation signal is subjected to voltage-current conversion. An example of a circuit supplied to the inductors L1 and L2 coupled to the inductors L01 and L02 of the core circuit unit 102 is shown.
[0089]
In the circuit shown in FIG. 12, one end of an inductor L01 is connected to a current source 105 having a power supply voltage Vdd, and the other end is connected to a variable capacitor VC1 including a drain of a MOS transistor T1 and a diode. The source of the MOS transistor T1 is grounded.
[0090]
Similarly, one end of an inductor L02 is connected to a current source 105 having a power supply voltage Vdd, and the other end is connected to a drain of a MOS transistor T2 and a variable capacitor VC2 including a diode. The source of the MOS transistor T2 is grounded.
[0091]
Note that the drain of the MOS transistor T1 is connected to the gate of the MOS transistor T2, and similarly, the drain of the MOS transistor T2 is connected to the gate of the MOS transistor T1.
[0092]
The capacitance control voltage Vctrl is supplied to the variable capacitors VC1 and VC2, and the capacitance of the variable capacitors VC1 and VC2 is determined. The resonance frequency is determined by the parallel connection (L1-VC1) and (L2-VC2) of the variable capacitors VC1, VC2 and the inductors L1, L2.
[0093]
Although variable capacitances VC1 and VC2 are used in FIG. 12, fixed capacitance capacitors may be used. When a fixed-capacitance capacitor is used, only the variation of the frequency control depending on the variable capacitors VC1 and VC2 disappears, and the operation does not change.
[0094]
This oscillator 106 outputs an output 1 (Output_1) from the drain of the MOS transistor T1 and an output 2 (Output_2) from the drain of the MOS transistor T2.
[0095]
In order to change the oscillation frequency of the output signal, the output 1 and the output 2 (Output_1, Output_2) are input to a current-voltage conversion circuit (VI Converter) 108 of the oscillation frequency control circuit 104 and are electromagnetically coupled to the inductor L01. The current flowing through the inductor L2, which is electromagnetically coupled to the inductor L1 and the inductor L02 by the coupling coefficient k2, is controlled.
[0096]
Here, k1 = 0.7, k2 = 0.7, and the inductance value is equal to L for each of the inductors (L01, L02, L1, L2), and a current of the same magnitude flows in each inductor. Assume.
[0097]
When the direction of the current between the electromagnetically coupled inductors is a direction that reinforces the magnetic field, the value of the inductance of the inductor L01 increases from L0 to 1.7L0.
[0098]
When a current flows in a direction to weaken the magnetic field, the inductance value of the inductor L01 decreases to 0.3L0.
[0099]
Therefore, if the lower oscillation frequency is near fLo = 2 GHz, the higher oscillation frequency is fHi = (0.3 / 1.7) − ／ · fLo, and an oscillation frequency of 4 GHz or more is obtained. You can see that you can do it.
[0100]
FIG. 13 is a specific example in which the voltage-current conversion circuit 108 shown in FIG. 12 is configured by cascode-connected transistors. The voltage control oscillation circuit 16 of the differential LC resonance type is the same as the circuit configuration shown in FIG. 12, and the same reference numerals are given in FIG. Hereinafter, the frequency control unit 104 will be described.
[0101]
The drain of the transistor T12 is connected to the inductor L1, and the source of the transistor T12 is connected to the drain of the transistor T11 whose output receives the output 1 (Output_1). The source of the transistor T11 is grounded via the current source 110. The source of the transistor T13 whose drain is connected to the inductor L2 is connected to the drain of the transistor T11. The control signal φ + is input to the gate of the transistor T12, and the control signal φ− is input to the gate of the transistor T13. That is, an inductor L1 is connected to a transistor T12 whose source is grounded via a current source and a transistor T11 whose output 1 (Output_1) is supplied to the gate is cascode-connected, and a control signal φ + is connected to the gate of the transistor T12. Supplied. An inductor L2 is connected to the transistor T13 that is cascode-connected to the transistor T11, and the control signal φ− is supplied to the gate of the transistor T13.
[0102]
Similarly, the drain of the transistor T15 is connected to the inductor L2, and the source of the transistor T15 is connected to the drain of the transistor T14 whose gate receives the output 2 (Output_2). The source of the transistor T14 is grounded via a current source. The source of the transistor T16 whose drain is connected to the inductor L1 is connected to the drain of the transistor T14. The control signal φ + is input to the gate of the transistor T15, and the control signal φ− is input to the transistor T16. That is, the inductor L2 is connected to T15, which is cascode-connected to the transistor T14 whose source is grounded and the output 2 (Output_2) is supplied to the gate via the current source, and the control signal φ + is supplied to the gate of the transistor T15. You. An inductor L1 is connected to the transistor T16 that is cascode-connected to the transistor T14, and the control signal φ− is supplied to the gate of the transistor T16.
[0103]
By changing the control signals φ + and φ−, for example, the directions of the currents flowing through the inductors L1 and L2 can be reversed. The amplitude of the current flowing through inductors L1 and L2 can be changed by appropriately setting the potentials of control signals φ + and φ−.
[0104]
By controlling the control signals φ + and φ− in this manner, the current flowing through the inductors L1 and L2 can be controlled, and as a result, the inductance values of the inductors L01 and L02 coupled to the inductors L1 and L2 can be controlled. Become. For example, when the current flowing through the inductor L01 (L02) and the current flowing through the inductor L1 (L2) are in phase, the value of the inductance of the inductor L01 (L02) increases, and as a result, the oscillation frequency decreases. When the current flowing through the inductor L01 (L02) and the current flowing through the inductor L1 (L2) are in opposite phases, the inductance value of the inductor L01 (L02) decreases, and as a result, the oscillation frequency increases.
[0105]
FIG. 14 shows an oscillator according to an embodiment in which a plurality of differential pairs are arranged.
[0106]
One end of the inductor L01 is connected to a current source having the power supply voltage Vdd, and the other end is connected to the drain of the MOS transistor T1 and the capacitor C1. The source of the MOS transistor T1 is grounded via the current source I0.
[0107]
Similarly, one end of the inductor L02 is connected to a current source having the power supply voltage Vdd, and the other end is connected to the drain of the MOS transistor T2 and the capacitor C2 including a diode. The source of the transistor T2 is grounded via the current source I0.
[0108]
The drain of the transistor T1 is connected to the gate of the transistor T2, and similarly, the drain of the transistor T2 is connected to the gate of the transistor T1.
[0109]
In the circuit shown in FIG. 14, the capacitances of the capacitors C1 and C2 are fixed, but variable capacitors may be used as in FIG.
[0110]
In the circuit shown in FIG. 14, the oscillation frequency control unit includes a plurality of differential pairs. An inductor L1n, for example, an inductor L11, having one end connected to the power supply voltage Vdd, is electromagnetically coupled to the inductor L01 of the VCO circuit 106 with a coupling coefficient k1n, for example, a coupling coefficient k11, and the other end is a MOS transistor T1n, for example, It is connected to the drain of transistor T11. A capacitor C1 (n), for example, a capacitor C11 is branched and connected between the drain of the transistor T1n and the inductor L1n. The output 1 (Output_1) of the VCO circuit is supplied to the gate of the transistor T1n, and the source of the transistor T1n is grounded via a variable current source In, for example, a variable current source I1.
[0111]
Similarly, an inductor L2n having one end connected to the power supply voltage Vdd, for example, the inductor L21, is electromagnetically coupled to the inductor L02 of the VCO circuit with a coupling coefficient k2n, for example, a coupling coefficient k21, and the other end has a MOS transistor T2n. For example, it is connected to the drain of the MOS transistor T21. Note that a capacitor C2n, for example, a capacitor C21 is branched and connected between the drain of the transistor T2n and the inductor L1n.
[0112]
The output 2 (Output_2) of the VCO circuit 106 is supplied to the gate of the transistor T2n, and the source of the transistor T2n is grounded via the variable current source In.
[0113]
The other ends of the capacitors C1n and C2n, for example, the capacitors C11 and C21, which are connected to the inductors, are connected.
[0114]
A plurality of such differential pairs are prepared, and the current values of the respective variable current sources I0 to In are changed or turned on / off to change the inductances of the inductors L01 and L02 of the VCO circuit 106 and control the oscillation frequency. can do.
[0115]
FIG. 15 shows an oscillator according to an embodiment using a variable phase shifter and a variable gain amplifier.
[0116]
VCO circuit 106 has the same circuit configuration as that shown in FIG. The output terminals are connected to buffer circuits 112 and 114 having a large impedance as viewed from the outside.
[0117]
Output signals (Output_1, Output_2) from the oscillator are input to variable phase circuits 116 and 118 of the oscillation frequency control circuit 104. The phase shift of the signal of the oscillation circuit is appropriately controlled by the variable phase shifters 120 and 122, and then the current is controlled via the variable gain amplifiers 124 and 126. The current is controlled by the inductor L01 and the coupling coefficient k1. Is supplied with a controlled phase / current amplitude value.
[0118]
Similarly, the current flowing through the inductor L2 coupled to the inductor L02 by the coupling coefficient k2 is similarly controlled.
[0119]
By appropriately controlling the phase shift and the current value, the inductances of the inductors L01 and L02 of the VCO circuit 106 can be made variable, so that the oscillation frequency can be changed and controlled.
[0120]
In the circuit example described above, the differential type has been described, but a single phase can be similarly applied. A circuit example in which a variable inductor is applied to a Colpitts oscillation circuit will be described with reference to FIG.
[0121]
The inductor L having one end connected to the power supply voltage Vdd has the other end connected to the drain of the MOS transistor T, and the source of the transistor T is grounded via the resistor R. The gate of the transistor T is grounded. An inductor L0 and a transistor T are branched from a connection point, and capacitors C1 and C2 are connected in series and grounded. The source of the transistor T1 is connected to the connection point between the capacitors C1 and C2.
[0122]
An output from a connection point between the inductor L0 and the capacitor C1 is input to a frequency control unit 128 (Frequency Control), connected to the power supply voltage Vdd, and coupled to the inductor L1 electromagnetically coupled to the inductor L0 with a coupling coefficient k. By controlling the flowing current, the oscillation frequency can be changed and controlled.
[0123]
In the embodiment described above, the MOS transistor is used. However, it goes without saying that a circuit having the same function can be realized by a bipolar transistor or the like.
[0124]
Further, the inductor constituting the variable inductor type oscillator described above, that is, the inductor L1 electromagnetically coupled with the coupling coefficient k to the inductor L0 on the LC resonance circuit side can be realized by various configurations. For example, as shown in FIG. 17, the spiral conductor forming the inductor L can be configured to be symmetrical with the spiral conductor forming the inductor L1. In this case, the respective conductors intersect at the center of FIG. 17, but the intersections are configured so that the conductors overlap with an insulating film interposed therebetween. it can.
[0125]
In order to realize a plurality of inductors L11... L1n that are electromagnetically coupled to the inductor L01 as shown in FIG. 18, one turn of L11. A configuration can be adopted.
[0126]
It is not necessary to arrange the conductor patterns on the same plane. For example, as shown in FIG. 19, L01, L11,... L1n can be stacked using a multilayer wiring. In FIG. 19, an insulating layer exists between the coil conductors constituting each of the inductors L01, L11... L1n. However, in FIG. 19, the insulating layer is omitted for simplification of the drawing. I have.
[0127]
By using an oscillator having a wide variable frequency range, a wireless terminal suitable for a plurality of communication systems having different frequency bands can be reduced in size. That is, in a conventional wireless terminal, it is necessary to mount an oscillator corresponding to each oscillation frequency in order to supply an oscillation frequency corresponding to each communication system. However, since the oscillator according to the embodiment of the present invention has a wide variable frequency range, it is possible to cope with a communication system in which one oscillator uses various oscillation frequencies. FIG. 20 shows a block diagram of a wireless terminal supporting various communication systems using communication frequencies of different communication frequencies, for example, 2 GHz, 2.4 GHz, and 5 GHz. Signals of the individual communication systems include antennas 130-1, 130-2, and 130-3, low noise amplifiers (Low Noise Amplifiers) 132-1, 132-2, and 132-3, and mixers (MIXs) 134-1 and 134. -2, 134-3, and is supplied to intermediate frequency processing units (IF) 136-1, 136-2, 136-3. Outputs from the intermediate frequency processing units (IFs) 136-1, 136-2, and 136-3 are supplied to a baseband processing unit. Note that signals of oscillation frequencies corresponding to the communication system are supplied to the individual mixers (MIX) 134-1, 134-2, and 134-3. In this example circuit, a local signal having a suitable frequency is supplied from the oscillator 138 to each wireless system. As the oscillator 138 for supplying the oscillation frequency, the oscillator according to the embodiment of the present invention described above is used. A signal of a desired frequency obtained by changing the inductance is switched by a switch and supplied to the mixer MIX of each communication system.
[0128]
As described above, according to the radio terminal according to the embodiment of the present invention, a plurality of communication systems having different frequency bands, for example, a mobile phone (PDC, W-CDMA, etc.), Bluetooth, without having a plurality of oscillators. (Registered trademark), wireless LAN (2.4 GHz, 5 GHz) and the like can be supported by one wireless terminal. According to the oscillator according to the embodiment of the present invention, a wide variable frequency range can be realized, and as a result, one oscillator can supply an oscillation frequency to a plurality of communication systems.
[0129]
<Amplifier Having Variable Inductor and Wireless Terminal Having This Amplifier>
Next, an amplifier including the variable inductor according to the embodiment of the present invention will be described.
[0130]
FIG. 21 is a block diagram schematically illustrating an amplifier according to an embodiment of the present invention.
[0131]
The amplifier shown in FIG. 21 includes an amplifier circuit 140 including an inductor La, and a control unit 142 including an inductor Lc electromagnetically coupled to the inductor L with a coupling coefficient k.
[0132]
The control unit 142 receives the input signal Input_1 to the amplifier circuit 140 and supplies a signal current to the inductor Lc. The characteristic of the output Output_1 of the amplifier is changed in accordance with the value of the inductance of the inductor La which is changed by the signal current supplied to the inductor Lc.
[0133]
In the circuit shown in FIG. 21, only one inductor Lc is shown, but the number is not limited to one, and a plurality of inductors Lc-1 to Lc-n may be provided. When a plurality of inductors Lc-1 to Lc-n are provided, various controls can be realized. For example, an inductor Lc-n having the same coupling coefficient and the same flowing current is prepared, and the current is turned on / off to change the number of Lc-1 to Lc-n through which the current flows, thereby changing the inductor Lc-n. It is also possible to respond by changing the activated Lc-n that is electromagnetically coupled.
[0134]
In addition, it is also possible to prepare inductors Lc-1 to Lc-n having different coupling coefficients k1 to kn and use them by switching. Further, it is also possible to change the current flowing through each of the inductors Lc-1 to Lc-n to perform finer control.
[0135]
The inductance of the inductor La can be apparently changed by changing the phase in the same manner as the control of the current amount including the ON / OFF control.
[0136]
The simplest control may be a system in which one inductor Lc is prepared and the current flowing through the inductor Lc is binary-controlled between ON and OFF. Also in this control system, the effect of varying the amplification characteristics can be sufficiently exhibited.
[0137]
A specific example of the differential amplifier will be described with reference to FIG.
[0138]
As shown in FIG. 22, the source of the MOS transistor M1 whose gate receives the input signal Input_1 is connected to the current source I1 via the degeneration inductor L1. The other end of the current source I1 is grounded. Further, an inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd.
[0139]
The input signal Input_2 is input to the gate of the MOS transistor M2 forming the differential pair, and the source of the MOS transistor M2 is connected to the degeneration inductor L2 in the same manner as the connection of the inductor L1, the MOS transistor M1, and the inductor L3. Is connected to the drain, the inductor L4 is connected, the inductor L2 is connected to the current source I1, and the inductor L4 is connected to the power supply potential Vdd.
[0140]
Output signals Output_1 and Output_2 are output from the drains of the MOS transistors M1 and M2, respectively. The control unit includes MOS transistors M3 and M4 for inputting input signals Input_1 and Input_2 of the differential amplifier to respective gates. An inductor L5 coupled to the inductor L1 by a coupling coefficient k1 is connected to a source of the MOS transistor M3, and is grounded via a variable current source I2. Similarly, the source of the MOS transistor M4 is connected to the variable current source I2 via an inductor L6 coupled to the inductor L2 by a coupling coefficient k2. The drains of the MOS transistors M3 and M4 are connected to the power supply potential Vdd.
[0141]
To change the characteristics of the output signal from the differential amplifier, for example, the distortion characteristics, the current value of the variable current source I2 is changed. It is assumed that the current change of the variable current source I2 is a binary value of current ON / OFF. When the variable current source I2 is ON, it is assumed that the pair of inductors L1 and L5 and the pair of inductors L2 and L6, which are electromagnetically coupled to each other, are configured so as to be oriented so as to mutually weaken the magnetic field. In such a circuit, when the variable current source I2 is ON, the inductance of the degeneration inductor looks small, so that high gain and low noise can be realized. That is, a high gain and low noise mode can be set. Further, when the variable current source I2 is OFF, a magnetic field in a direction to cancel the magnetic field generated in the inductors L1 and L2 is not generated, so that the degeneration inductance looks larger than when the variable current source I2 is ON. Will be. In this case, good distortion characteristics can be obtained. That is, it is possible to set the mode in which the distortion characteristic is emphasized.
[0142]
Therefore, when configuring the differential amplifier, the degeneration inductor may be designed with a relatively large inductance.
[0143]
In the above description, the current value of the variable current source I2 is controlled by two values, On / OFF. However, a desired amplifier characteristic can be obtained by changing the current value stepwise or continuously.
[0144]
FIG. 23 is a block diagram schematically showing another embodiment of the present invention.
[0145]
The amplifier shown in FIG. 23 includes a first amplifier circuit 150 including an inductor La, and a second amplifier 152 including an inductor Lc electromagnetically coupled to the inductor La with a coupling coefficient k.
[0146]
The second amplifier 152 receives the same input signal Input1 as the first amplifier 150 and performs a function of a control unit that controls the amplifier characteristics of the first amplifier 150, for example, a distortion characteristic. The output of the amplifier 152 is added to the output of the first amplifier 150 and output as an output signal Output_1.
[0147]
Similarly to the above embodiment, the second amplifier 152 corresponding to the control unit receives the input signal Input_1 to the amplifier circuit 150 and supplies a signal current to the inductor Lc. The characteristic of the output Output_1 of the first amplifier 150 changes according to the inductance value of the inductor L which changes according to the signal current supplied to the inductor LC. At this time, the output of the second amplifier 152 is also returned to the output Output_1, thereby increasing the amplification factor for the input signal Input_1.
FIG. 24 shows a specific circuit example of the differential amplifier shown in FIG.
[0148]
The source of the MOS transistor M1 whose gate receives the input signal Input_1 is connected to the current source I1 via the degeneration inductor L1. The other end of the current source I1 is grounded. Further, an inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd.
[0149]
The input signal Input_2 is input to the gate of the MOS transistor M2 forming the differential pair, and the degeneration inductor L2 is connected to the source of the MOS transistor M2 in the same manner as the connection of the inductor L1, the transistor M1, and the inductor L3. The drain is connected to an inductor L4, the inductor L2 is connected to a current source I1, and the inductor L4 is connected to a power supply potential Vdd.
[0150]
Output signals Output_1 and Output_2 are output from the drains of the MOS transistors M1 and M2, respectively. The second amplifier 152 corresponding to the control unit includes a MOS transistor M3 and a MOS transistor M4 that input the input signals Input_1 and Input_2 of the differential amplifier to their gates, respectively.
[0151]
An inductor L5 coupled to the inductor L1 by a coupling coefficient k1 is connected to a source of the MOS transistor M3, and is grounded via a variable current source I2. Similarly, the source of the MOS transistor M4 is connected to the variable current source I2 via an inductor L6 coupled to the inductor L2 by a coupling coefficient k2.
[0152]
The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M1, and the current from the variable current source I2 flowing through the MOS transistor M3 is added to the output signal Output_1. Similarly, the drain of the MOS transistor M4 is connected to the drain of the MOS transistor M2, and the current from the variable current source I2 flowing through the MOS transistor M4 is added to the output signal Output_2.
[0153]
By employing such a circuit configuration, the current flowing through the control unit can be used as the output of the amplifier circuit, and the current utilization efficiency of the amplifier as a whole increases.
[0154]
For example, if the current flowing through a normal amplifier is about 5 mA, depending on the design, it may be necessary to flow a current up to about 10 mA in some cases in order to realize low distortion characteristics. On the other hand, in the circuit shown in FIG. 24, when realizing high gain and low noise, when setting such that I1 = I2 = 2.5 mA, a characteristic equivalent to a case where 5 mA flows in a normal amplifier is obtained. We can expect to get. On the other hand, when realizing low distortion characteristics, low distortion characteristics can be realized by the effect of a large degeneration inductor even when the current source I2 is turned off and 2.5 mA flows to the current source I1.
[0155]
In the above-described circuit, the characteristics of the amplifier can be made variable, but the input impedance of the amplifier also changes accordingly. It is generally undesirable for the input impedance of the amplifier to change. Therefore, it is preferable to provide an input impedance control section at the input section of the amplifier.
[0156]
A circuit including a variable resistor as an input impedance control unit will be described with reference to FIG.
[0157]
In the circuit shown in FIG. 25, a variable resistor Rv is inserted between the input terminal Input_1 and the input terminal Input_2. The circuit shown in FIG. 25 has the same configuration as that of FIG. 24 except for the variable resistor Rv.
[0158]
The variable resistor Rv may be configured using an FET as shown in FIG. 26A, or may be a combination of a fixed resistor and a switch as shown in FIG.
[0159]
In the circuit shown in FIG. 25, as shown in FIG. 27, the input impedance of the amplification stage is represented by the following equation (5).
[0160]
Zin = L gm / C + j (ωL−1 / ωC). . . (5)
In this equation (5), when a bias current is flowing, there is a real part defined by the first term of the equation (5). However, when the bias current is turned off, gm = 0, so gm = 0. The input impedance has no real part. Therefore, in order to make the input impedance when the bias current is turned off close to the input impedance when the bias current flows, it is necessary to compensate the real part of the impedance. For this purpose, the real part of the impedance is compensated by using the aforementioned variable resistor Rv.
[0161]
To simplify the description, the resistance value of the variable resistor Rv is binary, that is, open / ON. In the circuit shown in FIG. 25, it is assumed that the inductor L1 and the inductor L5 and the inductor L2 and the inductor L6 are arranged in directions in which magnetic fluxes cancel each other. Under this assumption, when both the current sources I1 and I2 are ON, the total degeneration inductance becomes small and the device operates with high gain and low noise. Here, it is assumed that the variable resistor Rv is in an open state. On the other hand, when only the current source I1 is turned on and the current source I2 is turned off, the total degeneration inductance increases and operates with low distortion, but no bias current flows through the transistors M3 and M4. , The real part of the impedance disappears. When the resistance value of the variable resistor Rv is set to ON and an appropriate resistance value is set for the variable resistor Rv in order to compensate for the real part of the impedance, the input impedance can be made close to that when operating in the high gain and low noise mode. .
[0162]
When the current control by the variable current source I2 is controlled by two or more values, such as stepwise or continuous, the resistance value of the variable resistor Rv may be controlled by stepwise or continuous change. In such a case, the voltage φ applied to the gate may be changed in a type using an FET as shown in FIG.
[0163]
FIG. 28 shows a circuit configuration in which the gain can be made variable even in the circuit shown in FIG.
[0164]
The MOS transistors M11a and M11b whose gates are grounded are interposed between the MOS transistor M1 and the inductor L3 in FIG. 24. The transistor M11a is turned ON / OFF by a gate signal φ2, and the gate of the transistor M11b is connected to, for example, a power supply. Is always ON.
[0165]
The MOS transistor M12 is arranged so that the drain of the MOS transistor M1 is connected to the power supply potential Vdd, bypassing the inductor L3 and the MOS transistor M11a. This MOS transistor M12 is driven by a gate signal φ1.
[0166]
Similarly, the MOS transistors M21a and M21b are interposed between the MOS transistor M2 and the inductor L4, the transistor M21 is turned on / off by the gate signal φ2, and the transistor M21b is always turned on. The MOS transistor M22 is arranged such that the drain of the MOS transistor M2 is connected to the power supply potential Vdd, bypassing the inductor L4 and the MOS transistor M21. This MOS transistor M22 is driven by a gate signal φ1.
[0167]
It is assumed that the transistors M11a and M12 and the transistors M21 and M22 are constituted by MOS transistors of the same size, and the transistors M11b and M21b are constituted by MOS transistors of a small size. When the gate signal φ1 is turned off and the gate signal φ2 is turned on, a signal supplied via the transistor M11a, the transistor M11b, the transistor M21a, and the transistor M21b is output. On the other hand, when the gate signal φ1 is turned on and the gate signal φ2 is turned off, only the signal supplied through the transistors M11b and M21b is output, and the other signals are output through the transistors M12 and M22. The gain is low because it is thrown away. By using this gain switching operation together with the above-described gain switching for changing the degeneration inductor, a wider gain switching width can be realized.
[0168]
FIG. 29 shows a circuit example in which an inductance variable circuit is applied for input impedance adjustment.
[0169]
25. An impedance adjusting circuit 160 having a circuit configuration similar to that of the transistor M1 and the inductor L1, the transistor M2 and the inductor L2, the transistor M3 and the inductor L5, and the transistor M4 and the inductor L6 instead of the variable resistor Rv shown in the circuit shown in FIG. Is provided. That is, the drain of the MOS transistor M5 is connected to the power supply potential Vdd, the source is connected to the current source I3 via the inductor L7, and the input signal Input_1 is supplied to the gate.
The drain of the MOS transistor M6 forming a differential pair with the MOS transistor M5 is connected to the power supply potential Vdd, the source is connected to the current source I3 via the inductor L8, and the input signal Input_2 is supplied to the gate. .
As circuits for controlling the inductances of the inductors L7 and L8, there are provided MOS transistors M7 and M8 for inputting the input signals Input_1 and Input2 of the differential amplifier 162 to the gates, respectively. The source of the MOS transistor M7 is connected to an inductor L9 having an inductor L7 and a coupling coefficient k3, and is grounded via a variable current source I4. Similarly, the source of the MOS transistor M8 is connected to the variable current source I4 via the inductor L8 and the inductor L10 having the coupling coefficient k4. The drains of the MOS transistors M7 and M8 are connected to the power supply potential Vdd.
[0170]
To simplify the description, the inductances of the inductors L1 to L10 are set to the same value, and the coupling coefficients are set as k1 = k3 = k2 = k4. The variable current sources I2 and I4. It is assumed that inductors that change and couple in two values of current ON / OFF are connected so as to mutually weaken a magnetic field.
[0171]
When the variable current source I2 is turned on, the inductance values of the degeneration inductors L1 and L2 appear to be small, and operate in a high gain and low noise mode. At this time, if the variable current source I4 is turned off, the impedance adjustment circuit operates in a pseudo low distortion mode, and the total input impedance of the high gain / low noise mode circuit and the low distortion mode circuit is reduced. It becomes the value connected in parallel.
[0172]
On the other hand, when the variable current source I2 is turned off, the inductance values of the degeneration inductors L1 and L2 appear large and operate in the low distortion mode. If the variable current source I4 is turned on at this time, the impedance adjustment circuit operates in a pseudo high-gain, low-noise mode, and the total input impedance is low. The circuit in the high gain / low noise mode and the circuit in the low distortion mode have a value connected in parallel, and do not change even when the operation mode is switched.
[0173]
Further, in the above-described embodiment, a circuit example of a differential pair has been described, but a similar effect can be obtained with a single-phase circuit. FIG. 30 is a circuit diagram showing an embodiment applied to a single-phase amplifier circuit.
[0174]
As shown in FIG. 30, the source of the MOS transistor M1 whose gate receives the input signal Input_1 is connected to the current source I1 via the degeneration inductor L. The other end of the current source I1 is grounded. Further, an inductor L3 is connected to the drain of the MOS transistor M1, and the other end of the inductor L3 is connected to the power supply potential Vdd. An output signal Output_1 is output from the drain of the MOS transistor M1.
The control unit includes a MOS transistor M2 for inputting an input signal Input_1 to a gate. The source of the MOS transistor M2 is connected to an inductor LC having an inductor L and a coupling coefficient k, and is grounded via a variable current source I2. Have been. The drain of the MOS transistor M3 is connected to the drain of the MOS transistor M1, and the current flowing from the variable current source I2 via the MOS transistor M2 is added to the output signal Output_1.
[0175]
The value of the inductance of the degeneration inductor L can be controlled by, for example, ON / OFF control of the power supply of the variable current source I2.
[0176]
The degeneration inductor described above and a control inductor electromagnetically coupled with the coupling coefficient k, for example, the inductors L1 and L2 shown in FIG. 21 can be realized by various configurations. For example, as shown in FIG. 17, the spiral conductor forming the degeneration inductor L and the spiral conductor forming the control inductor LC can be configured to be symmetrical. In this case, the respective conductors intersect at the center of the drawing, and by forming this portion via an insulating film, it is possible to realize a single-layer wiring except for the intersection region. The inductor is not limited to the configuration shown in FIG. 17, but may have any configuration as long as it can be electromagnetically coupled, and may be formed of a conductor pattern as shown in FIG. 18 or FIG.
[0177]
Although the MOS transistor is used in the circuit of the embodiment of the present invention described above, it goes without saying that an active element such as another transistor may be used.
[0178]
The amplifier according to the embodiment of the present invention can be used for a wireless communication terminal such as a mobile phone. FIG. 31 shows an example thereof. FIG. 31 shows a block diagram of the wireless communication terminal.
[0179]
An RF input signal from the antenna (ANT) is supplied to an RF signal processing unit (RF). That is, in the RF signal processing unit (RF), the signal is supplied to the RF bandpass filter 1 (RF-BPF1), the low noise amplifier (LNA), and the RF bandpass filter 2 (RF-BPF2) via the switch (T / R). Then, the signal is multiplied by a local signal (RF-VCO) by a multiplier (DC) and frequency-converted into an intermediate frequency signal. This intermediate frequency signal is supplied to an intermediate frequency processing unit (IF-Stage) and a baseband signal processing unit (BB-Stage).
[0180]
The low noise amplifier (LNA) is supplied with a gain control signal (Gain Control) from a reception field strength determination unit (RSSI) in a baseband signal processing unit (BB-Stage).
[0181]
The transmitted signal is processed in the reverse of the above. That is, the signals supplied from the baseband signal processing unit (BB-Stage) and the intermediate frequency processing unit (IF-Stage) are processed by the RF signal processing unit (RF). In the RF signal processing unit, the intermediate frequency signal is multiplied by a local signal (RF-VCO) in a multiplier (UC) and frequency-converted, and the converted signal is passed through an RF band-pass filter (BP-BPF). The power is supplied to a power amplifier (PA) and supplied to an antenna (ANT) via a switch (T / R).
[0182]
The above-described amplifier is used for a low noise amplifier (LNA) of such a wireless terminal.
[0183]
There are various standards for wireless terminals, but when the level of an input RF signal is low, a low-noise amplifier (LNA) may be required to have amplifier characteristics for amplifying signals with low noise. is there. In such a case, control is performed to reduce the inductance value for degeneration of the LNA. Conversely, if the RF signal is sufficiently large, it is important not to distort the RF signal, so that the LNA degeneration inductance value is controlled to be large.
[0184]
Such characteristic control of the low noise amplifier (LNA) can be performed using, for example, a gain control signal (Gain Control). The amplifier characteristic control may be performed based on criteria other than RSSI.
[0185]
As described above, by using the amplifier having the variable amplification characteristic of the present invention for the LNA of the RF processing unit of the wireless terminal, it is possible to adaptively realize the desired LNA amplification characteristic without increasing current consumption.
[0186]
In the amplifier described with reference to FIGS. 21 to 31, at least a pair of inductances La, Lc, L1, L5, L2, L6, L7, L9, L8, and L10 are:
The description is given assuming that they are mutually coupled by coupling coefficients k1, k2, k3, and k4 and have mutual inductance. However, the amplifier can be realized without the pair of inductances being coupled to each other and having no mutual inductance. An amplifier using mutual inductance has an advantage that a large degeneration can be realized and a good distortion characteristic can be realized with a smaller current. On the other hand, an amplifier having an inductance having no mutual inductance does not need to use an inductor having a complicated shape, and thus has an advantage that the design becomes easy.
[0187]
Hereinafter, an amplifier including an inductor having no mutual inductance according to another embodiment of the present invention will be described in detail with reference to FIGS.
[0188]
In the following description, an example using a bipolar transistor will be described, but it is also possible to use another active element such as an FET.
[0189]
FIG. 32 shows a basic circuit configuration of an amplifier according to another embodiment of the present invention. As shown in FIG. 32, the plurality of amplification stages A1 to An are connected in parallel to the input side Input, and a gain switching function is realized by selectively operating the amplification stages A1 to An. . Here, the amplification stages A1 to An may have the same amplification characteristics, may have different amplification characteristics, or may have the same and different amplification characteristics. When the gain is switched, the input impedance changes, but the change in the input impedance Zin is compensated by the variable resistor Rx. That is, the value of the variable resistor Rx is set to an appropriate value so that the input impedance Zin of the entire amplifier does not greatly change when any of the amplification stages A1 to An is operating. Here, for example, the amplification stage A1 is an amplification stage having high gain and good distortion characteristics with a certain amount of current consumption, and the amplification stage A2 is an amplification stage having low gain and good distortion characteristics with small current consumption. Then, by operating the amplification stage A1 when a high gain is required, and by operating the amplification stage A2 when a low gain is required, desired gain and distortion characteristics can be realized and current consumption is required. Can be minimized. By setting the variable resistance Rx to an appropriate value, it is possible to suppress a change in the input impedance Zin when the gain is changed by switching the operation of the amplification stages A1 and A2.
[0190]
FIG. 33 shows a circuit according to a first embodiment for realizing the variable resistor Rx of the circuit of FIG. The variable resistor Rx for adjusting the input impedance is realized using fixed resistors R1 to Rm and switches SW1 to SWm. By appropriately switching the switches SW1 to SWm, the fixed resistors R1 to Rm are selected, and a desired resistance value is given to the input-side resistor Rx.
[0191]
FIG. 34 shows a circuit according to a second embodiment for realizing the variable resistor Rx of the circuit of FIG. The variable resistor Rx for adjusting the input impedance is constituted by the FET 172, and a desired resistance value can be realized by applying an appropriate control voltage to the control gate Vctrl of the FET 172.
[0192]
FIG. 35 shows a circuit according to a third embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit shown in FIG. 35, two grounded emitter stages A1 and A2 are connected in parallel, and a MOSFET is connected to the input stage to adjust the input impedance. The degeneration inductor L1 of the amplification stage A1 has a small inductance value so that a high gain can be realized. Further, the degeneration inductor L2 of the amplification stage A2 has a large inductance value so as to realize low gain and good distortion characteristics. FIG. 36 shows the result of simulating the input impedance (admittance) of the circuit shown in FIG. 35 except for the MOSFET 172 in the input section. FIG. 36 shows the first case where the amplification stage A1 is turned on and the amplification stage A2 is turned off to realize a high gain by appropriately setting the circuit constant and the operating point, and the amplification stage A1 is turned off, and The admittance shows the reflection coefficient in the second case where the amplification stage A2 is turned on to realize a low gain. As is apparent from FIG. 36, the impedance in the first and second cases shown in FIG. 36 is substantially reduced by connecting appropriate resistors in parallel, that is, by turning on the MOSFET 172 connected to the input section. It is possible to do the same. The current consumption for realizing a high gain is about 3 mA, the current consumption for realizing a low gain is about 1.5 mA, and the current consumption is low at a low gain. When the gain is low, the current consumption is small, but since the value of the degeneration inductor L2 is large, good distortion characteristics can be realized.
[0193]
FIG. 37 shows a circuit according to a fourth embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit shown in FIG. 37, two differential amplifier stages A1 and A2 are connected in parallel, and a MOSFET 172 is connected to the input stage to adjust the input impedance. Here, when realizing a high gain, both the amplification stages A1 and A2 are turned on. When the two amplifiers A1 and A2 are operated, the inductor of the degeneration looks small, so that a high gain can be realized. In order to realize a low gain, only the amplification stage A1 is turned on and the amplification stage A2 is turned off, so that only the inductor L1 can be seen in the degeneration inductor. It is possible to realize various distortion characteristics.
[0194]
FIG. 38 shows a circuit according to a fifth embodiment for realizing the variable resistor Rx of the circuit of FIG. In the circuit of FIG. 38, a series circuit of fixed resistors R1 to R3 and switches SW1 to SWm is connected in parallel between inputs Input_1 and Input_2. In this circuit, the input side resistor Rx can be appropriately set by selectively turning on and off the switches SW1 to SWm. That is, by appropriately switching the switches SW1 to SWm, the fixed resistors R1 to Rm are selected, and a desired resistance value is given to the resistor Rx on the input side.
[0195]
FIG. 39 shows a circuit example in which the amplifier circuit shown in FIG. 32 is applied to a low-noise amplifier of a wireless device. In the wireless terminal circuit shown in FIG. 39, an RF input signal from an antenna (ANT) is input to an RF signal processing unit (RF), and the RF signal processing unit (RF) will be described with reference to FIG. Supplied to the low noise amplifier (LNA). An output signal from the low noise amplifier (LNA) is multiplied by a local signal (RF-VCO) in a multiplier (DC) and frequency-converted to an intermediate frequency signal. This intermediate frequency signal is supplied to an intermediate frequency amplifier (IF-AMP) of an intermediate frequency processing unit (IF-Stage) via a band pass filter (BPF). The output from the intermediate frequency amplifier (IF-AMP) is supplied to a baseband signal processing unit (BB-Stage) via a quadrature demodulator (QDEM) and processed.
[0196]
In this wireless terminal circuit, the amplifier circuit shown in FIG. 32 is applied to a low-noise amplifier to realize gain switching and to keep input impedance constant. The gain switching function widens the dynamic range of the wireless device and does not change the input impedance of the low-noise amplifier. Therefore, it is possible to always easily match a desired impedance such as 50Ω using a single input matching circuit. Further, a circuit used for a wireless device is required to have low power consumption. However, since a low-noise amplifier to which the present invention is applied can be operated with a minimum necessary current, the low noise This also leads to power consumption.
[0197]
It should be noted that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying constituent elements in an implementation stage without departing from the scope of the invention.
[0198]
Various inventions can be formed by appropriately combining a plurality of constituent elements disclosed in the above embodiments. For example, some components may be deleted from all the components shown in the embodiment. Further, components of different embodiments may be appropriately combined.
[0199]
【The invention's effect】
As described above, according to the present invention, by using a distributor that can be formed on an integrated circuit to change the inductor by electromagnetic means, the electrical characteristics are good, and miniaturization and cost reduction are achieved. And a variable inductor suitable for integration into an integrated circuit can be provided.
[0200]
Further, according to the present invention, it is possible to obtain an oscillator that controls the oscillation frequency by changing the inductance value.
[0201]
Further, according to the present invention, it is possible to obtain an amplifier whose amplification characteristic can be controlled by changing the inductance value.
[Brief description of the drawings]
FIG. 1 is a circuit diagram schematically showing a variable inductor according to a first embodiment of the present invention.
FIG. 2 is a circuit diagram schematically showing a modification of the variable inductor shown in FIG.
FIG. 3 is a circuit diagram schematically showing a variable inductor according to a second embodiment of the present invention.
FIG. 4 is a circuit diagram schematically showing a modification of the variable inductor shown in FIG. 3;
FIG. 5 is a circuit diagram schematically showing a variable inductor according to a third embodiment of the present invention.
FIG. 6 is a circuit diagram schematically showing a variable inductor according to a fourth embodiment of the present invention.
FIG. 7 is a circuit diagram schematically showing a variable inductor according to a fifth embodiment of the present invention.
FIG. 8 is a circuit diagram schematically showing a variable inductor according to a sixth embodiment of the present invention.
FIG. 9 is a circuit diagram schematically showing a variable inductor according to a seventh embodiment of the present invention.
FIG. 10 is a circuit diagram schematically showing a variable inductor according to a seventh embodiment of the present invention.
FIG. 11 is a block diagram schematically showing an oscillator according to an embodiment of the present invention.
FIG. 12 is a block diagram schematically showing an oscillator according to another embodiment of the present invention.
13 is a block diagram schematically showing a circuit in which the voltage-current conversion circuit shown in FIG. 12 is configured by cascode-connected transistors.
FIG. 14 is a block diagram schematically showing a transmitter according to still another embodiment of the present invention in which a plurality of differential pairs are arranged.
FIG. 15 is a block diagram schematically showing an oscillator using a variable phase shifter and a variable gain amplifier according to still another embodiment of the present invention.
FIG. 16 is a block diagram schematically showing a Colpitts oscillation circuit according to still another embodiment of the present invention.
FIG. 17 is a plan view schematically showing a circuit pattern of the inductor shown in FIGS. 1 to 16;
FIG. 18 is a plan view schematically showing another circuit pattern of the inductor shown in FIGS. 1 to 16;
FIG. 19 is a plan view schematically showing another circuit pattern of the inductor shown in FIGS. 1 to 16;
FIG. 20 is a block diagram schematically illustrating a wireless terminal including the oscillator illustrated in FIGS. 1 to 16;
FIG. 21 is a block diagram schematically showing an amplifier according to an embodiment of the present invention.
FIG. 22 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG.
FIG. 23 is a block diagram showing a differential amplifier according to another embodiment of the present invention.
FIG. 24 is a circuit diagram showing a specific circuit of the differential amplifier shown in FIG.
25 is a circuit diagram showing another specific circuit of the differential amplifier shown in FIG.
26 (a) and (b) show circuit examples of the variable resistor shown in FIG. 23.
FIG. 27 is an equivalent circuit diagram showing the input impedance of the amplification stage shown in FIGS. 24 and 25.
FIG. 28 is a circuit diagram showing a circuit according to a modification of the circuit shown in FIG. 25.
FIG. 29 is a circuit diagram showing still another specific circuit of the differential amplifier shown in FIG.
FIG. 30 is a circuit diagram showing a circuit according to a modified example of the differential amplifier shown in FIG.
FIG. 31 is a block diagram schematically showing a wireless terminal incorporating the amplifier described with reference to FIGS. 23 to 30;
FIG. 32 is a block diagram showing a basic circuit configuration of an amplifier according to another embodiment of the present invention.
FIG. 33 is a block diagram showing a circuit for realizing the variable resistor shown in FIG. 32;
FIG. 34 is a block diagram showing another circuit for realizing the variable resistor shown in FIG. 32;
FIG. 35 is a block diagram showing still another circuit for realizing the variable resistor shown in FIG. 32;
36 is a graph showing the result of simulating the input impedance of the circuit shown in FIG. 35 excluding the MOSFET of the input section.
FIG. 37 is a block diagram showing still another circuit for realizing the variable resistor shown in FIG. 32;
FIG. 38 is a block diagram showing another circuit for realizing the variable resistor shown in FIG. 33;
39 is a block diagram illustrating a circuit example of a wireless terminal in which the amplifier circuit illustrated in FIG. 32 is applied to a low-noise amplifier of a wireless device.
[Explanation of symbols]
11 ... Signal input terminal
12 ... Distributor
13a to 13n: signal path
14a to 14c, 76a, 76b, 77a, 77b ... inductor
15, 16 ... variable inductor terminals
21a, 21b: Common source transistor
22 ... Load circuit
23a, 23b ... current sources
24a, 24b: High frequency bypass capacitor
31a to 31n: First transistor (common-gate transistor)
32a to 32n: second transistor (gate-grounded transistor)
34a to 34n: control signal input terminals
41a, 41b... Transistors of source follower circuit
43a, 43b ... current sources
51: first transistor
52 ... second transistor
53, 54 ... current sources
55... Third transistor (cascode transistor)
56a, 56b... Fourth transistor (cascode transistor)
57a, 57b ... control signal input terminals
61 ... transistor
62 ... Buffer circuit
63… Current source
64: High frequency bypass capacitor
71a, 71b... First differential pair transistors
72a, 72b... Second differential pair transistors
73a, 73b, 74a, 74b, 75a, 75b... Cascode transistors
L0, L11, L1n, L01, L02, L1, L2, L2n ... inductor
k, k1, kn, k1n, k2n ... coupling coefficient
VC1, VC2 ... variable capacitor
C1, C2, C11, C21, C1n, C2n ... capacitors
L, L1, L2 ... degeneration inductor (first inductor)
L ', L5, L6 ... second inductor
102: Core circuit (VCO core)
104, 104-1 to 104 -n: oscillation frequency control unit
105 ... current source
106 ... voltage controlled oscillator circuit
VC1, VC2 ... variable capacity
T1, T2 ... MOS transistors
T11 to T16: Transistor
T11 to T1n: transistors
T21 to T2n: transistors
I0-In ... Variable current source
116, 118: Variable phase circuit
120, 122: Variable phase shifter
130-1, 130-2, 130-3 ... antenna
132-1, 132-2, 132-3 ... Low Noise Amplifier
134-1, 134-2, 134-3 ... mixer (MIX)
136-1, 136-2, 136-3 ... Intermediate frequency processing unit (IF)
M1, M2, M11a, M11b, M21a, M21b, M22... MOS transistors

Claims (25)

A signal input terminal to which an input signal is input,
A distributor that includes first and second active elements, and that changes a distribution ratio of first and second currents flowing through the first and second active elements;
An output terminal from which the first and second currents are output;
A first inductor through which the first current flows, and a second inductor through which the second current flows, the second inductor being mutually coupled to the first inductor;
A variable inductor comprising: The variable inductor according to claim 1, wherein the first and second active elements include first and second transistors, each of which is grounded at the source or grounded at the emitter. The first and second active elements each have a gate terminal or a base terminal commonly connected to the signal input terminal, and a source terminal or an emitter terminal connected to one ends of the corresponding first and second inductors, respectively. 2. The variable inductor according to claim 1, further comprising first and second transistors. The first and second active elements each have a gate terminal or a base terminal commonly connected to the signal input terminal, and a drain terminal or a collector terminal connected to one end of the first and second inductors. 2. The variable inductor according to claim 1, including a first and a second transistor. 2. The controller according to claim 1, wherein the control unit includes first and second current sources that supply first and second currents, and at least one of the first and second current sources varies the supplied current. 3. A variable inductor as described. 2. The variable inductor according to claim 1, wherein the first and second active elements include first and second transistors that are grounded at a gate or grounded at a base. The variable inductor according to claim 1, wherein the distributor includes a source follower circuit or an emitter follower circuit. The variable inductor according to claim 1, wherein the distributor includes a cascode connection circuit. An amplifier having an input terminal connected to the signal input terminal and an output terminal connected to the first inductor;
2. The variable inductor according to claim 1, further comprising a variable gain buffer circuit having an input terminal connected to an input terminal or an output terminal of the amplifier, and an output terminal connected to the second inductor. The variable inductor according to claim 1, wherein the distributor includes a differential circuit. A voltage controlled oscillation circuit having a first inductor;
A frequency control circuit having a first inductor and a second inductor that is electromagnetically coupled to each other, and supplies a control current to the second inductor. A frequency control circuit that controls the oscillation frequency of the voltage-controlled oscillation circuit by changing the inductance value of the inductor;
An oscillator comprising: The oscillator according to claim 11, wherein the frequency control circuit includes the plurality of second inductors, and controls a control current flowing through each of the plurality of second inductors. The oscillator according to claim 11, wherein the frequency control circuit changes at least one of an amplitude value and a phase of a current flowing through the second inductor. The oscillator according to claim 11, wherein the frequency control circuit includes a plurality of second inductors that are coupled with different coupling coefficients from the first inductor. A wireless terminal comprising the oscillator according to claim 11 and supporting a plurality of communication systems having different frequency bands. An amplifier circuit having a first inductor;
A control unit that includes a second inductor that is electromagnetically coupled to the first inductor and supplies a control current to the second inductor. The control unit changes the control current to change the control current of the first inductor. A control unit that changes an inductance value and changes an amplification characteristic of the amplification circuit;
An amplifier comprising: 17. The amplifier according to claim 16, wherein the control unit changes at least one of an amplitude value and a phase of a current flowing through the second inductor. 18. The amplifier according to claim 17, further comprising an input impedance control unit that controls an input impedance of the amplifier at an input unit of the amplifier. 19. The amplifier according to claim 18, wherein the input impedance control unit includes a variable resistor. A wireless terminal comprising the amplifier according to claim 18 as a low noise amplifier of an RF signal processing unit. A variable gain amplification stage in which a plurality of amplification stages are connected in parallel;
An input impedance adjustment circuit that includes a variable resistor connected to the input unit and adjusts a change in input impedance that occurs when the gain is switched by adjusting the resistance value of the variable resistor;
An amplifier comprising: 22. The amplifier according to claim 21, wherein the variable resistor connected to the input comprises a plurality of fixed resistors and a switch for selecting the fixed resistors. 22. The amplifier according to claim 21, wherein the amplifying stages have different amplification characteristics, and change the gain of the variable gain amplifying stage by selectively operating an operating amplifying stage. 22. The amplifier according to claim 21, wherein the amplification stages have the same amplification characteristics and change the gain of the variable gain amplification stage by selectively operating an operating amplification stage. A wireless communication device comprising the amplifier according to claim 21 as a low noise amplifier of an RF unit of the wireless communication device.

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